RU2138922C1 - Method for controlling image repetition blanking circuits for tv sets and video cassette recorders - Google PatentsMethod for controlling image repetition blanking circuits for tv sets and video cassette recorders Download PDF
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- RU2138922C1 RU2138922C1 RU93053753A RU93053753A RU2138922C1 RU 2138922 C1 RU2138922 C1 RU 2138922C1 RU 93053753 A RU93053753 A RU 93053753A RU 93053753 A RU93053753 A RU 93053753A RU 2138922 C1 RU2138922 C1 RU 2138922C1
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- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
- H04N5/00—Details of television systems
- H04N5/14—Picture signal circuitry for video frequency region
- H04N5/21—Circuitry for suppressing or minimising disturbance, e.g. moiré or halo
- H04N5/211—Ghost signal cancellation
The invention relates to schemes for suppressing image repetitions, both used in a television receiver and VCR, and, more importantly, to methods for calculating filter parameters used to suppress image repetitions in composite video signals supplied from a video detector of a television or VCR.
Background to the invention.
Television engineers have devoted considerable attention to repeated image suppression schemes included in television sets, which also include a visual display device for reproducing an image in a form suitable for human vision. Multiple images caused by multi-path reception and commonly referred to as “repeats” typically occur in television images that are distributed through the air or transmitted via cable.
The television receiver is synchronized by the strongest signal it receives, which is called the call signal, and it is usually a direct signal received via the shortest reception path. Multipath signals transmitted along other paths are usually delayed in this way with respect to the reference signal and appear as lagging repeated images, however, it is possible that a direct signal or a signal arriving along the shortest path is not a synchronization signal of the receiver. When the receiver is synchronized by the reflected (long path) signal, there will be a leading repeated image caused by the direct signal, or there will be many leading repeats caused by the direct signal and other reflected signals arriving with a lower delay than the reflected signal along which the receiver is synchronized. The parameters of multi-path signals, that is, the number of messages arriving on different paths, their relative amplitudes and the difference in delay times between different messages arriving on different paths, vary from location to location and from channel to channel at a given location. These parameters may also change over time.
The visual effects of multipath distortion may well be qualified in two categories: multiplicity of images and distortion of the channel frequency response. Both effects are due to temporary and amplitude changes among the multipath signals arriving at the receiving site. When the relative delays of the multipath signals with respect to the reversal signal are substantially large, the visual effect is observed as multiple copies of the same image on the screen, horizontally offset from each other. These copies are sometimes referred to as “macro repeats,” in contrast to the “micro repeats,” which will now be described. In the usual case, when the direct signal dominates and the receiver is synchronized by the direct signal, the repeated images are shifted to the right with variable position, intensity and polarity. This is known as lagging retries or "post-repeat" images, which will now be described. In the more rare case, when the receiver is synchronized with the reflected signal, there will be one or more repeated images shifted to the left of the transmitted image. This is known as leading repeats or "pre-repeat" images.
Multipath signals with relatively short delays with respect to the reference signal do not cause separately distinguishable copies of the prevailing image, but cause distortion of the channel frequency response. The visual effect in this case is observed as an increase or decrease in image sharpness and, in some cases, loss of some image information. These short-circuited, close-up, or nearby repeats are usually caused by the disconnection or improper connection of the radio frequency transmission lines, such as antenna inputs or subscriber cable taps. In a cable television environment, many approximate repeats are possible, caused by reflections distorted by several incorrectly connected subscriber taps of various lengths. Many of these close repeats are commonly referred to as “micro repeats.”
The effects of long multi paths or macro repeats are usually reduced by blanking schemes. The effects of short multi-paths or micro-repeats are mitigated by equalizing the waveform, in most cases by attenuating and / or compensating for the delay group of the video frequency characteristic.
Since the characteristics of a transmitted television signal are initially known to be possible, at least in theory, to use such characteristics in a system for detecting signal repetition and blanking. Nevertheless, various problems limit this approach. Instead, it is desirable to find a reference signal under certain conditions for repeated transmission, for example, in a section of television signals that are not currently used for video purposes, and use this reference signal to detect repeat signals before preparing to cancel the repeat signals. Typically, lines are used in a vertically blanked interval (VBI) (VBI). Such a signal in this case is called a reference blanking signal - the GCP signal (GCR) and many different GCP signals are described in patents and other technical publications.
As a standard for television in the United States of America, it is recommended to accept chirp (linear frequency modulated signal) Bessel pulsed signals used in UCP signals. The energy distribution in the Bessel chirp pulse signal has a wide frequency spectrum, continuously stretched around the video frequency band. The chirp starts at the lowest frequencies and continues up to a frequency of 4.1 MHz. The chirp is placed in the first half of the selected nosocomial lines, the 19th line of each field is consistently preferred. The chirp, which is based on +30 IRE, ranges from -10 to +70 IRE and there is a prescribed time after the trailing edges of the previous horizontal synchronization pulses. Chirp signals appear in a cycle of eight fields, in which the first, third, fifth and seventh fields have a polarity of the color pulse defined as positive, and the second, fourth, sixth and eighth fields have the opposite polarity of the color pulse, defined as negative. The initial fraction of the ETR chirp signal, which appears in the first, third, sixth and eighth fields of a cycle of eight fields, fluctuates upwards from the base +30 IRE to the level +70 IRE. The initial share of the chirped ETR signal, which is manifested in the second, fourth, fifth and seventh fields of the eight-field cycle, fluctuates down from the +30 IRE base to the level of -10 IRE and is a complement to the chirp ETR signal.
The strategy for the elimination of shadows in a television receiver is based on the transmission of an OGP signal experiencing the same multipath distortions as other television signals. The circuits in the receiver can then verify the received distorted GCP signal, and with the initially known voluntarily distorted signal, a procedure known as a channel characteristic can be performed in which the magnitudes, phases, and times of the event are determined according to the reference signal. This is done by calculating the discrete Fourier transform of the DFT (DFT) of the OGP signal with repetition and dividing the members of this DFT by the corresponding members of the DFT of the OGP signal without repetition, known in advance, so as to generate the corresponding members of the DFT channel. All of these DFTs are present in the time interval. The temporary presence of repeats and the amplitudes of their infase components are then used to calculate the adjustable load coefficients of the digital filter through which the composite signal from the video detector passes to transmit a message in which the repeats are canceled, this filter is called a “blanking filter” in this description. The members of the channel DFT are analyzed to determine the largest of them, which is replaced by zero in the modified DFT. Other members replace the opposite characters in the modified DFT, which is the required DFT that suppresses filter repeats. The load coefficients of the filter blanks are adjusted to get as close to this desired DFT as possible, and the DFT signals can also be used to calculate the adjustable load coefficients of a stabilizing filter connected to the cascade with a filter for blanking to provide the desired width of the frequency spectrum characteristic upon completion of the reception path through the amplitude modulator of the transmitter of the remaining part of the range, the transmitting medium, the front panel of the television receiver and the cascade and quencher repeats, and a stabilizing filter.
The inventors constructed a shadow-suppression (blanking) filter as a cascade connection of a recursive digital filter, mainly used for blanking post-repeated images, and a non-recursive digital filter, mainly used for blanking pre-repeated images. A recursive digital filter has an infinite impulse response (IIR filter), therefore it is usually referred to as an IIR filter. A non-recursive digital filter has a limited impulse response (FIR filter), therefore it is usually referred to as a FIR filter. You can try to independently adjust the IIR IIR and FIR FIR filters by directly linking the smallest DFT terms of the aforementioned receive channel, later in time than the largest DFT term of the aforementioned receive channel for tap loads, in the IIR filter, and directly linking the smallest DFT terms of the aforementioned receive channel earlier in time than the largest DFT term of the aforementioned receive channel for tap loads in the FIR filter. Direct binding of DFT members to tap loads in filters is a relatively simple computational procedure that was pursued in the prior art when it was post-iterations or pre-iterations that were corrected. Suppose then these parts of the visual results of the characteristics of the post-retry channel are used to adjust the filtering coefficient of the IIR filter, which is usually used to suppress post-retries.
Suppose further that these parts of the visual results of the characteristics of the pre-repeat channel are used to adjust the filtering coefficient of the FIR filter, which is usually used to quench pre-repeats. When the regulation of the IIR filter and the regulation of the FIR filter are carried out independently, the repetitions are well quenched if only post-repetitions are present and not in a very large amount. Repetitions are also well extinguished if only pre-repetitions are present and in not very large numbers.
Repeat cancellation tends to be bad, however, when both post-repetitions and pre-repetitions of high energy are present, even this reduction in repetitions is significant. The problem of too many post-repeats and too different delays between these repeats can be solved by using an IIR filter with a large number of taps with non-zero load factors and by increasing the number of devices with a programmable delay. The problem of too many pre-repeats and too different delays between these repeats can be solved by using an FIR filter of a more complex configuration. When both post-retries and pre-retries are quenched, however, the problem of weak quenching of retries is not resolved by dividing into separate problems the post-retries and the problem of pre-retries.
The inventors have noticed that a good blanking occurs when one of the filters of the cascade does not significantly respond to the reaction of the others. The problem is that when both post-retries and pre-retries are to be extinguished, the characteristics of the IIR and FIR filters are interactive with each other. To illustrate this interaction, let us assume that the IIR filter precedes the FIR filter in their cascade connection with each other. For each post-repetition extinguished by the IIR filter, the pre-repetition applied to this filter will give an increase in repetition of the pre-repetition image. Repeat from the pre-repeat image lags behind the pre-repeat image by the same interval that the canceled post-repeat lags behind the dominant signal.
SUMMARY OF THE INVENTION
The blanking scheme is preferably used with the methods of the invention; repeats accompanying a complex video signal from a video detector in a television receiver or VCR are quenched in a cascade of three digital filters having filtered parameters that are adjustable by digital programmable signals; one, an IIR filter is used to blank out tracking macro repeats, or post-repeats; another, an FIR filter, is used to quench leading macro repeats, or pre-repeats, and another additional FIR filter is used to quench or "equalize" the characteristics of the receiving channel. Filtered parameters of these three digital filters are calculated according to the methods of embodiment of the invention in its various aspects.
These methods each include acquiring data and measuring channel characteristics, followed by the final step of comparing most of the main series of channel characteristics, leading to the next most basic series of channel characteristics, which determines whether the conditions for retries are constant. This final step makes the loop the inverse method by acquiring data and measuring the characteristics of the channels until constant repeat conditions are established, so the most basic series of results of the characteristics of the channels can rely on supporting the accuracy of the calculation in the next steps of calculating the filtered parameters for the IIR filter used to quench servo macro repeats and a FIR filter used to quench leading macro repeats.
After the steps of calculating the filtered parameters for the filters used to quench the macro repeats, each of these methods performs the step of calculating the filtered parameters for the FIR filter used to quench the micro repeats. This step is followed by data acquisition and further measurement of the channel characteristics, followed by the final step of comparing the most basic series of additional channel characteristics, leading to the next most basic series of additional characteristics, leading to the determination of whether they are really the same. If the series of additional channel characteristics results are indeed the same, each method will loop back to the step of additional data acquisition; if the series are noticeably different, then each method restores the adjustable filtering parameters for the filters to the initial values and the method restarts automatically.
The two methods for implementing the invention are somewhat different in aspects, each performing initial calculations of the adjustable filter parameters of the IIR filter used to quench the lagging macro repeats, regardless of the initial calculation of the regulation of the filtered parameters of the FIR filter used to quench the leading macro repeats. These initial-independent calculation techniques significantly reduce the amount of computation over previously known techniques, even taken as a result of subsequent computations, which is sometimes required to correct individual computations. In one preferred method, a subsequent approximation method, subsequent calculations to provide this correction are prepared to return to acquiring data and measuring the channel response and recalculating the adjustable filter parameters of the filters used to quench the macro repeats until the macro repeats drop below the prescribed threshold level. In another method, which is independent of subsequent approximations, subsequent calculations to provide this correction are based on the fact that the corrected DFT corresponds to the prescribed degree of DFT convolution of the frequency characteristics of the filters used to suppress macro repeats.
A brief description of the drawings.
FIG. 1 is a general schematic diagram of a system including a television receiver or video recorder with a snuff-out circuit in which the methods of the invention can be successfully applied.
FIG. 2 is a schematic diagram of a snuff-out circuit suitable for inclusion in the combination of FIG. 1, this repetitive quenching circuit includes a circuit sensing an OGP signal.
FIG. 3 is a schematic diagram of a circuit for reconstructing a field counter modulo eight in a repeat extinguishing circuit of FIG. 2.
Each of FIG. 4, 5, and 6 is a flow chart of one of two alternative quenching methods that can be used with the shadow quenching circuit of FIG. 2, a method embodying the invention in one of its aspects.
FIG. 7 is a block diagram of the steps that can be performed in step 84 of the method of FIG. 4, FIG. 5, FIG. 6 or FIG. 7.
FIG. 8 is a flowchart of the steps that may be in the first execution path of step 90 of the method of FIG. 4 or the method of FIG. 6.
FIG. 9 is a schematic diagram of one way of performing a lightly loaded IIR filter, which can be used in the methods of the invention to quench lagging macro repeats or post repeats.
FIG. 10 is a schematic diagram of one way of performing a lightly loaded FIR filter, which can be used in the methods of the invention to quench leading macro repeats or pre-repeats.
FIG. 11 is a more detailed block diagram of the steps that can be performed to calculate filterable parameters for repetitive filters formed from cascaded lightly loaded IIR filters and lightly loaded FIR filters when steps 96 and 97 of the method of FIG. 5.
FIG. 12 is a flowchart of substeps that may be performed in step 94 of the method of FIG. 5.
FIG. 13 is a more detailed block diagram of substeps that can be used to correct for “filter-generated” repeats when step 98 of the method of FIG. 5.
Various modifications that may be made to the circuits shown in FIG. 2 and 3 are marked in parentheses in these figures.
The term "television set" is used in this description to denote the front panel of a television receiver with an accompanying kinescope, a power source for a kinescope, a disconnecting circuit for a kinescope, a node of a television receiver associated with the conversion of a complex video signal to color signals to excite the kinescope, reproducer (ditch), stereo detector or sound amplifier. A conventional type BKM VCR (VCR) includes a front panel of a television receiver with associated elements called a “television monitor” in this description and drawing support. If the BKM and the television set are combined into a single hardware unit called "combo", it is desirable to be able to simultaneously record a program received on one channel and show a program received on another channel, two front panels should be provided, one for a video recorder with recording capability, and the other for a television receiver with the ability to display an image. As taught in the US patent with serial number N 07 / 955,016 in the name of Chandrakant B. Patel and Min Hung Chung on October 1, 1992, entitled "VCRs with a front panel of a television receiver and a blanking circuit" and attributed to Samsung Electronics Company Limited , it is preferable to include the respective sets of cancellation patterns after the video detectors of each of the front panels of the television receivers, although a single mic can be used to calculate the filtered parameters of both sets of cancellation schemes computer.
Raster scanning converts two-dimensional spatial regions of consecutive image fields that form a three-dimensional region in space and time into a one-dimensional temporal region of a video signal. Television engineers send this combination of various sample values of the video signal to the filtering circuits, indicating the intended results of such a combination in a three-dimensional region in space and time. A filter circuit combining sampled video signal values that places elements of a picture to be drawn, or “pixels,” along a scan line of an image field, is called a “horizontal spatial factor”. A filter circuit combining sampled values of a video signal separated in time by scanning length intervals that places selectively drawn pixels across the scan line of an image field is called a "vertical spatial filter." A filter circuit combining sample values of a video signal separated in time by the intervals of a scan line that shows selectively displayed pixels at the same positions in successive image fields or frames is called a “time filter,” that is, a filter that works better in the time domain than in any of the spatial areas. A temporal filter can combine pixels for only one spatial position, but usually combines pixels arranged appropriately from multiple frames for each particular spatial position. For example, in the averaged line filter, pixels are averaged for each position along the scan line. By analogy, which is used in this description or the following claims, the term "temporal filter" is intended to include filters for combining the corresponding "pixels" of the respective scan lines selected from successive fields, which scan lines include UCP signals. This temporal filter of a special type is referred to in this description and the following paragraphs as the “filter of the averaged UCP signal,” although the averaging is not done on the basis of real values, but only after correction according to the standard polarity of the chirp.
Storage devices that provide temporary storage of the number of samples corresponding to the positions of pixels in the video scan line are referred to in this description as “temporary storage devices of one line”, even if such storage devices can actually be included in batteries that accumulate several lines on a pixel-on basis pixel. Similarly, storage devices that temporarily store the number of samples corresponding to pixel positions in two consecutive scan lines of a video signal are called “temporary storage devices of two lines” in this description, even if such storage devices can actually be included in batteries that accumulate several pairs of sequential pixel-to-pixel lines. The inclusion of several temporary storage devices of one line or temporary storage devices of two lines in a single data bank, of course in the scope of the invention, is disclosed in this description and the corresponding drawings.
The front panel of the television receiver 20, in response to the radio frequency television signal thus received, produces an audio signal and a false video signal for devices 10, these devices 10 may be a television monitor or a VCR that receives these signals for recording. The VCR with recording power can be a VHS, Super VHS or Betamax type video cassette recorder. For example, an improved VHS recorder of the type described in US Pat. No. 1,5,113,262, issued May 12, 1992, K.X., may serve as a VCR. Stroull et alii, entitled "Video Recording System to Limit Recording and Playback Bandwidth".
An RF television signal can be transmitted over the air and then received by an air antenna 30 for use in the front panel of a television receiver, as shown in the example. Alternatively, the radio frequency television signal may be transmitted by a public antenna cable or other cable television service. The front panel of the television receiver 20 includes nodes of a conventional television receiver, normally working in combination with either a television monitor or a video recorder. These nodes typically include a radio frequency amplifier, an understanding of a converter or a "first detector", at least one intermediate frequency amplifier, a video detector or a "second detector", and an audio demodulator (usually a portable type).
The front panel of the television receiver further includes separate circuits for horizontal synchronization pulses and vertical synchronization pulses.
The sound signal from the sound demodulator on the front panel of the television receiver 20 is demodulated from a frequency-modulated sound carrier frequency, which is heterodyned into the middle frequency by a buck converter. Before demodulation, the carrier frequency-modulated sound frequency is limited in its change in amplitude bias, and the capture phenomenon dampens the characteristics of the repeats in the sound signal from the sound demodulator. Accordingly, the sound signal from the sound demodulator on the front panel of the television receiver 20 is fed directly to the device 10, where it is used in a conventional manner.
The complex video signal from the video detector on the front panel of the television receiver 20 is supplied to the blanking repeats of the circuit 40 to remove or cancel the accompanying repeats. Repeat blanking schemes 40, which may be of any type known in the art, include adaptive filter circuits and a computer for calculating filter parameters for these adaptive filter circuits. The result of the repetition-free complex video signal is supplied from the blanking circuit of repeat 40 to the device 10, where it is used in a conventional manner. The separator of the UCP signal (or the acquisition scheme of the UCP signal) 45 separates the UCP signal and its accompanying duplicate from the complex video signal from the video detector on the front panel of the television receiver 20. The separator of the UCP signal transfers the highlighted signal to it to the computer in the blanking circuit 40, in this computer, the OGP signal with repetition correlates with the initially known information about the OGP signal, which is free from repetition, as a basis for calculating filter parameters for adaptive filter circuits in repetition quenching circuits pa 40. In the invention, the GCR signal separator 45 takes the form of the circuits receiving the GCR signal, which is averaged at baseline pixel-by-pixel Bessel chirp, what is extracted from the GCR signals in a number of consecutive fields. The UCP signals, including ETR chirps, form the first series of UCP signals, and the UCP signals, including ETR chirps, form the second series of UTR signals.
Rapid changes in multipath conditions occur over time, such as when one or more planes fly over the antenna of the television receiver 30, for example, causing a rapidly changing selective attenuation, known as “airplane flutter”. The filter coefficient computer within the blanking circuit 40 typically provides a sufficient computation speed for adjusting filter parameters over time fields. However, multi-path conditions can change so quickly during airplane flutter that adjusting filter parameters by calculating the correct shape of one UCP signal with a repeat selected from the 19th line of the field is no longer intended for a later time in the field when calculating these filter parameters finished. When, in accordance with the invention, noise-reducing GCP signals are generated by averaging GCP signals from several consecutive fields, the calculation of the filter parameter correction is even less possible to track the rapid change in multi-path conditions. Accordingly, it is generally accepted that it is more practical to determine filter parameters for quench patterns 40 only after stopping a rapid change in multi-path conditions and restoring static multi-path conditions.
FIG. 2 can serve as an illustration of one form of repeat damping circuits, which is suitable for use with the BHP signals of the Bessel chirp inserted in the 19th line of the XBI of each field. The composite video signal supplied to the circuits damping the repeats in FIG. 2, it is digitized from the front of the television receiver 20 to an analog-to-digital converter (ADC) 50. The ADC 50 typically provides samples of eight parallel bits of a complex digitized video signal. The digitized composite video signal is used as an input signal for cascading a quenching post-repetition filter 51, which is adaptive, such as an IIR filter; a quench pre-filter 52, which is adaptive, such as a FIR filter; and a stabilizing filter 53, which is adaptive, such as an FIR filter.
The output of the filter cascade is a “depot” digital complex video signal, which is converted into an analog repetitive free video signal by a digital-to-analog converter 54.
An analogue repetition-free complex video signal is supplied to the instruments 10. The digital-to-analog converter 54 is prepared with an advanced design when instruments of the type using a digital signal are used as the instruments 10 better than those using the analog signal.
The filter coefficient computer 55 calculates load coefficients for the adaptive filters 51, 52 and 53. These load factors are binary numbers that the filter coefficient computer 55 writes to the registers inside the digital filters 51, 52 and 53. In the IIR filter 51, the load factors, recorded in these registers are used as multiplier signals for a digital multiplier that receives a filter output with various delay components, as multiplier signals. The product signal from the digital multiplier is algebraically combined in a digital adder / calculator to obtain an IIR filter characteristic. In each of the FIR filters 52 and 53, the load coefficients recorded in these registers are used as multiplier signals for a digital multiplier that receives the filter input signal with various delay components as multiplier signals. In each of the FIR filters 52 and 53, the product signals from the digital multiplier are algebraically combined in the digital adder / subtractor to obtain the characteristics of the weighted total dependence of the FIR filter.
Pre-repeats taking place off-air can be moved as much as 6 microseconds from the direct signal, but usually the offset is no more than 2 microseconds. In a cable receiver, direct off-air interference may precede a signal transmitted over the cable by as much as 30 microseconds. The number of taps of the FIR filters 52 and 53 depends on which region the quenched repeats are wanted. To maintain filter prices in line with commercial requirements, typically the FIR filter 52 has about 64 taps to dampen repeats with a deviation of as much as 6 microseconds from the direct signal. The FIR filter 52 used to stabilize the frequency needs only 32 taps or so. An FIR filter 53 typically requires adjustments to the in-band video response, which can usually be sweeped as much as 20 dB at 3.6 MHz, but the 3.6 MHz sweep is usually less than 10 dB. Scanning is typically a characteristic of the incorrect orientation of the antenna in an off-air receiver. The cascade of FIR filters 52 and 53 with the same design is replaced by a single FIR filter with about 80 taps.
Typically, the deviation region for post-repeats reaches 40 microseconds from the direct signal, with 70% or so of the post-repeats being located in a subregion that reaches 10 microseconds. The quench post-repeats IIR filter 51, designed to quench post-retries, can have up to 600 taps. However, since post-repetitions usually do not overlap and deviations are discrete, the load factor for many of these taps approaches zero. Branches requiring load factors significantly different from zero are combined into groups of ten or less, except for those where the repeats overlap. From the point of view of saving hardware, it is advisable to use only as many digital multipliers as it is required for the load factors to have a value significantly greater than zero. Correspondingly, the diverting delay line in the IIR filter 51 is usually designed as a cascade connection of ten-by-one branches of the delay lines alternating with programmable “load” delay devices, the resulting filter 51 is sometimes called a “light-loaded” filter. Ten or so branches of the delay line provide signals for loading digital multipliers. The incremental delay between successive taps of each of these ten or so branches of the delay lines is an interval of one pixel. Programmable load delay devices, each containing delay lines of different lengths arranged in a chain together, which can be controlled depending on the control signals supplied in the form of binary numbers. Such underloaded filters will include registers for binary numbers indicating delays of programmable delay devices; the contents of these registers are also controlled by a filter coefficient computer 55.
The FIR filter, which extinguishes repetitions, can also be made in the form of a loosely loaded filter, and this description is written taking into account this possibility. Practically speaking, pre-repetitions are mostly close to repetitions and are much ahead of the main image, so the FIR filter that extinguishes pre-repetitions is usually not an underloaded construction.
Let us now consider the ways in which UHF signals are transmitted by the filter coefficient computer 55 from the front panel of the television receiver 20. Horizontally and vertically synchronized pulses are received from the front panel 20. The horizontal clocks are read by an eight-digit digital counter 56, called a “scan line counter”, periodically reset by vertical sync pulses, and vertical sync pulses are read modulo 8 by a three-digit counter counter 57, called a “field counter”. These counters make the filter coefficient computer 55 useful for use in timing its operations, although the connections for communicating these counters with computer 55 are omitted in FIG. 2 so as not to complicate it. Decoder 58 responds to the nineteenth account of the scan lines of the counter of line 56, corresponding to the scan line in each field containing the UCP signal, provided that the output signal of the multiplexer 59 corresponds to a complex digital video signal from the output of the cascade connection of the filters 51, 52 and 53 supplied to it as the first input, which is better than supplying zero as its zero input.
The filter coefficient computer 55 controls the operation parameters of the filters 51, 52, and 53. Thus, by manipulating these operational parameters, the computer 55 can select the cascade connection point of these filters 51-53 from which the OGP signal is separated by the OGP signal separator. (The UCP signal separator includes elements 58 and 59 in FIG. 2 and elements 58 and 101-108 in FIG. 5). For example, the input signal supplied to the cascade connection of filters 51-53 can be selected in the UCP signal separator by computer 55 by setting the load coefficients of the recursive paths in the IIR filter to zero, thus, the output characteristic of the IIR filter 51 is determined solely by its input signal ; setting to zero all load factors, except for one, with a single value that defines the axial center of the filter 53. Alternatively, you can set the circuit to provide a more direct and faster selection of the point in the cascade connection of the filters 51-53, from which the OGP signal is extracted. The fact that the point in the cascade connection of the filters 51-53 from which the UCP signal is separated can be selected is important for understanding, since this fact will help in understanding the procedure for calculating the variable parameters of the filters 51-53, which is explained later in this description using the flowchart of FIG. 4.
Read-then-write random-access memory is provided by the line (scan) temporary storage device 60 of FIG. 2, this storage device 60 may be replaced by serial memory in other embodiments of repetitive quenching circuits. This is a temporary storage device of line 60 for accumulating the 19th VBI line of UCP signals based on a pixel basis for eight consecutive fields; in this temporal filtering operation, Bessel chirp information is separated from other information present on these 19 XBI scan lines. Elements 59-69 in FIG. 2 are combined to form the UHF signal of an averaging filter that performs this temporal filtering operation, which correlates the Bessel chirp information present on these 19 XBI scan lines to provide a signal-to-noise ratio, which is compared using simple gating, to extract the chirp information Bessel from 19 scan lines where it is present. When the corresponding pixels of eight GCP accumulation signals in the 19th line of FIELD 000, the eighth and last field of a sequence of eight fields, the separated Bessel chirp information is sequentially loaded one pixel at a time into the register of filter coefficient computer 55 in the continuation of any line of the field 000 after the information is highlighted Bessel chirp recorded in filter coefficient computer 55. Transfer of accumulated information from line 60 storage device to computer 55 and sequential erasure of accumulated information and from the storage device, line 60 may also occur during any two of the 1st through 18th scan lines of field 001.
In more detail, the temporary storage device of line 60 should be able to store a full scan line of a sample of sixteen parallel bits, assuming that eight lines from a sample of eight parallel bits of a digitized complex video signal supplied from the ADC 50 via the filter cascade 51 are accumulated (on an algebraic or sign basis) -53. For their symbolic, algebraic summation, complementary to two arithmetic is preferred. In a partial embodiment of the installation for temporary storage of the line 60 operating as an accumulator for UCP signals, the digital adder / subtractor 61 supplies an output signal of sixteen parallel bits to the temporary storage of the line as its recording input signal. The digital adder / subtractor 61 receives as the first input to it the output signal of the multiplexer 62, which usually corresponds to reading from a temporary storage device line 62, taken as the zero input of the multiplexer 62. The digital adder / subtractor takes as its second input output signal from eight parallel the bit of the multiplexer 59 along with eight neutral conductors as an bit-and-bit extension.
Decoder 69 decodes a field count modulo eight, which is one, three, six, or zero (i.e., eight), to provide a logical zero to digital adder / subtractor 61, provided that it adds up its input signals. Decoder 69 decodes a field count modulo eight, which is two, four, five, or seven, to provide a logical unit to digital adder / subtractor 61, provided it subtracts its second input signal (supplied from multiplexer 59) from its first input signal (supplied from multiplexer 62). This layout is accumulated in the storage device by the following function:
(FIELD 001 line 19) - (FIELD 010 line 19)
+ (FIELD 011 line 19) - (FIELD 100 line 19)
- (FIELD 101 line 19) + (FIELD 110 line 19)
- (FIELD 111 line 19) + (FIELD 000 line 19)
In the last line of the eighth field of each sequence of eight fields, as a rule, the control signal ZERO in the multiplexer 62 is called for installation in ONE. This state of multiplexer 62 ONE should provide an output signal corresponding to the first input signal in it, which is an arithmetic zero containing sixteen parallel bits of ZERO conductors. This leads to the reset accumulated in the temporary storage line 60 of the result in arithmetic zero. The control signal for multiplexer 62 is shown in FIG. 2 as generated by an I-type 63 two-input gate. Decoder 64 decodes the counter of the scan line counter corresponding to the last line of the current field to generate one of the input signals of the I-type 63 gate. Decoder 65 decodes the field count modulo eight from the counter 57 to generate other input signals of the I-type 63 valve. The eighth field of each sequence of eight fields generates a count of 000 modulo eight from the field counter 57. Both input signals of the I-type 63 gate are ONE only during the last line of the eighth field of each sequence of eight fields, during this line, the I-type 63 valve supplies ONE to the multiplexer 62, as its control signal, causing the accumulated result stored in the temporary storage device of line 60 to be reset to zero.
The I-type 66 two-way valve supplies ONE to the filter coefficient computer 55 when the accumulated result recorded in the temporary storage device of line 60 is suitable for transferring to the register of shaded Bessel chirp in the internal memory of the computer 55. The output signal of the detector 65 is one of the input signals I-type gate 66 and is ONE only during the eighth field of each sequence of eight fields. A two-input gate of type NON-OR 67 generates other input signals of an AND-type gate 66. A gate of NON-OR-type 67 receives the output signal of decoder 64, which detects the last field line in the line counter 56, and the output signal of decoder 68, which detects vertically blanked interval coming from the line counter account 56. Accordingly, the output signal of the NON-OR-type valve 67 is ONE, except for the time of the vertically blanked interval or the last line of the field. Thus, the accumulated result stored in the line temporary storage device is allowed to be transferred to the internal memory of the computer 55 at any time during the eighth field of each sequence of eight fields, except for the time of its last line or during a vertically blanked interval.
Now we will consider the synchronization for the timed pixels selected by the analog-to-digital converter 50 and the addressing of the temporary storage device of the line 60. The generator 70, which has automatic frequency and phase control, AFPC, generates sinusoidal oscillations at the second harmonic of the color frequency subcarrier as the primary clock signal. An intersection decoder with zero 71 detects the middle axis intersecting the sinusoidal oscillations in the generated pulses at a frequency four times the subcarrier of the color frequency. These pulses time the sampling from a complex video signal to digitize the ADC 50: and they will time the data advance in the temporary storage device of the line 60, if it is a serial memory. In FIG. 2, the temporary storage device of the blanking circuit line 60 is a random access memory for a read-then-write operation, while each cell of this storage device is addressable. The addresses of these memory cells are scanned recursively according to the count of pixels supplied from a ten-digit digital counter, referred to as a “pixel counter”, which counts the pulses from the intersection decoder with zero 71. The same addresses supplied to the filter coefficient computer 55, in it is used to address the register that stores the line when a separated UCP signal from the temporary storage device of line 60 is transmitted to it.
Typically, if it exists, the pulsed color signal has the most stable frequency basis in the complex video signal and is the preferred reference signal for the AFFC of the generator 70. The overflow signal from the second bit of the pixel counter 72, preferably a square 3.58 MHz signal, is provided as a feedback signal communication to the first AFFC decoder 73 for comparison with the selected pulse signal in order to generate an error signal, the multiplexer AFFC signal 74 selectively refers to the pixel counter 72 to control the frequency and of the phase of its oscillations. Pulse valve 75 receives pulses from a pulse valve control signal generator (GSUIV) 76 to isolate from the complex analog video signal supplied from the front panel of the television receiver 20, a color pulse signal supplied from the first AFFC detector 73. Horizontal clock signals from the front panel of the television receiver 20 are supplied to the control signal generator of the pulse valve 76 and their trailing edge is used to timing the pulses that the generator 76 generates during the pulse inter shafts. A cascade of unstable triggers or (one-shots) “one-shots” is commonly used to generate these pulses.
The circuits of the decoder 68 receive a scan line count, which provides a counter of lines 56, which correspond to the VBI lines in each field, to generate a prohibition signal. The inhibit signal is supplied to the control signal generator of the pulse valve to prohibit the generation of its pulses, so that the pulse valve 75 will select only these return intervals during the field, which may have a color pulse. (In another embodiment, the control signal generator of the pulse valve 76 is not locked from generating pulse pulses of the pulse valve during a vertically blanked interval and the time constant of the first AFFC decoder is longer than necessary in the circuits of Fig. 2).
An amplitude detector 77, referred to as a “color pulse presence detector”, detects when a pulse is present in the output signal from the pulse valve 75 for supplying UNITS to cause the multiplexer of the AFC signal 74 to select the output signal from the first AFC detector 73 as the first error signal to be fed to the controlled Oscillator 70 as its AFC signal. Preferably, in terms of noise immunity, the amplitude detector 77 includes a cascade of a synchronous detector, followed by a cascade of a threshold detector, followed by a short pulse eliminator. It is advisable to prepare the setting of the pixel counter 72 to provide a phase shift of 90 o relative to each other, a pair of 3.58 MHz square-wave oscillations for feeding detectors 73 and 77 to the synchronization detection units. Setting up the counters to provide a phase shift of 90 o relative to each other Square-wave oscillation is common when designing television circuits, commonly used in television stereo sound decoders. A short pulse eliminator is known from the radar and is usually constructed using circuits for the logical addition of the input signal that is differentially delayed in it, so an output signal is generated from there.
When a television signal is received without an accompanying color pulse as a black-and-white television signal, the access signal for the AFC of the generator 70 will have to be separated by horizontal clocks supplied to the AFC from the front panel of the television receiver 20. The presence of the color pulse 77 will transmit ZERO when the video signal is complex transmitted from the front panel of the television receiver 20, does not have a concomitant color pulse, creating the conditions for the choice of the AFC signal multiplexer 74 signal the input signal from the second detector AFFK 78 to control the generator 70, as its AFFC signal. The synchronization decoder 79 receives ONE in the count (s) from the pixel counter 72, theoretically, in the presence of a horizontal clock or a part prescribed therein, such as its edge. The output signal from the synchronization decoder 79 is fed as a feedback signal to the second AFFK detector 78, which compares this feedback pulse with the access signal received from the horizontal clock pulses transmitted from the horizontal clock separator on the front of the television receiver 20 and generates a second error signal for selective the multiplexer feeds the AChPK signal 74, into a controlled generator 70, as its AChPK signal. This ACFK circuit is called by television engineers "lockable line synchronization."
The oscillation stability of the controlled generator is required in the number of fields from which the 19th scan lines for accumulation in the temporary line recorder 60 come from, so that the accumulation procedure in which the Bessel chirp is separated from these lines is adequate to suppress the horizontal clock, front spoilage ( pad of the horizontal blanking interval), rear spoilage, including color impulse and base +30 IRE. It is practically necessary to control the crystal of the generation frequency; and automatic phase control - AChPK (ARS) of the aspect of the need for the prevalence of AChPK, with automatic frequency control of the AFC aspect of the too long time constant of the AChPK, i.e. several fields long.
The recovery circuits of the counters 56, 57 and 72 are omitted from FIG. 2 to avoid unnecessary complication. The counter of the scan lines can simply be restored by the leading edges of the vertical clock pulses supplied from the vertical synchronization separator on the front panel of the television receiver 20.
The pixel counter from the pixel counter 72 is restored, when necessary, in order to resynchronize it with the scan lines in the complex video signal supplied from the video detector of the front panel of the television receiver 20. The front and rear edges of the horizontal synchronization pulses supplied from the horizontal synchronization separators of the front panel of the television receiver 20 are detected using a differentiator following the comparator of the assigned level. The leading edge detection result is used to load a temporary memory register register by a pixel counter count. The pixel count is fed to the comparator window to determine if it is in the expected range and to generate an error indication, if not. The counter of the pixel counter 74 is restored to zero provided that the trailing edge of the detector result is supplied. The condition for recovery may be the only indication of pixel count error. However, better noise immunity is obtained by counting errors in a reversible counter configured so that a given number of consecutive errors must be calculated before adjusting the pixel count.
FIG. 3 shows schemes for reconstructing a field counter modulo eight 57, since its counter is either correctly phased or out of phase with four fields, the temporary storage device of line 31, shown as random-access memory, is addressed by the pixel count supplied from the pixel counter 72. Line memory 31 is adapted for read-then-write operations. Logical ZERO issued by decoder 58 only during the 19th scan line of each field is provided to multiplexer 310 for adjusting the temporary storage device of line 31 by digitizing a sample of the 19th scan line supplied from ADC 50. During other scan lines, logical ZERO output decoder 58, causes the data to be output to multiplexer 310 read from the temporary storage device of line 31 for writing back to it.
The temporary storage device of line 31 is provided with a shutter of pixels 32 and 33, timed by the output signal from the intersection detector with zero 71. The shutters of pixels 32 and 33 are used to temporarily store the last pixel recorded in the temporary storage device of line 31 and the last pixel read from the temporary storage devices of the line 31, respectively, the alignment of these samples in time, in accordance with the subtracted and reduced input signals of the digital subtractor 34. Pixel samples of the difference signal From the subtractor 34, all will be zero, except for the corresponding time of the 19th scan line. The difference signal from the subtractor 34 is provided in the absolute value circuit 35, which may include a battery of two-input exclusive-OR gates, each receives a sign bit of the difference signal as the first input and receives a corresponding other bit of the difference signal for selective addition, and each can further include a digital adder for summing the sign bits of the difference signal with selectively supplemented bits of the difference signal to generate the absolute value of the difference as the total output signal th signal.
An accumulator 36 for sequentially sampling the output signal of the absolute value circuit 35 includes a gate 361 for sequentially recorded temporary values of the accumulated result, a digital adder 362 for adding sequential samples of the output signal of the absolute value circuit of the accumulated result in addition to its value, and a multiplexer 363 for selectively supplying the average accumulated result result in the output gate 361 to adjust its contents. Multiplexer 363 serves as a jumper to input arithmetic zero to the output gate 361 as soon as the decoder 58 detects the nineteenth scan line supplied by the counter 56. Decoder 364 receives the pixel count from counter 72, described by these portions of the scan line, which may contain Bessel chirp information to provide ONE, which is logically added to the output signal from the intersection detector with zero 71 in I-gate 365. The output gate 361 is synchronized to the received input information sensitive only to ONE received from I-gate 365.
A sequential sampling of the absolute values of the difference of the nineteenth lines of the current and previous fields, which is supplied by series from the absolute value circuit 36, is accumulated using the accumulator 36. The accumulated result should have a tangible value if the current field is not FIELD 001 or not FIELD 101. 19th FIELD lines 000 and FIELD 001 both contain an ETR signal, therefore, their difference is a zero value, with the exception of noise. The 19th lines of FIELD 100 and FIELD 101 both contain an ETR signal, therefore, their difference is zero, with the exception of noise. The output signal of the threshold decoder 37, which is ONE when the accumulated result is significantly larger than the arithmetic zero and is otherwise NULL, is supplemented by a NOT gate 38 to supply one of the four input signals to the I-gate 39. The decoder 41 detects the field count from the counter 57, other than 001 or 101 for supplying ONE to the I-gate, this unit indicates that the field count is out of phase and counter 57 can be restored. The output of decoder 58, which detects the presence of the 19th line of the field, and the output of decode and 42, which receives the pixel count from the counter 72 to detect the end of the scan line, are the other two inputs of the I-gate 39. Provided that the field count is not 001 or 101, the I-gate 39 produces ONE to restore the counter 57 to the count field 001, and at the end of the 19th line of FIELD 000 or FIELD 100 in the television signal received by the front panel of the television receiver 20. Alternatively, the counter 57 can be restored at 101 or the provision can be made to restore only the two least significant bits of the count and the fields, restoring them to 01.
Returning to FIG. 3, if the field count modulo eight, provided by the field counter 57, is correctly phased, the accumulated result obtained in the temporary storage line 60 during the FIELD 000, the last field in the accumulation cycle, will be the eighth cycle of the ЕТР signal of the Bessel chirp, devoid of the accompanying horizontal pulse timing, front spoilage, rear spoilage, including color impulse and base +30 IRE. On the other hand, if the modulo eight count provided by the field counter 57 is out of phase with four fields, the accumulated result obtained in the line temporary storage device during FIELD 000, the last field in the accumulation cycle, the future eighth cycle of the Bessel chirp signal ETP, devoid of the accompanying horizontal clock, front spoilage, rear spoilage, including color pulse and base +30 IRE. The shift of the four combined binary digits in the direction of decreasing the value divides the accumulated result obtained in the temporary storage device of the line 60 during the FIELD 000 by eight and the resulting quotient is fed as an ETR or ETR signal to the filter coefficient computer 55.
A filter-coefficient computer 55, which is well adapted to correlation, performed against the non-repeatable Bessel chirp function ЕТР or ETR, recorded in its internal register, is programmed to perform a correlation substep that determines whether it receives the input from the line temporary recording device 60 during the FIELD 000 with an ETP signal, an ETR signal, or an unrelated ETR or ETR signal. This procedure allows the filter coefficient computer 55 to determine when the UCP signals are not included in the television signal received by the front panel of the television receiver 20. Computer 55 can then apply the previously determined “bypass” load factor as recorded in the register inside the filters 51, 52 and 53. Alternatively, the computer 55 may be adapted to calculate the load factor of the filters 51, 52 and 53 coming from the relative data of the received repeats supplied by the average, which does not rely on UCP signals included in the television signal received by the front panel of the television receiver 20.
In other embodiments of the circuitry of FIG. 3 circuits external to the computer 55 are designed to analyze the UCP signal recorded in the temporary storage device of line 31 (during the scan line following it acquired, for example) to determine whether it is an ETR or ETR signal and this definition is used to determine whether the majority of the restoration condition signifying bits for the field counter 57 is ZERO, therefore, the restoration falls on the account of field 001, or ONE, therefore, the restoration falls on the account of field 101. The contents of the temporary storage On the device line 31 is scanned in accordance with the count of pixels from the counter 72 during the analysis procedure.
In the example of the analysis procedure for the portion of the pixel count, the corresponding initial fraction of the Bessel chirp is decoded into the selective ONE generation, which is used to allow accumulation in each of the two batteries. One battery requires, in addition, that the sign bit of the current UCP signal should be ZERO in order to accumulate its value (absolute value) in excess of the threshold value T. Another battery further requires that the sign bit of the current UCP signal be ONE, for in order to accumulate its value (absolute value) in excess of the threshold value T. After the portion of the pixel count corresponding to the initial share of the Bessel chirp is scanned, each value of the contents of the batteries is compared in the corresponding omparatorah with a threshold value T that is almost as large as the integral of the absolute value of the initial lobe of the Bessel chirp. If the contents of the batteries, which requires that the sign bit of the current UCP signal, is ZERO, in order to accumulate excesses of these threshold values of T, after the initial fraction of the Bessel chirp, the comparator associated with this battery supplies ONE to the filter coefficient computer 55, this The ONE together with the ZERO of the other comparator sets the presence of the ETP signal. Conversely, if the contents of the battery, which requires that the sign bit of the current UCP signal, is ONE in order to accumulate an excess of this threshold T after the initial fraction of the Bessel chirp, the associated comparator supplies ONE to computer 55, this ONE together with ZERO of the other comparator sets the presence ETR signal. If this threshold T is not exceeded by the contents of any of these batteries, after the initial fraction of Bessel chirp. Two associated comparators, both feed zeros to computer 55, which determines that neither ETR nor ETR signals are present in the television signal, which the devices of FIG. 2 are trying to free from repetition. In further improvement of this scheme, the threshold value of T is regulated in response to noise and amplitude conditions of the OGP signal.
Changing the damping patterns of the repeat of FIG. 2 are possible when data is transferred from the temporary memory of the line 60 to the line memory register in the filter coefficient of the computer 55, the front addressing of the temporary memory of the line 60 and the line memory register is generated in the computer 55, instead of the pixel counter 72. The multiplexer is controlled by the decoder 58 or computer 55 can use the addresses in the temporary recording device of line 60, selecting them from the pixel counter 72 during the 19th line of each field and in another way, ensuring their selection by computer rum 55. Change slaking repeat FIG. 2 is also possible when multiple temporary line storage devices are used, instead of a single line temporary storage device 60, allowing the computer 55 to control filter coefficients 51, 52, and 53 more often than in an eight-field cycle.
Another modification that can be made in the damping repetition schemes of FIG. 2 is an accumulation of temporary storage of a line 60 of 19 scan lines with sixteen consecutive fields, better than eight. This contributes to the correlation of the information of individual Bessel chirps, which significantly improves the signal-to-noise ratio supplied to the filter coefficient computer 55. In such changes of the damping schemes of the repetition of FIG. 2, the field counter modulo eight 57 is replaced by a field counter modulo sixteen, and decoder 65 is replaced by a decoder that produces ONE if and only if the FIELD COUNT from this field counter modulo sixteen is 0000. The temporary storage of one line 60 will then accumulate signals from sixteen consecutive fields, which can be divided into sixteen, using a 4-bit combined shift, to feed the computer 55 corrected low-noise OGP signal with the accompanying repetitions. Further accumulation, i.e. 19 scanning lines with 24 consecutive fields, provides a slightly smaller improvement in the signal-to-noise ratio of the information of an individual Bessel chirp fed to a filter-coefficient computer 55.
In further or still other changes to the damping repetition scheme of FIG. 2, the temporary storage device of a single scan line 60 is replaced by a temporary storage device of two scan lines, and the decoder 58 is replaced by a decoder to determine the presence of the 18th or 19th scan line to supply ONE for two consecutive scan lines, to create conditions for the multiplexer 59 to load a temporary storage device of two scan lines. The decoder 64 is replaced by a decoder to determine the presence of the 261st or 262nd scan lines to supply ONE for two consecutive scan lines to the I-gate 63. The I-gate 63 receives a ONE during the 261st or 262nd scan lines from each field identified by the FIELD COUNTER with all ZERO bits to create conditions for the multiplexer 62 to free the contents of the temporary storage of the two scan lines. Or, the temporary storage device of a single scan line can be replaced by a temporary storage device of three scan lines, and the decoder 58 can be replaced by a decoder to determine the presence of the 18th to 20th scan lines, to create conditions for the multiplexer 59 to load a temporary storage device of three lines scans with suitable software for periodically erasing the memory of the three lines. These devices facilitate a two-wire combination of VBI intervals, including GCP signals in antiphase and infase to other circulation signals, in order to suppress macro-repetitions of long delay.
From a hardware simplification point of view, decoder 64 of FIG. 2, it is preferable to replace with any of the simpler decoders, each of which provides ONE for one or a pair of lines after the 19th line, but before the 262nd line. For example, a decoder that detects ONE in the eight least significant bits of the LINE COUNT can be used with decoder 65 to provide two input signals to I-gate 63. I-gate 63 will then create conditions for multiplexer 62 to clear the contents of temporary storage on line 60 to each of the scan lines after the 255th in the field identified by the FIELD COUNT to zero in each binary digit.
FIG. 4 is a schematic illustration of one method for setting the parameters of the filters 51, 52, and 53; this procedure is performed by a filter coefficient computer 55. This method uses sequential approximation methods to correct the interaction between the characteristics of the filters 51 and 52, when both leading and lagging macro repeats accompany the prevailing image. The procedure for entering START conditions 81 of FIG. 4 occurs during power-up of the television receiver, when a new channel is tuned in, or when the prescribed time elapses from the last release procedure. RESET ALL FILTERS Step 82 preferably sets the coefficients of the filters 51, 52 and 53 to values predetermined for the channel onto which the front panel of the television receiver 20 is switched and recorded in the channel addressing memory. Alternatively, during an increase in power or a reset, the filter coefficients 51, 52, and 53 can be set to a “bypass mode” of values associated with a non-repeat signal, and during periodic deterioration, the previous values of the filter coefficients are maintained during the “restore”.
Then the ACQUIRED DATA step 83 follows, this step 83 ends after the number of fields that the computer 55 has to wait for the accumulation by the temporary storage device of the line 60 to end, in order to generate a separate UCP signal that corresponds to the input data for the computer 55 Step ACQUIRED DATA 83 includes substep correlation not shown in FIG. 4, this substep determines whether the input of computer 55 receives from the temporary storage device of line 60 during the FIELD 000 ETR signal, an ETR signal, or a signal not connected to either the ETR or the ETR.
Then comes step 84 CHANNEL CHARACTERISTIC. Computer 55 does this by correlation, in the time domain; An OGP signal that is free from repetition is recorded in its permanent memory with an OGP signal with a repeat repeated from the received complex video signal. The time position of the prevailing characteristic of the data transmitted by the computer 55 is detected when the corresponding time distribution of each successively less than one significantly larger repeat characteristic increases the number of post-repeats that can be canceled by the filter 51 and increases the number of pre-repeats that can be suppressed by the filter 52. The corresponding time location of the dominant characteristic and the multipath characteristics of the data transmitted to the computer 55, calculated and temporarily The internal memory of computer 55 is used as a basis for programming the spread of load delay lines between the groups of IIR filter taps 51. The relative power of the dominant characteristic and multipath characteristics in the data transmitted to computer 55 are calculated by computer 55 and temporarily stored in its internal memory to to be used as a basis for assigning loads to the IIR filter group taps 51 and the FIR filter taps 52.
FIG. 7 shows a set of steps that a filter coefficient computer 55 may be programmed to take in order to perform step 84 CHANNEL CHARACTERISTICS. Immediately after condition 840, the START STEP, in the initial substep 841, the discrete Fourier transform (DFT) of the acquired OGP signal is calculated. Then, in substep 842, computer 55 divides the members of this DFT into the corresponding members of the DFT of a repeatless UCP signal, this last DFT is known in advance and recorded in the internal memory of computer 55. This dividing-by-member division of substep 842 produces a DFT of the receive channel, which temporarily stored in the internal memory of the computer 55.
As part of step 84 CHANNEL CHARACTERISTIC, it is preferable to normalize the DFT terms of the receive channel according to the energy in the prevailing image. In substep 843, computer 55, in accordance with its program, selects the DFT term of the receive channel of the largest magnitude. Then, in sequential substep 844, the mean square energy of this term and its nearest members (that is, twelve on each side) is determined and temporarily stored in the internal memory of the computer 55. Normalization can be done on the prevailing image and other repeated images, but preferably with In terms of reducing computations, normalize only on these terms, called “strong” repeats or repeats with significant energy, and replace the remaining terms with zeros.
The definition of terms describing strong repetitions follows in substep 845. The root-mean-square energy of the DFT term of the receiving channel of the largest magnitude and its closest members, which terms describe the predominant image, are understood to provide a threshold level at which to compare other terms of the DFT of the receive channel. A threshold level of -30 dB below the root mean square energy of the prevailing image was found to be satisfactory. Each DFT member of the receive channel that exceeds this threshold level is taken into account to be described in a strong repeated way and compared with its neighboring members to determine the largest local term and the location in time of the center of strong repetition. The largest local member and its closest members, as the described image of strong repetition, are marked in the internal memory of computer 55. Storage of DFT members in computer 55 can be done in the corresponding memory cells, which are addressed by a temporary bin (discretization element), and then marked by setting the prescribed value of the label bit associated with the bits describing the size of the member.
Then, in substep 486, which discards weak duplicate DFT images of the receive channel to generate an approximate DFT of the receive channel, computer 55 replaces with zeros the values of all unlabeled members of the DFT receive channel temporarily recorded in its internal memory.
In the normalized substep 487, the computer 55 divides each of the marked terms in the approximated DFT by the mean square energy of the prevailing image to generate a normalized approximated DFT of the receiving channel. This normalized approximated DFT of the receive channel, temporarily recorded in the internal memory of computer 55, was used to suppress the remaining portion of the calculations before condition 848 FINAL STEP is finally reached in step 84. The marks are stored on the members of the normalized approximated DFT of the receive channel temporarily recorded in the internal the memory of computer 55, therefore, tags can be used to perform a later step of decision 88 in the procedure of FIG. 4.
Solution Step 85 STABLE REPEAT? follows step 84 CHANNEL CHARACTERISTIC. This step is advanced in computer 55 using a subroutine where the results of step 84 CHANNEL CHARACTERISTICS only preceded the more recent results of step 84 CHANNEL CHARACTERISTICS obtained from the internal memory register of computer 55 and are replaced in this register by the results of the current step 84 CHANNEL CHARACTERISTIC. Computer 55 cross-correlates the results of the later come step 84 CHANNEL CHARACTERISTIC with the results of only the previous step 84 CHANNEL CHARACTERISTIC to determine whether the correlation is good enough so that the state of the repeat can be considered stable or unchanged. Only if the repeat states are practically unchanged, a signal Y (yes) is generated, indicating the presence of a basis for going further with the detailing procedure, using the later results of CHANNEL CHARACTERISTICS. If the solution step is STABLE REPEAT? 85 produces a signal N (no) indicating an alternate state of repetition, the actions of computer 55 return to step 83 ACQUISED DATA, and the adjustable filtering parameters of the IIR filter 51 and the FIR filter 52 remain unchanged. If the solution step is 85 STABLE REPEAT? generates a Y signal (yes), the procedure goes to steps 86-88, which use the newer results from step 84 CHANNEL CHARACTERISTICS, as a basis for adjusting the adjustable filtering parameters of the IIR filter 51 and the FIR filter 52.
In step 86, ADJUSTING the IIR filter coefficients, programmable delays and nonzero load factors of the IIR filter are adjusted by computer 55 using the most recent results from step 84 CHANNEL CHARACTERISTICS as the basis for the adjustment. In more detail, part of the post-repetitions of the most recent normalized approximated DFTs of the receiving channel, these labeled terms, later in time than the largest-magnitude term, complements the generation of the DFT of the required IIR filter characteristic 51, the corrected filter coefficients 51 are taken from this DFT DFT members of the required IIR filter characteristic 51 are used to determine load factors. If the IIR filter 51 is of an exploded characteristic type, computer 55 acts to measure the length of the intervals containing only unlabeled zero coefficients to determine the delay values for the adjustable load delay devices. Computer 55 supplies the adjusted IIR filter parameters to the IIR filter 51.
Step 87 CORRECTING FIR-FILTER FACTORS is performed by computer 55 after completing Step 86 of CORRECTING FIR-FILTER FACTORS. Nonzero load coefficients of the FIR filter 52 are adjusted by computer 55 using the most recent results from step 84 CHANNEL CHARACTERISTICS as the basis for the correction. In more detail, part of the pre-repetitions of the most recent results of the normalized DFT of the receiving channel, earlier in time than the term of the largest magnitude, serves as an addition to the generation of the DFT of the required characteristic of the FIR filter 52, the corrected coefficients of the FIR filter are taken from this DFT. If the FIR filter 52 is an exploded characteristic type, computer 55 acts to measure the length of the intervals containing all the zero coefficients to determine the adjustable delay of the load delay devices. Computer 55 feeds the adjusted FIR filter coefficients to the FIR filter 52.
FIG. 4 shows the solution step 88 REPEATS BELOW THE THRESHOLD?, Which is transmitted by computer 55 after performing steps 86 and 87, ADJUSTING FIR-FILTER COEFFICIENTS to CORRECTING FIR-FILTER COEFFICIENTS, respectively. Step 88 can be a tool to continue from the count of the number of tags attached to the members of the normalized approximated DFT of the receive channel in step 84 CHANNEL CHARACTERISTICS, the score is fifteen or lower (i.e., the score was not significantly larger than the number of tags associated with the predominant image) a Y (yes) signal is generated, and if the count was sixteen or more, an N (no) signal is generated. Alternatively, step 88 may be a tool to continue from the count of the number of unlabeled members, since the number of labeled members and unlabeled members is added to the prescribed total number of members.
Signal N (no), coming from step 68 REPEATS BELOW THE THRESHOLD?, Directs the computer 55 to step 89 MAXIMUM NUMBER OF ITERATIONS? The iteration counter in computer 55 counts the number of consecutive N (no) signals coming from the decision step 88 REPEATS BELOW THE THRESHOLD ?, and restores the count to zero with the signal Y (yes) coming from step 88 REPEATS BELOW THE THRESHOLD? Signal N (no), coming from step 88 REPEATS BELOW THRESHOLD?, Is transmitted by this counter before the maximum count, returning back to step 83 DATA ACQUISITION.
If the solution in step 88 is Y (yes), all meaningful repetitions are canceled, or if the solution in step 89 is Y (yes) it indicates that enough iterations have been done to ensure that filters 51 and 52 are not able to be further adjusted to extinguish at least one more repetition, the part of the procedure for extinguishing macro repeats is completed in computer 55 and computer 55 proceeds to step 90 ADJUSTMENT OF STABILIZATION COEFFICIENTS, in which load coefficients for an amplitude-stabilized filter 53 are calculated. Iteration counter either is restored to zero by the signal Y (yes) coming from one of the decision steps 88 and 89, or it is tipped to zero.
FIG. 8 shows a routine that computer 55 can use to perform step 90 ADJUSTING STABILIZATION FACTORS. Passing from condition 900 STARTING STEP, computer 55 in substep 901 responds with a calculation to its program, depending on the characteristics of the macro-quenching filter, including cascade-connected filters 51 and 52, DFT of only a fraction of the Bessel chirp of the UCP signal with a repeat that is separated from the rest of this signal by the window overlay procedure. This DFT has nonzero terms only near its largest member, and in substep 902 it divides the term by term into the DFT of the ideal characteristic in the repetition-free chirp that is copied from the computer’s permanent memory 55. This generates a DFT of the stabilizing filter 53. which is the basis for computing computer 55 in substep 903, which corrects the load coefficients of the retraction of the FIR filter 53 used for the counter counter effects of micro-repeats. The width of the bin of the DFT members can be the same as the width of the bin of the DFT members used in calculating the adjustable filter parameters of the filters 51 and 52 used to quench the macro repeats. The number of taps for the FIR filter 53 is usually not more than 31, nevertheless, this number of spectral bins in the truncated DFT is justified little and eliminates the unreasonable stretching of the time required for these calculations. The division procedure in these stabilization methods is prone to error when the OGP signal is noisy or when division into small numbers occurs, as is sometimes required to limit the upper quotient region. In the final substep 904, immediately preceding the FINAL STEP condition 905, it is achieved that the corrected load factors calculated by the computer 55 in substep 903 are supplied to the coefficient registers of the FIR filter 53.
Following step 90, CORRECTION OF STABILIZATION FACTORS in the procedure of FIG. 4, the computer 55 proceeds to another DATA ACQUISITION, step 91. Step 91 ends after a certain number of fields have passed that the computer 55 must wait until the accumulation in the temporary storage device of line 60 is completed in order to generate a dedicated UCP signal, which is a suitable input for computer 55. Step 91 ACQUIRED DATA includes a correlation substep not shown in FIG. 4, in this substep, the computer 55 determines what input it receives from the temporary storage device of line 60 during the FIELD 000 — the ETR signal, the ETR signal, or a signal that is not connected to either the ETR or the ETR.
Then computer 55 performs another CHANNEL CHARACTERISTIC, step 92, using the ETR or ETR signal obtained in step 91 to recalculate the DFT of the receive channel. In the decision step 93 REPEATS THE SAME? computer 55 determines whether the DFT of the receive channel, recalculated in step 92, CHANNEL CHARACTERISTIC, is correlated with the DFT of the receive channel, calculated in advance in step 84 CHARACTERISTIC OF THE CHANNEL. From the point of view of ease of implementation, it is preferable to do the correlation indirectly, by checking by eye whether all the residual repeats in step 84 CHANNEL CHARACTERISTIC and in step 92 CHANNEL CHARACTERISTIC are below the prescribed threshold level. If the correlation is good, as indicated by the fact that the repetitions have not undergone any noticeable changes, the decision step 93 generates a signal Y (yes), which returns to step 91 ACQUISITION OF DATA to continue the check to detect whether or not a noticeable change in the repetitions occurs. Computer 55 does not change the filtering parameters of filters 51-53.
If the correlation is poor, as indicated by the change in repetitions, the solution step 93 generates a signal N (no), so that the computer 55 returns to step 82 RESTORE ALL FILTERS TO REMOVE FROM REPEAT. This procedure allows you to damp out repetitions when there is a rapid change in multipath conditions or when various receive channels are selected. The filtering parameters of the filters 51-53 are then recalculated by the computer 55, following the previously described steps.
In the procedure of FIG. 4, the computer 55 performs step 86 ADJUSTING FIR-FILTER COEFFICIENTS and step 87 ADJUSTING FIR-FILTER COEFFICIENTS, independently of each other during each passage through these two consecutive steps. The correction of filter coefficients in the initial unit of cascade-coupled quenching filters, here it is a post-repeat filter 51, gives rise to "generated by the filter" repeats of the type that can be suppressed by adjusting the filter coefficients of the last unit, these filters, here it is a pre-repeat filter 52. Since step 86 ADJUSTING FIRST FILTER COEFFICIENTS and step 87 CORRECTING FIRST FILTER COEFFICIENTS do not take into account filter-generated repetitions, sequential recalculation of load coefficients s cascaded filters 51 and 52 during the next passage through these two successive steps will introduce compensatory repetitions, which will reduce the "filter- generated" ghosts in the final filter response. Since this reduction cannot be complete, measures must be taken to recalculate the load coefficients of the last unit of cascade-coupled filter-blanking filters. Computer 55 loops around steps 83-89.
FIG. 5 shows a flowchart of an alternative method for setting operating parameters of filters 51, 52, and 53, this procedure is performed by filter coefficient computer 55.
This method is similar to the method of FIG. 4, but without a step 90 CORRECTION OF STABILIZATION FACTORS after the decision step 88 REPEAT SAME? through step 94 CORRECTION of the SYNCHRONIZATION COEFFICIENTS before the decision step 88. Step 94 CORRECTION of the STABILIZATION COEFFICIENTS is included in the iteration loop so that the load factor of the stabilization filters can be adjusted slowly, over the course of many steps of obtaining the GCP signal data. This reduces the sensitivity of the load factors of the stabilization filter to the noise associated with the UCP signal. The OGP signal is coherent over many steps of obtaining the data of the OGP signal to sequentially adjust the load factors of the stabilization filter to the required values. The noise accompanying the UCP signal is incoherent, i.e., erratic during many steps of obtaining UCP signal data; therefore, its effect on the values of the load coefficients of the stabilization filter are averaged to zero on the basis of the lowest error squared value.
FIG. 12 shows the routine used by computer 55 in step 94 ADJUSTING STABILIZATION FACTORS. The load coefficients of the filter 53, adjusted by the computer 55 in step 94 so that the cascade connection of the filters 51-53, accumulated in the temporary storage device of the line 60, in the best way matches the ideal characteristic of the OTP Bessel-free chirp, which is recorded in the computer 55. Satisfying condition 904 STARTING STEP, computer 55 in substep 941 responds with a calculation to its program, depending on the characteristics of the macro-quenching filter, including cascade-connected fil Sections 51 and 52, the DFT of only a fraction of the Bessel chirp of the UCP signal with a repeat that is extracted from the rest of this signal by the window blending procedure, after this initial substep 941, the routine proceeds to substep 942. In substep 905, computer 55 responds to its program by generating cross-correlation DFT of only a fraction of the Bessel chirp of the UCP signal with repetition of the DFT of the ideal characteristic of the Bessel chirp, which has been superimposed by the 58th window, and which is copied from the computer's permanent memory 55. In order to generate a crossover After a closely correlated result, computer 55 examines the members of one of the DFTs in the reverse time sequence and sequentially uses them to multiply by the members of the other DFT. (See pages 69-71 and 471 PROCESSING DIGITAL SIGNALS IN VLCI by Richard J. Higgins, Prentice Hall, Englewood Cliffs, New Jersey). Cross-correlation is performed on a cyclical basis, where each DFT revolves around itself in a time interval. The result of this correlation procedure determines the time basis to which the central tap of the synchronization filter 53 will refer and with respect to which the relative delays of the other taps of the filter 53 will be determined.
In a subsequent substep 943, computer 55 calculates the changes in the “middle” terms of the cross-correlation characteristic required to bring them into line with the results of the cross-correlation of the non-repeating Bessel chirp of an undetermined band that has been superimposed by windows with the ideal characteristic of a non-repetitive Bessel chirp that has been superimposed by the windows. The ideal characteristic of a free-from-repeat Bessel chirp that has been overlaid with windows, used in this routine and routine of FIG. 8 may have (sin X) / X convolution in the time domain, called the low-pass step characteristic in the frequency domain. However, a video image in which some high-frequency attenuation is observed for a short-term performance improvement is usually subjectively more enjoyable; therefore, the ideal characteristic of the Bessel's repeat-free UCP chirp recorded in the memory of computer 55 for use in stabilization can preferably have a suitable high-frequency attenuation.
In the subsequent substep 944, which ends with condition 945 FINAL STEP, computer 55 increases (or decreases) each of the loads in the stabilization filter by crushing the required changes to bring it into line with the results of cross-correlation of the undetermined Bessel chirp repeated by windows with an ideal characteristic of a free-from-repeat Bessel chirp that has been overlaid with windows. This procedure, together with an iteration loop around steps 88, 89, 83-87 and 94 in the procedure performed by computer 55, slowly adjusts the load factors in the stabilization filter 53. This makes it less likely that the pulsed noise associated with the OGP signal will lead to an erroneous calculation load factors of the stabilization filter 53.
FIG. 6 shows a flowchart of another alternative method that the filter coefficient computer 55 may use to set operation parameters of filters 51, 52, and 53. In the method of FIG. 6, the computer 55 calculates, without sequential approximation, a correction for the interaction between the responses of the filters 51 and 52, when both the leading and lagging macro repeats accompany
prevailing image. The entry into condition 81 START of the procedure of FIG. 6 is the same as for the procedure of FIG. 4, and step 82 RESTORE ALL DEPOSITOR FILTERS, step 83 DATA RECEIVED, step 84 CHANNEL CHARACTERISTICS OF THE SOLUTION 85 STABLE REPEATS and the return loop from step 85 to step 83 are the same as in the procedure of FIG. 4, and in the procedure of FIG. 6.
When is the solution step 85 STABLE REPEATS? produces an output signal Y (yes), the procedure of FIG. 6 proceeds to step 95 CALCULATION OF THE FIRST COEFFICIENTS and to step 97 CALCULATION OF THE FIRST COEFFICIENTS. The calculation of the adjustable filtering parameters for the filters 51 and 52 is performed by the computer 55 in steps 96 and 97 of the procedure of FIG. 6, these steps can be performed in any order, in accordance with the calculations performed in step 86 CORRECTING THE FIRST COEFFICIENTS and step 87 CORRECTING THE FIRST COEFFICIENTS of the procedure of FIG. 4. However, in the procedure of FIG. 6, the computer 55 delays the application of the adjustable filtering parameters of the filters 51 and 52 until the adjustment of these filtering parameters is complete. Step 98 CALCULATION OF AMENDMENTS FOR “REPEATED GENERATED FILTERS”, which will be described in more detail below, follows steps 96 and 97 of the procedure of FIG. 6, in this step 98, the computer 55 performs the adjustment, which should achieve compliance with the filtering parameters of the filters 51 and 52, previously calculated in steps 86 and 87. Then, in step 99, ADJUSTING ALL DEPOSITOR FILTERS, the computer 55 supplies the adjustable filtering parameters to the filters 51 and 52. In the procedure of FIG. 6, the computer 55 then proceeds to step 90 ADJUSTING THE STABILIZATION FACTORS. This transition occurs invariably, since the computer 55 does not need to provide a return to step 83 RECEIVED DATA, as it does in the procedure of FIG. 4.
Step 90 ADJUSTING THE STABILIZATION FACTORS, step 91 DATA RECEIVED, step 92 ADDITIONAL CHANNEL CHARACTERISTICS, solution step 93 REPEAT SAME ?, returning from step 93 to step 91 and returning from step 93 to step 82 are the same . 4, and in the procedure of FIG. 6.
In the procedure of FIG. 4 and FIG. 6, the stabilizing filter 53 makes adjustments for the micro-repeats obtained by correcting the micro-repeats, in addition to providing adjustments for the received micro-repeats associated with the complex video signal supplied from the video detector, as described above. These micro-repeats take place due to the fact that the suppression of macro-repeats is not performed on a video signal considered as a complex signal, but (in accordance with the previous description) is performed on a video signal considered only as a valid signal and because of this, the number of taps in each segment grouped taps of a quenching filter with distributed taps is limited to only ten or so. The use of a stabilization filter 53 for correcting micro-repeats generated during the quenching of macro-repeats is a concept adopted for a configuration where the quenching of macro-repeats occurs on a filter that is configured differently from the cascade connection of the IIR filter 51 and the FIR filter 52, observation of the inventors.
The method of FIG. 6 described above can be adapted for use when blanking of macro repeats in complex video signals supplied from a video detector of a television receiver or VCR is performed using only an IIR filter. In such an adaptation, step 99 follows immediately after step 96, and steps 97 and 98 are omitted. There is no possibility to cancel pre-repetitions, but pre-repetitive conditions, which are relatively rare when compared with post-repetitive conditions, are relatively rare, usually occur in urban overpopulation during non-air reception using an omnidirectional or non-directional antenna.
The method of FIG. 6 described above can also be adapted for use when the blanking of macro repeats in a complex video signal supplied from a video detector of a television receiver or VCR is performed only using an FIR filter. In this adaptation, step 96 is omitted, step 99 follows immediately after step 97, and step 98 is omitted. Quenching macro repeats using only the FIR filter avoids the problem of quenching repeats, lasting indefinitely, at constantly decreasing levels, which can sometimes cause subtle depot artifacts. The group delay problems inherent in the IIR filter design can be avoided.
FIG. 9 shows one form in which the divided loads of the IIR filter 51 can be applied. In this form of FIG. 9 The IIR filter 51 is a filter with a loaded output, with loads applied to differentially separated signals. The input signal to the IIR filter 51 is supplied as one of the input signals of the multi-input digital adder 510, which generates the output signal of the IIR filter 51. The output signal of the IIR filter 51 is supplied to the delay line used as the initial component of the cascade from the load delay device 511, sections FIR filter 512, load delay device 513, FIR filter sections 514, load delay device 515, and FIR filter section 516, load delay device 517, and FIR filter section 518. The output signals from the corresponding summed-and-loaded to and FIR filter sections with a loaded output 512, 514, 516 and 518 are linearly connected to each other (and output signals from any other FIR filter sections in the cascade after them) and to the input signal of the filter 51 of the adder 510, to generate an output IIR filter signal 51. Each of the sections of the FIR filter 512, 514, 516, and 518 can dampen the corresponding post-repeat shape of this output signal.
FIG. 10 shows one form in which the divided loads of the FIR filter 52 can be applied. In this form, FIG. 10 FIR filter 52 is a filter with a loaded output, with loads applied to differentially separated signals. The input signal to the FIR filter 52 is fed to the delay line, which has as its final component a load delay device 529, a FIR filter section 528, a load delay device 527, a FIR filter section 526, a load delay device 525, a FIR filter section 524 , load delay device 523, FIR filter section 522, load delay device 521. The output signals from the corresponding summed-and-loaded fraction of the FIR filter section with the loaded output 522, 524, 526 and 528 are linearly connected to each other (and output signals from any other FIR filters, toyaschih in cascade after them) and with the output signals of delay unit load 523, thereby generating an output signal of the FIR filter 52. Each of the FIR filter sections 522, 524, 526, 528 can cancel a respective post-ghost form this output signal.
The respective delays of each of the load delay devices 511, 513, 515, 517, 519, 521, 523, 525, 527 and 529 are programmed depending on the digital signal received in this temporary storage device register. These sections of the FIR filter 522, 524, 526 and 528, each is a multi-tap delay line having a sum-and-load circuit for applying output loads to their tap signals, each of these loads is programmed depending on the digital signal received in the corresponding register temporal storage device of this section of the FIR filter.
Anyone versed in the design of digital filters can easily convert load-loaded designs of FIG. 9 and 10 into a mold with a loaded input using well-known development procedures, but molds with loaded outputs are often preferred for filters with adjustable parameters due to the fact that the filtering functions can be adjusted immediately without cleaning old samples. In filters with spaced loads, measurements in load delays are explained by the need to wait until the old samples are cleaned so as not to introduce artifacts into the image, and these changes are best made during the vertical back blanking interval. Changing the filter characteristics by adjusting the loads of the tap can be done almost instantly, but it is best done during the horizontal horizontal blanking interval.
The procedure of FIG. 4 can be modified in such a way that computer 55 performs step 86 CORRECTING THE FIRST COEFFICIENTS, after step 87 CORRECTING THE FIRST COEFFICIENTS, which is better than vice versa. The procedure of FIG. 6 can be modified so that computer 55 performs step 96 CALCULATING FIR FACTORS after step 97 CALCULATING FIR FACTORS, which is better than vice versa.
According to the block diagram of FIG. 11 the procedures included in steps 96 and 97 of the procedure of FIG. 6 will now be discussed in more detail. Detailing steps 96 and 97 of the procedure of FIG. 6 is also appropriate for a more complete understanding of steps 86 and 87 of FIG. 4. In this more detailed description of the filters 51 and 52, it is assumed that they have the structure of the examples of FIG. 9 and 10, respectively.
Initial Step 95 DETERMINATION OF THE MIDDLE MEMBER OF THE PREVALING IMAGE in the block diagram of FIG. 11 begins with computer 55, when in the solution step 85 STABLE REPEAT? output signal Y (yes). In step 95, the group of members with very high energy of the normalized approximated DFT of the receive channel obtained in step 84 and FIR filter 52 used to quench repetitions. This group of members with the highest energy was determined in step 84 CHANNEL CHARACTERISTICS and the results of this determination can be transferred to the internal memory of computer 55 for use in steps 96 and 97, or alternatively, computer 56 can be programmed to repeat these calculations at the beginning of the series of steps 96 and 97. Members of the normalized approximated DFT of the receive channel, which are later in time than the group of members having the highest energy, are considered as a post-repetitive region, thus, it should be used as a basis for calculating the adjustable filter parameters of the IIR filter 51. The members of the normalized approximated DFT of the receive channel, which are earlier in time than the group of members with the highest energy, are considered as a pre-repeated region and, thus , should be used as a basis for calculating the adjustable filter parameters of the FIR filter 55. The boundary between the post-repeat region and the pre-repeat region is more accurately determined within the group of members s, with the greatest energy, the computer 55 is programmed to select the greatest of these members (or "prefer" one of them when there are many members of the largest) as the middle term of the temporal spectrum of the predominant image.
Assume that step 96 of calculating the FIR coefficients precedes step 97 of calculating the FIR coefficients in the procedure of FIG. 6, steps 961 and 962, which are substeps of step 96, are executed by computer 55 after step 95. Then, steps 551 and 972 are executed by computer 55, which are substeps of step 97.
In step 961, DETERMINATION OF LOAD DELAYS TO PAY OUT REPEATS The delays provided by load delay devices in the 51 IIR filter with spaced load are calculated by computer 55. In step 84, CHANNEL CHARACTERISTICS, the largest local members and their closest members, as being descriptors of the prevailing image and strong repeating images are marked in the internal memory of the computer 55. The number of terms in the approximated DFT for the receive channel, which are later in time than the largest term, and which I describe t the prevailing image, or are in a subsequent series of unlabeled members of the zero value, calculated by computer 55, which temporarily stores the count results in the internal memory and uses scores to program the delay of the load delay device 511. The number of members in each series of unlabeled members of the zero value, consecutively late in time is calculated by computer 55, which temporarily stores the corresponding sequence of accounts in its internal memory, and uses accounts to program delays in load delay bores 513, 515, 517, ... respectively, step 961 generates indexed information for placement in the FIR sections of the filter 512, 514, 516 and 518 at suitable delay intervals for the subsequent step 962.
At step 962, DETERMINATION OF LOADS OF TAPES FOR PAYMENT OF POST REPEATS, computer 55 considers each successive group of marked members in revolution and calculates the load of the tap for the corresponding section of the FIR filter with the loaded output 512, 514, 516, 518, ... This subroutine is similar to a subroutine FIG. 8 or the routine of FIG. 9 to calculate the loads of the taps of the stabilization filter 53, except that the DFT of the chirp with the repeat of the Bessel DFT and the non-repeat Bessel chirp drawn from the internal memory of the computer 55 both overlap the time interval of a full scan line or two, and are indexed in time vis-a -vis each other. Indexing is a summation of the time intervals covered by load delays preceding one of the sections of the FIR filter 512, 514, 516, 518, ..., for which the loads of the taps, delays through the multi-tap delay lines of each previous section of the FIR filter with the loaded output were calculated and half the delay of the multi-tap delay line of the FIR filter section for which the loads of the taps were calculated.
In step 971, DETERMINATION OF DELAY LOADS FOR EXTINGUISHING PRE-REPEATS is the number of terms in the approximated DFT for the receive channel that are earlier in time than the largest term and which either describe the prevailing image or are in the nearest earlier series of unlabeled zero-value terms is calculated by computer 55, which temporarily stores the count results in its internal memory and uses the count to program the delay of the load delay device 521. The number of members in each series of unlabeled members a value of zero, successively earliest in time, is calculated by a computer 55, which temporarily stores the corresponding sequence of accounts in its internal memory and uses accounts to program the delay in load delay devices 523, 525, 527, ... respectively. The loading of the initial load delay (529 in FIG. 10) from the delay line of the FIR filter 52 is calculated by subtracting the delay of all subsequent elements in this delay line from the delay value associated with the earliest pre-repeats whose blanking is always possible: this maintains a constant delay, which is provided in the prevailing image, therefore, these changes in the suppression of pre-repetitions do not introduce any jitter from side to side in this image. Step 971 generates indexed information to accommodate the FIR filter section 522, 524, 526, and 528 inappropriate delay intervals for the subsequent step 972.
In step 972, DETERMINING LOADS OF TAPES FOR PAYING OFF PRE-REPEATS, computer 55 considers each sequentially earlier group of marked members into rotation and calculates the load of the tap for the load circuit and summing the corresponding FIR filter with the loaded output 522, 524, 526, 528, .. This subroutine is similar to the subroutine of FIG. 8 and the routine of FIG. 9 for calculating the loads of the taps of the stabilization filter 53, in addition to the fact that the DFT of the Bessel repeater and the DFT of the Bessel-free repeater drawn from the internal memory of the computer 55 both overlap the time interval of a full scan line or two, and are indexed in time vis-a -vis each other. Indexing is a summation of the time intervals covered by the load delays following one of the sections of the FIR filter 522, 524, 526, 528, for which the load of the taps, the delays through the multi-tap delay lines of each subsequent section of the FIR filter with the loaded output and half the delay multi-tap delay line of the FIR filter section, for which the load of the taps was calculated. After completing step 971, for steps 96 and 97, condition 950 is reached. FINAL STEP.
Steps 95, 961, 962, 971 and 972 that have been described so far, together with the application of the calculated filter parameters of the IIR filter 51 and the FIR filter 52, are essentially similar to steps 86 and 87 of the method of FIG. 4. In steps 95, 961, 962, 971, and 972, computer 55 calculates the filtering parameters for the IIR filter 51 and the FIR filter on separable bases. This produces a general correction of the blanking characteristic for cascades of filters 51 and 52, only if they are not macro repeats, if there are only lagging macro repeats and there are no leading macro repeats, or if there are only leading macro repeats and there are no lagging macro repeats.
Take the cascade connection of the IIR filter 51, the FIR filter 52, and the stabilization filter shown in FIG. 2. If pre-repeats are present as well as post-repeats, independent calculation of the IIR filter 51 parameters will make them such that it will delay the prevailing image and will combine it with each post-repeat destructively. All post-repeats are canceled without generating repetitions, IIR filter 51 delays each pre-repeat, similar to the prevailing image, but each delayed pre-repeat does not usually connect destructively with anything and gives rise to a “filter-generated” repeat. Each pre-repeat gives rise to one “filter-generated” repeat for each canceled post-repeat, ignoring the repeats. As an example, suppose that relative to the prevailing image, there are two pre-repeats at -10 mS and -3 mS, respectively, and that there are three post-repeats at 4 ms, 8 ms and 20 ms, respectively. Each pre-repeat gives rise to three “generated filter” repeats delayed from there by the delay component of the post-repeat, respectively, of the prevailing image, and attenuates in accordance with the size of the post-repeat, respectively, of the dominant image. At the output of the IIR filter 51, an initial pre-repetition of -10 mS gives rise to a “generated filter” repetition located in time at -6 mS, -2 mS and +10 mS relative to the prevailing image; and a pre-repetition of -3 mS gives rise to "generated by the filter" repeats located in time at +1 mS, +5 mS and +17 mS relative to the prevailing image. The magnitude of any of these "filter-generated" repeats is the product of the pre-repeat and post-repeat values that gave rise to this "filter-generated" repeat. This process of “internal” repetition generation is repeated with gradually decreasing amplitude. For perfect repetition of repetitions, these repeated repetitions must also be canceled or reduced below the level of subjective perception.
If the structure of the snooze filter includes an IIR filter 52 next to the FIR filter 51, then the post-snoozes give rise to the "generated filter" snoozes at the output of the FIR filter 51. The location of the "filter-generated" snaps moves forward relative to the post of the repetitions giving them a start, the component of advancement depends on the time component of the pre-repetitions facing the prevailing image.
Since the location and magnitude of the initial internally generated repeats can be previously calculated, computer 55 can be programmed to calculate the coefficients for a filter that extinguishes repeats, which will "generate" repeats of the "opposite" magnitude at the intended locations. Thus, when a filter that quenches repetitions gives rise to a “filter-generated” repetition, it achieves a quench due to the “existence” of a repetition of opposite polarity in the same place. At the same time, the repeated generation of repeats is also prevented. Step 98 CALCULATION OF AMENDMENTS FOR "FILTER-GENERATED" repeats in the method of FIG. 6 makes corrections for the interaction of filters 51 and 52, which would otherwise give rise to a “generated filter” by repetition, when the leading and lagging macro repeats accompany the predominant image.
A more detailed block diagram of FIG. 13 shows the substeps of step 98 of FIG. 6 performed by computer 55 after steps 95-972. START condition 980 is entered when the last of steps 96 or 97 ends or is "completed." The initial decision step 981 AVAILABILITY OF PRE-REPEATS AND POST-REPEATS is performed by computer 55 and as a result, signal N (no) immediately sends the procedure to condition 988 FINAL STEP, followed by step 99 ADJUSTING ALL FILTERS FROM REPEATING the main program of FIG. 6.
As a result of the Y signal (yes), the computer will continue the subroutine from substep 982 CALCULATING DFTs OF FOLDED FIR- AND FIR-FILTERS EXTINGUISHING THE REPEATS. In substep 962, computer 55 is programmed to multiply the magnitude of each of the DFT terms of the characteristics of one of the filters 51 or 52 by the value of each of the DFT terms of the characteristics of another filter, respectively, the product determines the values of the corresponding terms of the DFT of minimized characteristics, i.e., the DFT of the cascade connection of the filters 51 and 52. In substep 982, computer 55 is further programmed to linearly combine the advance of pre-retries and the delay of post-retries, giving rise to each DFT member of collapsed characteristics, thus determining the position of the member in time.
Then, the computer performs a substep 983 DEFINITION OF STRONG "FILTER-GENERATED" REPEATS AND CHANNEL REPARCHING. In a procedure like this described previously, when considering sub-step 845 of the routine of FIG. 7, it was determined whether the filter-generated repeat has enough energy to be of interest. The “filter-generated” repeat defined in substep 983 must have enough energy to be of interest, this repeat is present during the time interval, otherwise occupied by unlabeled zero terms in the channel response results and in the listing of the corresponding filter parameters, replaces these unlabeled zero terms, member per member, to the labeled basis of the "re-characterized" channel.
The substeps 984, 985, 986, and 987 that follow the substep 983 are performed by the computer 55 in virtually the same way as the substeps 961, 962, 963, and 984, respectively, but the results from the “over-characterized” channel response are better than the initial channel characteristics. At the end of substeps 984, 985, 985, and 987, computer 55 reaches condition 988 FINAL STEP followed by step 99 ADJUSTING ALL FILTERS FROM REPEATING the main program of FIG. 6.
When the OGP signal is being received, there is also a small probability of significant concomitant noise, step 90 ADJUSTING THE STABILIZATION FACTORS in the method of FIG. 4 or in the method of FIG. 6 should be performed in the following alternative manner. Computer 55 executes a routine similar to the routine of FIG. 12 used in step 94 CORRECTING STABILIZATION FACTORS, in addition, substep 944 is modified to increase (or decrease) the taps loads for the stabilization filter 53, not an adjustment that splits the calculated changes in cross-correlation, but a better adjustment that evens out the calculated changes in cross correlation. This one-count control procedure does not recognize noise associated with the UCP signal.
The damping methods described here apply to the UCP signal differently from the other methods described, although suitable modifications may be necessary with respect to the circuits used to obtain the UCP signals for use in a filter coefficient computer 55. In any case, the data related to the original information about the repeat-free UCP signal, which is contained in the constant part of the internal memory of the computer 55, must be modified in accordance with the standard UCP signal. When formulating patent claims for a method announced after a specification, this applicability of the methods of various UCP standards should be taken into account.
Anyone who understands the programming of microcomputers, after reading the procedures for suppressing repeats in television signals, will be able to make a large number of variations based on the method described here, and this fact should be taken into account when formulating patent claims for a method that follows so that the expansion area of any of these items includes as many possible options as the spirit of the invention embodies.
Priority Applications (3)
|Application Number||Priority Date||Filing Date||Title|
|US07/984,486 US5331416A (en)||1992-12-02||1992-12-02||Methods for operating ghost-cancelation circuitry for TV receiver or video recorder|
|Publication Number||Publication Date|
|RU93053753A RU93053753A (en)||1996-06-10|
|RU2138922C1 true RU2138922C1 (en)||1999-09-27|
Family Applications (1)
|Application Number||Title||Priority Date||Filing Date|
|RU93053753A RU2138922C1 (en)||1992-12-02||1993-11-30||Method for controlling image repetition blanking circuits for tv sets and video cassette recorders|
Country Status (8)
|US (1)||US5331416A (en)|
|EP (1)||EP0600739B1 (en)|
|JP (1)||JP3025402B2 (en)|
|KR (1)||KR0164053B1 (en)|
|CN (1)||CN1088044A (en)|
|DE (2)||DE69326051T2 (en)|
|RU (1)||RU2138922C1 (en)|
|TW (1)||TW229347B (en)|
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|Publication number||Priority date||Publication date||Assignee||Title|
|RU2372741C2 (en) *||2006-05-16||2009-11-10||Сони Корпорейшн||System of data transmission, transmission device, receiving device, method of data transmission and program|
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|US6937292B1 (en) *||1992-04-22||2005-08-30||Samsung Electronics Co., Ltd.||Ghost cancellation reference signal with bessel chirps and PN sequences, and TV receiver using such signal|
|US5532755A (en) *||1994-01-05||1996-07-02||Samsung Electronics Co., Ltd.||Apparatus for suppressing ghosts in signals modulating a carrier in quadrature phasing with a video carrier|
|US5483292A (en) *||1994-03-09||1996-01-09||Samsung Electronics Co., Ltd.||Symbol clock regeneration in digital signal receivers for recovering digital data buried in NTSC TV signals|
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|MM4A||The patent is invalid due to non-payment of fees||
Effective date: 20101201