KR0169617B1 - Video tape recorder with tv receiver front end and ghost-suppression circuitry - Google Patents

Abstract

The ghost suppression circuit is connected to receive the composite video signal from the video detector in front of the TV receiver. Suppressing at least one ghost in response to the composite video signal, the response being fed to the recording electronics of the videotape device, the recording electronics together with the sound signal supplied from the sound detector in front of the TV receiver. Use helical injection to record the response. This is done instead of supplying the recording electronics of a videotape machine with a composite video signal taken directly from the video detector in front of the TV receiver. It also suppresses ghosts on the screen of TV receivers which produce TV images in response to TV signals supplied from videotape devices during playback from recorded videotape, without having to rely on ghost suppression circuitry at the TV receivers. Since the time-base stability of the ghost cancellation reference signal in the regenerated composite video signal during playback from the recorded video tape is weakened due to headswitching during helical scanning, the performance of any ghost suppression circuit in the TV receiver is weak. Eliminate potential problems.

Description

Video tape recorder with television receiver front end and ghost suppression circuit

1 is a front end of a television (hereinafter referred to as TV) receiver comprising a sound detector and a video detector used to supply a sound signal and a composite video signal for recording as a combination according to the present invention, and the composite video signal is a video tape recorder. Schematic diagram showing the coupling relationship between a ghost suppression circuit passing before being supplied to a video tape device with recording capability.

2 is an internal schematic diagram of a ghost suppression circuit in FIG.

3 is a schematic diagram of a circuit for resetting the modulo-8 field counter of FIG.

4 is a flowchart of a ghost suppression method using FIG.

5, 6 and 7 are schematic diagrams of a combination of a TV receiver and a video tape device, each constructed according to the principles of the present invention and called a combo.

TECHNICAL FIELD The present invention relates to video tape recorders and players, and more particularly to cassette types using helical scan recording and playback.

TV technicians have been deeply interested in ghost suppression circuits contained in TV receivers having a display device that reproduces TV images in a form suitable for human viewing. Ghost images, generated by multiplexing reception and commonly called ghosts, typically occur on TV screens that are broadcast wirelessly or transmitted over the wire.

A signal synchronized with a TV receiver is called a reference signal, which is usually a direct signal received on the shortest transmission path. Multiplex signals received on different paths are generally delayed with respect to the reference signal and appear as a rear ghost image. However, the direct signal, that is, the shortest signal, may not be the signal to which the TV receiver is synchronized. When the receiver is synchronized to the reflected (longer path) signal, the front ghost image is present by the direct signal, or many front ghosts are present by the reflected signal with less delay than the reflected signal synchronized with the direct signal and the receiver. Multiplex signals vary in terms of number, amplitude, and position-to-position delay time from channel to channel. In addition, the parameters of the ghost signal may be time variable.

The visual effects of multi-channel distortion can be broadly classified into two categories, namely, distortion of the frequency response of multiple images and channels. Both visual effects are caused by the variation in time and amplitude of the multiplex signals reaching the receiver. When the delay associated with the multiplex signal with respect to the reference signal is sufficiently large, the visual effect is viewed as a plurality of identical images on the TV display which are moved horizontally with each other. These same images are sometimes referred to as macroghosts to distinguish them from microghosts which will be explained soon. Normally the direct signal prevails and the TV receiver is synchronized to this direct signal. In this case, the ghost image is shifted to the right in the change position, intensity, and polarity. These images are known as trailing ghosts or post-ghosts. If the receiver is not synchronous with the reflected signal, one or more ghost images are moved to the left of the reference image and these ghosts are known as leading ghosts or pre-ghost images.

A relatively short delay multiplex signal with respect to the reference signal does not cause the same identifiable image of the dominant image separately, but leads to distortion in the frequency response characteristics of the channel. In this case, the visual effect is represented by the sharpness of the increased or decreased image, and in some cases, the loss of the image information. Short delays or close-in ghosts are most commonly caused by unterminated or incorrectly terminated high frequency transmission lines such as antenna lead or CATV lead cables. In a CATV environment, there may be a large number of near ghosts caused by the multiple taps improperly terminating the incoming cable of various lengths. Many of these near-ghosts are mainly called micro-ghosts.

The long multiplexing effect, or macro ghost, is typically reduced by the cancellation scheme. Short multipath effects, or micro ghosts, are typically mitigated by waveform equalization, typically peaking and / or group delay compensation of high frequency video responses.

The characteristics of the transmitted TV signal are known in the art and it is possible, at least in theory, to use these characteristics in systems of ghost signal detection and cancellation. Nevertheless, several problems limit this proximity. Instead, for example, it is known that it is desirable to repeatedly transmit a reference signal located in a TV signal section that is not currently used for video purposes and uses this reference signal for detection and cancellation of ghost signals. Typically lines are used in the vertical blanking interval (VBI). This signal is referred to herein as a ghost can celing reference (GCR) signal.

The method of canceling ghosts in a TV receiver depends on the transmitted GCR signal due to the same multiplexing distortion, which is the remainder of the TV signal. In addition, the circuitry at the receiver can examine the received distortion GCR signal and can construct an adaptive filter that cancels or at least attenuates the distortion multiplely in the prior art on the waveform of the distortion-free GCR signal. The GCR signal does not need much time in the VBI (preferably just one TV line), but enough information to allow the circuitry at the receiver to configure the compensation filter to analyze the distortion in multiple and cancel the distortion. Should have

The GCR signals are used in a TV receiver to calculate the adjustable weighting factor of the ghost erase filter, through which the composite video signal from the video detector passes to provide a ghost suppressed response. Since the weighting factor of this ghost elimination filter is adjusted, it has a filter characteristic complementary to that of the transmission medium causing the ghost. The GCR signal can additionally be used to calculate the adjustable weighting factor of an equalization filter that is dependent on the ghost-erasing filter, which in turn feeds the intrinsic planar frequency spectral response to the transmitter residual band amplitude modulator transmission media, TV receiver shear and dependent ghost-erasing. To provide a pure transmission path through the / equalization filter.

W. Ciciora, IEEE Report on Consumer Electronics, Vol. 25, pp. 2/79, pp. 9-43, shows that GCR signals can be properly (sin x) / x waveforms in A Tutor-ial on Ghost Canceling in Television Receivers. He points out. Such a properly windowed waveform exhibits a constant spectral energy density relative to the frequency band involved. The ghost position can then be determined, and the filter can be configured for ghost signal cancellation to reduce the effect of long multiplexing, and can be configured for waveform equalization to reduce the effect of short multiplexing.

Under the heading GHOST CANCELLING CIRCUIT of U.S. Patent No. 4,897,725, filed with Tanaka on January 30, 1990, the transmission reference signal or GCR signal is actually a BTA (Japanese) GCR signal presented (sin x) / x waveform as the main reference signal. Or as a ghost suppression signal. This (sin x) / x waveform received with the ghost is Fourier transformed to provide a set of Fourier coefficients. The Fourier transform of the ghosted GCR signal is then subjected to an undamaged GCR to estimate ghost gain filter parameters: tap gain information for infinite impulse response (IIR) ghost suppression filters and finite impulse response (FIR) waveform equalization filters. It is processed with the Fourier transform of the signal.

In the title of GHOST CANCELLING REFER-ENCE SIGNAL TRANSMISSION / RECEPTION SYSTEM of U.S. Patent No. 4,896,213, filed on January 23, 1990 in Kobo, the ghost component is derived from group delay distortion and frequency-to-amplitude characteristic distortion. A system with a built-in ghost canceller for reducing or eliminating is disclosed. The digital signal consisting of the frame synchronizing signal, the clock synchronizing signal and the data signal is generated during the VBI scanning line and superimposed on the TV signal to be transmitted. At the receiving end, the digital signal is used as a ghosted CGR signal in an array that correlates with a ghostless CGR signal to control adaptive filtering of the video signal to reduce ghosting.

The Bessel pulse chirp signal is a component of the GCR signal used as a standard for TV broadcasting in the United States. The energy distortion in the Bessel pulse chirp signal has a frequency spectrum that extends continuously over the video frequency band. The chirp starts at the lowest frequency and sweeps from there to the highest frequency of 4.1 MHz. The chirp is inserted into the first half of the selected VBI line, with the 19th line of each field being preferred at present. The chirps are on a +30 IRE pedestal, rise sweep from -10 to +70 IRE, and start at a predetermined value after the back segment of the preceding horizontal sync pulse. Spy appears in the 8-field cycle, where the first, third, fifth, and seventh fields have a positive polarity of the burst, and the second, fourth, sixth, and eighth fields have negatively defined bursts. Have different polarity. The initial lobe of the spy signal ETP appears in the first, third, sixth, and eighth fields of an 8-field cycle sweeping up from the +30 IRE pedestal to the +70 IRE level. The initial lobe of the spy signal ETR appears in the second, fourth, fifth, and seventh fields of the 8-field cycle swing down from the +30 IRE pedestal to the -10 IRE level, and the initial lobe is the complement of the ETP spy signal. .

The title of ADAPTIVE TELEVISION GHOST CANCELLATION SYSTEM INCLUDING FILTER CIRCUITRY WITH NON-INTEGER SAMPLE DELAY, issued US Pat.

METHOD AND APPARATUS FOR COMMUNICATION CHANNEL IDENTIFICATION AND SIGNAL RESTORATION of US Patent No. 4,864,403, filed on September 10, 1991 in Koo, describes a method and apparatus for estimating ghost suppression filter parameters in a TV receiver.

The title AUTOMATIC WAVEFORM EQUALIZING SYSTEM FOR TELEVISION RECEIVER of US Patent No. 4,044,381, issued August 23, 1977 to Shimano, describes a waveform equalization filter that can be used to suppress microghosting.

In the title of METHOD OF DETECTING SIGNAL WAVEFORM DISTURBANCE IN RECEIVED TELEVISION SIGNAL of US Pat. A pair-wise combination of VBI intervals with an in-phase reference signal is described.

Since the known degree of ghost cancellation depends on the fineness of the cancellation method, the timebase stability of the GCR signal in the received TV signal is important for the procedure for determining the weighting for the ghost cancellation and equalization filter by analyzing the GCR signal. The theoretical effectiveness of the ghost cancellation procedure using weighted summation of differentially delayed video signals depends on the same signal as the other delay that caused the ghosted signal. If the length of the scan lines differs during the transmission of the GCR signal rather than during other parts of the video signal, the weighting determined to generate the ghost-free GCR signal by the weighted sum of the various delayed GCR signals is variously delayed. It would not be suitable to generate ghost-free images at other times due to the weighted sum of the video signals. In TV receivers with display devices and ghost cancellation circuits, the time-base stability problem of detected video signals is not a problem when receiving airwave signals or those signals relayed by cable or co-receipt systems. .

Problems that have not been evaluated by TV receiver designers working in the field of TV ghost cancellation are high frequency (rf) from a home video cassette recorder (VCR) in which a TV receiver with display and ghost cancellation circuits records TV signals with ghosts. It does not satisfactorily fulfill its ghost cancellation procedure when receiving a signal. In the prior art, the position of the ghost erase circuit is next to the video detector in the TV receiver with the display, where the ghost cancellation circuit could be used for high frequency signals received from air propagation, wired or video recording media.

The term television set refers to kinescopes, power supplies for kinescopes, reflection circuits and kinescopes for driving kinescopes, loudspeakers, stereophonic sound detectors or audio amplifiers. It is used in the specification to describe a TV receiver front end involving parts of the TV receiver combined to convert to a color signal for the purpose. Conventional video cassette recorders (VCRs) have a receiver front end without these additional items, and these additional items are referred to by the term TV monitor in the specification and the accompanying drawings. If a VCR and TV set combined into a single device called a combo is required to record a program received on one channel while displaying the program received on another channel, two TV receiver fronts, one recording function Branches for video tape equipment and another for a TV receiver with video display functionality.

Consider placing a ghost cancellation circuit after the video detector in front of a TV receiver used to supply composite video and sound signals for recording to an electromagnetic storage medium. Ghost suppression circuits are estimated to be $ 50 as part of the manufacturing value, which represents an increase of approximately $ 50 or more in retail value of the appliance containing the circuits. On average in terms of years of service, the average service is that the TV set is longer than the VCR, which is why the distinction between VCR and TV sets is commercially more popular than the TV / VCR combo.

Ghost suppression circuits are quite expensive, so it would be economically better to have ghost suppression circuits in longer serviceable instruments. A ghost suppression circuit is required in any case after a video detector in a TV receiver with a display, so to ghost suppress the composite video signal received over a cable TV classification system or over the airwaves with an antenna, up to about $ 150 VCR or TV. The expectation of increasing the value of the / VCR combo causes the device designer to avoid the consideration of installing a ghost suppression circuit after the video detector in front of the TV receiver used to supply the composite video signal for recording.

The inventors have recognized that it is not right that TV design engineers generally believe that ghost suppression circuits should only be installed in TV receivers with displays. Those unfamiliar with home video recording design are not happy to think about the time-base stability problem of signals based on ghost cancellation. This is because such situations do not occur, especially when using airborne broadcast signals, such as laboratory generators or high frequency signals, on TV receivers, and have been practiced in developing ghost suppression procedures and suppression circuits. Home VCRs use helical scanning of electromagnetic tape with headswitching that occurs immediately before the vertical return period. There is time-base instability in video signals recorded on electromagnetic tape during playback, and under real circumstances, time-base instability often persists during the vertical retrace period, and some of the video signals used to generate the peaks of the picture on the display of the TV receiver. It has continued to any extension line in the primary active line. Thus, the ghost suppression circuit in a TV receiver receiving a high frequency (r-f) signal modulated in accordance with the reproduced video signal will not attempt to operate properly. The weighting factor selected in response to the evaluation of the GCR signal occurring on one scan line will not be suitable as an active video signal on another scan line since the scan line of the active video does not have the same actual duration as the scan line on which the GCR signal occurs. Beneficial timebase stabilization is followed by a differential delay followed by a linear delay to separate the GCR signal component from the accompanying horizontal sync pulse, front porch, back porch with color burst, and the +30 IRE GCR signal pediment. This is necessary for the 19th scan line of several fields to be combined. If the scan lines are counted to facilitate differential delay using a temporary digital memory, these accompanying components will not erase well if there is an error in the sample timing of the ninth scan line. In general, home VCRs cannot provide timebase stability, which is a requirement for separating GCR signals.

According to the present invention, the basic lesson to be learned from the inventors is that it is not practically possible in a composite video signal retrieved from a video tape on which ghost suppression, in particular macro ghosts, is satisfied when a relatively clean video tape recording and playback device is used. Ghost erasing should be done before tape recording and playback procedures that cause timebase instability will interfere with ghost erase. Otherwise, when a TV signal with ghost is recorded, it is reproduced for supplying the TV signal to the TV set, even if it includes a ghost cancellation circuit that satisfactorily suppresses the ghost when receiving the TV signal over the air or by wire. When the videotape results, the ghost appears in the regenerated image.

According to one aspect of the present invention, the ghost erasure made before recording the tape becomes a tape that can be reproduced to supply the ghost suppressed signal to the TV set. The reproduced image will be free of ghost even if the TV set does not have a ghost suppression circuit. This aspect of the invention is implemented in the following apparatus. That device comprises a video tape device comprising elements and at least a video tape front end comprising a sound detector and a video detector, and at least a recording electronic element which receives for recording signals from the sound detector and the video detector; Circuits are connected to receive the composite video signal from the video detector in front of the TV receiver, and instead of supplying the recording electronics of the videotape device with the composite video signal taken directly from the video detector in front of the TV receiver, the composite video signal It is implemented as an improved combination in which a response to the video tape device is supplied to a recording electronic device and at least one ghost is suppressed in response to the composite video signal.

Another aspect of the invention is a TV having a display device, which is required next to a video detector in front of a TV receiver that supplies a composite video signal for recording by a ghost suppression circuit in a combination of a TV receiver and a video tape device called a combo. In addition to the ghost suppression circuits needed next to the video detector in the receiver, a single computer can be used to calculate the parameters for the filters in the two sets of ghost suppression circuits.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings.

1 shows a video tape device 10 having a recording function, which can be, for example, a VHS, super VHS or Betamax type video cassette recorder (VCR).

In addition, video tape device 10 was released on May 12, 1992 in C.H. It may be an improved VHS recorder of the type disclosed under the title VIDEO SIGNAL RECORDING SYSTEM ENABLING LIMITED BAND WIDTH RECORDING AND PLAYBACK of US Patent No. 5,113,262 to Strolle.

The TV receiver front 20 supplies a sound signal and a composite video signal recorded by the video tape device 10 in response to the received high frequency TV signal. The high frequency TV signal is captured by the aerial TV antenna 30 after airing, as an example. Optionally, the high frequency TV signal may be provided over the wire by co-receipt or other TV cable service. The TV receiver front 20 includes parts of a typical TV receiver commonly used in combination with a videotape device having a recording function. These parts mainly include high frequency amplifiers, down frequency converters or first detectors, at least one intermediate frequency amplifiers, video detectors or second detectors and sound demodulators (often of intercarrier type). The TV receiver front 20 additionally includes separate circuits for the horizontal and vertical sync pulses.

The sound signal from the sound demodulator in the TV receiver front 20 is demodulated from the frequency modulated sound carrier while being heterodyned to an intermediate frequency by the downlink frequency converter. Prior to this demodulation, the frequency modulated sound carriers are limited in order to eliminate amplitude variations, the capture of which suppresses the response to ghosts in the sound signal from the sound demodulator. Thus, the sound signal from the sound demodulator in the TV receiver front 20 is supplied directly to the video tape device 10 for recording in a conventional manner.

The composite video signal from the video detector in the TV receiver front 20 is fed to the ghost suppression circuit 40 to cancel or suppress the accompanying ghost. As a result, the ghost suppressed composite video signal is supplied from the ghost suppression circuit 40 to the video tape apparatus 10 for recording in the conventional manner. Ghost suppression circuit 40 may be any of the known technology types.

2 shows one form that the ghost suppression circuit 40 can do. Ghost suppression circuits are suitable for use with Bessel-chapped GCR signals inserted into the first half of the ninth VBI line of each field. The composite video signal supplied from the receiver front 20 to the ghost suppression circuit of FIG. 2 is digitized by an analog-to-digital converter (ADC) 50. The ADC 50 will typically supply eight parallel bit samples of the quantized composite video signal. The quantized composite video signal is an input signal as a cascade of post-ghost cancellation filter 51, an adaptive filter of IIR type, a cascade connection of preghost cancellation filter 52, an adaptive filter of FIR type, and an adaptive filter of type FIR. Applied as cascade of equalization filter 53.

The output signal of the filter dependence is a ghost suppressed digital composite video signal, which is converted into a ghost suppressed analog composite video signal by a digital to analog converter (DAC) 54. Ghost suppressed analog composite video signals are supplied to videotape devices 10 that record using VHS, Super VHS or standard Betamax as an example. The digital-to-analog converter 54 is not used in the prior design in which the videotape device 10 records digital signals rather than analog signals.

The filter coefficient computer 55 calculates weighting coefficients for the adaptive filters 51, 52 and 53. These weighting coefficients are binary numbers, and the filter-coefficient computer 55 records the binary numbers in the recorders in the digital filters 51, 52 and 53. The weighting coefficient stored in the recorder in the IIR filter 51 is used as a multiplier signal in which the digital multiplier receives a filter output signal as a multiplier signal having various delay sums. The product signals from the digital multiplier are combined by adding and subtracting in the digital adder / subtractor circuit to generate an IIR filter response. In each FIR filter 52 and 53, the weighting coefficient stored in the recorder is used as a multiplier signal by which the digital multiplier receives a filter input signal as a multiplier signal having various delay sums. In each FIR filter 52 and 53, the product signal from the digital multiplier is combined by addition and subtraction in the digital adder / subtracter circuit to generate the weighted sum response of the FIR filter.

Pre-ghosts in aerial reception can be shifted by 6 microseconds of displacement from the direct signal, but typical displacements are no more than 2 microseconds. Direct reception in wired reception can outstrip the signal fed into the wire by 30 microseconds. Band image response can be rolled off by 20dB at 3.6MHz, but rolloff at 3.6MHz is typically less than 10dB. The number of taps in the FIR filters 52 and 53 depends on the range of ghost suppression, and in order to maintain the filter value in commercial confines, the FIR filter 52 typically suppresses ghosts with displacements of 6 microseconds from the direct signal. To about 64 tabs. FIR filter 53, used for frequency equalization needs, has only 32 taps. Dependent FIR filters 52 and 53 are moved by a single FIR filter with about 80 taps.

Typically, the range of post ghost has about 70% of the post ghost occurring in the subrange, which extends to 10 microseconds, and extends to 40 microsecond displacement in the direct signal. The IIR postghost cancellation filter 51 needed to suppress postghost over the entire range can be as long as 600 taps. However, since postghosting occurs at discrete displacements, the weighting factor for many of these taps in filter 51 will be at or near zero. Taps that require a weighting factor far greater than zero are grouped together in groups of 10 or less. From an economical point of view of hardware, it is desirable to use only a large number of digital multipliers with weighting factors greater than zero. Thus, the tap delay line in IIR filter 51 is usually designed with a cascade of about 10 tap delay lines interspersed with a programmable delay device, making filter 51 sometimes called a sparse-weighting filter. . About 10 tap delay lines feed the signal to a digital multiplier for weighting. The programmable bulk delay devices consist of delay lines of various lengths whose chain formation can be controlled in response to control signals expressed in binary. This weighted filter includes a recorder for the binary number that specifies the delay of the programmable delay device, the content of which is also controlled by the filter coefficient computer 55.

Consider a means of supplying a ghosted GCR signal from the TV receiver front 20. Horizontal and vertical sync pulses are received from the receiver front end 20. The horizontal sync pulse is periodically reset by the vertical sync pulse and counted by a 9-stage digital counter 56 called a scan line counter, and the vertical sync pulse is counted modulo 8 by a 3-stage digital counter 57 called a field counter. do. Although these counts are omitted from FIG. 2 to reduce the complexity of the connection supplied to the filter coefficient computer 55, these counts can be used with the filter coefficient computer 55 used for timing operations. The decoder 58 responds to the scan line count from the scan line counter 56, which is 19, corresponding to the scan line in each field containing the GCR signal, and rather than the wired zero supplied with the output signal of the multiplexer 59 as the 0 th input signal. Rather, it responds to a situation for responding to the digitized composite video signal supplied from the ADC 50 as the first input signal.

RAM with read and write functionality provides temporary scan line storage 60 in FIG. 2 which may be replaced by serial memory in a selective embodiment of a ghost suppression circuit. The temporary scanning line storage section 60 is an arrangement for accumulating the 19th VBI scanning line GCR signal in each pixel unit for eight consecutive fields in the temporary filtering operation of separating the Bessel-composition information from other information generated during the 19th VBI scanning line. Is connected to. This temporal filtering operation correlates with Besseltack information that occurs during the 19th scanline to provide an improved signal / noise ratio compared to simply using gating to separate Besseltack information from the 19th VBI scanline. When the corresponding pixels of the eight GCR signals have accumulated during the 19th scan line of the field 000, which is the 8th and the last field of the 8th field order, the separated Bessel-Zip information is cleared after the 19th and the scanline storage 60 is cleared. Prior to any line of field 000, the filter coefficients are stored one pixel at a time in the recorder of the computer. In FIG. 2, the scanning line storage unit 60 erases data during the last line of the last field of the eight field order, or the erasure is performed by randomizing the field 000 after the separated Bessel-Zip information recorded in the recorder of the filter coefficient computer 55. Can occur during the line of. Further, the transmission of data accumulated in the filter coefficient computer 55 in the scan line storage 60 and the erasure of the accumulated data from the scan line storage 60 may occur during any two scan lines of the first to 18th scan lines of the field 001.

In particular, the temporary scan line storage unit 60 stores the total scan line of the 16-bit parallel sample if the 8-bit parallel sample of the 8-bit parallel sample of the quantized composite video signal supplied from the analog-to-digital converter 50 is to be accumulated as an encoded reference. Has Signed arithmetic is preferably two's complement arithmetic. In partial implementation of the arrangement to operate the temporary scan line storage 60 as a accumulator with a signal to the GCR signal, the digital adder / subtracter 61 sends a 16-bit parallel output signal to the temporary scan line storage 60 as its write input signal. Supply. The digital adder / subtracter 61 receives the output signal of the multiplexer 62 as a first input, and the output signal usually corresponds to the reading of information from the temporary scan line 60 received at the 0 th input of the multiplexer 62. The digital adder / subtracter 61 receives, as a sign bit extension, the 8-bit parallel output signal of the multiplexer 59 as a second input, as eight wired zeros.

Decoder 80 decodes a modulo-8 field count that is 1, 3, 6, or 0 (ie, 8) to supply logic 1 to digital adder / subtracter 61 on the condition of adding its input signal. Decoder 80 supplies logic 1 to digital adder / subtracter 61 to subtract logic 2 from the first input signal (supplied from multiplexer 62) from the second input signal (supplied from multiplexer 59). Decode modulo-8 field count of 5 or 7. This array stores the following functions in the temporary scan line storage unit 60.

That is, (field 001 line 19)-(field 010 line 19) + (field 011 line 19)-(field 100 line 19)-(field 101 line 19) + (field 110 line 19)-(field 111 line 19) + (Field 000 line 19) is accumulated.

During the last line of the eighth field in each order of eight fields, a control signal, usually zero, to the multiplexer 62 generates one. This 1 causes the multiplexer 62 to supply an output signal corresponding to a first input of arithmetic zero having a 16-bit parallel of zero wired. In the temporary scan line storage section 60, the result of the accumulation is reset to arithmetic zero. The control signal for multiplexer 62 is generated by a two-input and gate 63 as shown in FIG. Decoder 64 decodes the count from scan line counter 56 corresponding to the last line of the current field to generate one of the input signals to AND gate 63. Decoder 65 decodes 000 modulo-8 field counts from counter 57 to generate another of the input signals to endgate 63. The eighth field in each sequence of eight fields generates 000 modulo-8 counts from field counter 57. Both input signals to the AND gate 63 are only 1 during the eighth last line of each sequence of eight fields, and during that last Raan, the AND gate 63 brings the accumulated result stored in the temporary scan line storage 60 to be reset to arithmetic zero. 1 is supplied to the multiplexer 62 as a control signal.

The two-input end gate 66 supplies 1 to the filter coefficient computer 55 when the accumulation result stored in the temporary scan line storage 60 can be moved to the ghosted Bessel-lapper recorder in the internal memory of the filter coefficient computer. The output signal of the decoder 65 is one of the input signals to the AND gate 66 and is 1 only for the eighth field of each field of the eight fields. The two-input NOR gate 67 generates another of the input signals to the AND gate 66. Noah gate 67 responds to an output signal of decoder 64 that detects the last line of a field at a count from scan line counter 56 and responds to an output signal of decoder 68 that detects a vertical blanking interval resulting from the count from scan line counter 56. . Thus, the output signal of NOA gate 67 is 1 except during the vertical blanking interval or the last line of the field. Thus, the accumulation result in the temporary scan line storage 60 can be moved to the internal memory of the filter coefficient computer 55 at any time during the eighth field of each field order of eight fields except during the last scan line or during the vertical blanking interval.

The clocking of the timing pixel sampling by the analog-to-digital converter 50 and the addressing of the temporary scan line storage 60 are now described. Oscillator 70 with automatic frequency and phase control (AFPC) generates a sinusoidal oscillation at the second harmonic of the color subcarrier frequency as the original clock signal. The zero-crossing detector 71 detects the average intersection of sinusoidal oscillations to generate a pulse at a rate four times the color subcarrier frequency. These pulses adjust the sampling of the composite video signal that is digitized by the analog / digital converter 50, and if the temporary scan line storage 60 is serial memory, it will adjust the data progress there. In the ghost suppression circuit of FIG. 2, the temporary scan line storage section 60 is a RAM arranged for reading and writing when each of its storage locations is addressed. The address of each storage location is repeatedly scanned in accordance with the count of pixels supplied from the 10-stage digital counter 72 named as the pixel counter for counting pulses from the zero crossing detector 71. These same addresses are multiplied by the filter coefficient computer 55 to be used for addressing the scan line storage recorder when the separated GCR signal is moved from the temporary scan line storage 60.

In general, if a color burst signal is present, it is the most stable reference in the composite image signal and is the preferred reference signal for the AFPC of oscillator 70. The overflow signal from the second stage of the pixel counter 72 is probably a 3.58 MHz square wave, and the overflow signal is supplied as a feedback signal to the first AFPC detector 73 which compares the separated burst signal to generate an error signal, and the AFPC signal. Detector 74 optionally applies to pixel counter 72 which controls the frequency and phase of its oscillator. Burst gate 75 responds to a pulse from burst gate control signal generator 76 to separate the color burst signal to be supplied to the first AFPC detector 73 and the analog composite image signal to be supplied at the front end of the TV receiver. Horizontal sync pulses from the front end of the TV receiver 20 are supplied to the burst gate control signal generator 76 and their rear sections are used by the burst gate control signal generator 76 to adjust the pulses that occur during the color burst interval. Unstable flip-flops or one-shots are routinely used to generate these pulses.

The decoder circuit 68 responds to the scan line count provided by the scan line counter 56 and corresponding to the VBI line in each field to generate the inhibit signal. This prohibition signal is applied to the burst gate control signal generator 76 to prevent its generating pulse, such that burst gate 75 only selects those back porch intervals during a field that may have a color burst. (In another embodiment, the burst gate control signal generator 76 is not inhibited from generating a burst gate pulse during the vertical blanking interval and the time constant of the first AFPC detector is made longer than in the FIG. 2 circuit.)

An amplitude detector 77 called a color burst presence detector when a burst is present in the output signal from burst gate 75 to supply 1 which causes the AFPC signal multiplexer 74 to select the output signal from the first AFPC detector 73 as the first error signal. Detects a burst to apply to the control oscillator 70 as its AFPC signal. In terms of immunity to noise, the amplitude detector 77 preferably has a configuration in which a synchronous detector stage, followed by a threshold detector stage, is then a short-pulse canceller. An array of pixel counters 72 may be made to provide a 3.58 MHz square wave pair in a quadrature phase relationship to each other for application to the synchronous detection portions of detectors 73 and 77. Counter arrays for providing square waves that are orthogonal to each other are commonly used in TV stereophonic sound decoders and are familiar to TV circuit designers. The short-pulse canceller is known from the radar and is constructed using circuitry to AND gate the differentially delayed input signal to generate an output signal therefrom.

When a monochrome TV signal is received without the accompanying color burst, the reference signal for the AFPC of oscillator 70 should be a separate horizontal sync pulse supplied to the AFPC circuit from the front 20 of the TV receiver. When the color burst presence detector 77 does not have a color burst accompanying the composite video signal supplied from the front end of the TV receiver 20, the AFPC signal multiplexer 74 uses the output signal from the second AFPC detector 78 to the control oscillator 70 as its AFPC signal. We will supply 0, making a choice. The sync decoder 79 theoretically responds with a 1 to the count of the pixel counter 72 in response to the occurrence of a horizontal sync pulse such as an edge or a predetermined interval. The output signal from the synchronous decoder 79 is supplied as a feedback signal to the second AFPC detector 78. The second AFPC detector 78 compares the feedback pulse with the input reference signal obtained from the horizontal synchronous pulse supplied from the horizontal synchronous separator at the front end of the TV receiver. A second error signal that is selectively applied by the signal multiplexer 74 is generated to the control oscillator 70 as its AFPC signal. This AFPC array is referred to by TV engineers as line-locked-clock.

The oscillation stability of the controlled oscillator 70 is temporary for line accumulation, with the Bessel-Strap separate from the 19th scanline, the back porch containing the horizontal sync pulse, front porch, color burst and the +30 IRE pedestal fully accumulated. It is necessary over the number of fields from the 19th scanning line taken for the accumulation in section 60. The correction control of the oscillation frequency is practically necessary and the aspect of the automatic phase control (APC) of the AFPC should be superior to the automatic frequency control (AFC) of the AFPC which has a very long time constant, that is, several field lengths.

The circuit for resetting the counters 56, 57, 72 is omitted in Figure 2 to avoid unnecessary complexity. The scan line counter 56 can simply be reset by the front section of the vertical sync pulse supplied from the vertical sync separator at the front end of the TV receiver.

The pixel count from the pixel counter 72 is reset as necessary to resynchronize it with the scan line in the composite video signal supplied from the video detector 20 before the TV receiver. The front and rear sections of the horizontal sync pulses supplied from the horizontal sync separator in front of the TV receiver 20 are detected using a differentiator followed by a suitable level comparator. The forward segment detector result is used to command the loading of the temporary store recorder with the current pixel count. The pixel count is applied to the window comparator to determine if it is within the expected range and to generate an indication of an error if it is not within the expected range. The count of the pixel counter 72 is conditionally reset to zero in response to the back section detector result. The condition for reset can be a single indication of pixel count error. However, better noise immunity is obtained by counting errors in the configured reversible counter, so that a certain number of consecutive errors must be counted before the pixel count is corrected.

3 shows a circuit for resetting the modulo-8 field counter 57, the count of which is directly phased or incorrectly phased by the four fields. The temporary scan line storage section 31 is shown as RAM addressed by the pixel count supplied from the pixel counter 72. The scan line storage 31 is arranged for read and write operations. Logic 1, which occurs by decoder 58 only during the 19th scanline of each field, is supplied to multiplexer 310 to condition updating of temporary scanline storage 31 with the quantized 19th scanline sample supplied from analog / digital converter 50. . Logic 0 caused by decoder 58 during another scan line causes multiplexer 310 to apply the data read from temporary scan line storage 31 for rewriting.

The temporary scan line storage section 31 is supplied with pixel latches 32 and 33 clocked by the output signal from the zero-crossing detector 71. Pixel latches 32 and 33 are used to temporarily store the last pixel written to the temporary scan line storage 31 and the last pixel removed from the temporary scan line storage 31, respectively. The samples of the difference signal from subtractor 34 will all be zero except during the 19th scan line. The difference signal from the subtractor 34 is supplied to an absolute value circuit 35, which is a two-input exclusive which receives, as a first input, each other bit of the difference signal that is selectively complementary with the sign bit of the difference signal. An OR gate battery may be provided and a digital adder may be further added to add the remaining complementary bits of the difference signal to generate the absolute value of the difference signal as the sum output signal.

The accumulator 36 for successive samples of the output signal of the absolute value circuit 35 has an output latch 361 for temporarily storing the successive values of the accumulation result and the succession of the absolute value circuit 35 output signal in the accumulation result to expand its value. A digital adder 362 for adding typical samples, and a multiplexer 363 for selectively supplying an expanded accumulation result to an output latch 361 that updates its contents. The multiplexer 363 is wired to insert arithmetic 0 into the output latch 361 when the decoder 58 does not detect the counter 56 which supplies the 19th scan line count. Decoder 364 is responsive to the pixel counter from counter 72 representing portions of the scan line that may contain Bessel-stack information to supply 1 and is AND gated with the output signal from 0-crossing detector 71 at AND gate 365. Output latch 361 is clocked to receive input data that only responds to 1 received from endgate 365.

Successive samples of the absolute value of the difference of the 19 lines of the current and previous fields supplied in series from the absolute value 35 are accumulated using the accumulator 36. The accumulation result shall have a value that can be evaluated if the current field is not field 001 or 101. The 19th lines of fields 000 and 001 both contain ETP signals, so their difference is zero except noise. The 19th lines of fields 100 and 101 both contain an ETR signal, so their difference is zero except noise. When the accumulation result is substantially greater than or equal to zero, the output signal of the threshold detector 37 is zero, and the output signal is supplemented by a NOT gate 38 to supply one of the four input signals of the AND gate 39. do. Decoder 41 detects the field counts from counters 57 different from 001 and 101 to supply the AND gate with a 1 indicating that the field counts are out of phase and enabling the reset of counter 57. The output signal of the decoder 58 detects the occurrence of the 19th line of the field, and the output signal of the decoder 42 responds to the pixel count from the counter 72 to detect the end of the scan line, and the two output signals of the AND gate 39 The other two input signals. If the field count is not 001 or 101, the AND gate 39 generates 1 to reset the counter 57 to the 001 field count at the end of the 19th line of field 000 or 100 in the television signal received by the TV receiver front end 20. . Optionally, counter 57 can be reset to 101, or it can prepare to reset only the two least significant bits of the field count to reset them to 01.

Referring back to FIG. 3, if the modulo-8 field count provided by the field counter 57 is just in phase, the accumulation result obtained from the temporary scan line storage unit 60 during the field 000, which is the last field in the cycle of accumulation, is equal to the horizontal synchronization pulse, It will be eight times the front porch, the back porch with color burst, and the ETP vessel-chapped signal without the +30 IRE pedestal. On the other hand, if the modulo-8 field count provided by the field counter 57 is incorrectly phased by four fields, the accumulation result obtained from the temporary scan line storage unit 60 during the field 000, which is the last field in the accumulation cycle, is horizontal sync pulse, front. It will be eight times the porch, back porch with color burst, and ETR Bessel-chapped signal without the +30 IRE pedestal. The three binary position shifts wired in the reduced magnitude direction divide the accumulation result obtained in the temporary scan line storage 60 during field 000 by 8 and the share of the result is supplied to the filter coefficient computer 55 as an ETP or ETR signal.

The filter coefficient computer 55, which is well adapted to perform correlation to the ghost-prevessel superposition function ETP or ETR stored in its internal register, is the input received from the temporary scan line storage 60 during field 000 is an ETP signal, an ETR signal, or It is programmed to perform a correlation boost step to determine if it is not correlated with the ETP, ETR signal. This procedure enables the filter coefficient computer 55 to determine when no GCR signals are included in the telephone signal received by the TV receiver front 20. The computer 55 can then apply certain weighting factors stored in the recorder therein to the filters 51, 52 and 53. Optionally, computer 55 may weight factors for filters 51, 52, and 53 resulting from data relating to the received ghost supplied by means that do not depend on the GCR signal included in the television signal received by TV receiver front end 20. It can be arranged to calculate.

In another variation of the FIG. 3 circuit, an external circuit to the computer 55 is provided to analyze the GCR signal stored in the temporary scan line storage 31 to determine if it is an ETP or ETR signal (eg, during the scan line following the capture). This decision is used to determine if the most significant bit of the reset state for field counter 57 is zero so that reset is 001 field count or 1 is reset to 101 field count. The contents of the temporary scan line storage unit 31 are scanned according to the pixels from the counter 72 during the analysis procedure.

In an exemplary analysis procedure, portions of the pixel count corresponding to the initial lobe of the vessel chirp are used to enable accumulation by one of the two accumulators. One accumulator is further required that the sign bit of the current GCR signal is zero in order to accumulate its magnitude (absolute value) above the threshold T. The other accumulator further needs to have the sign bit of the current GCR signal equal to 1 to accumulate its magnitude (absolute value) above the threshold T. After the portion of the pixel count that corresponds to the initial lobe of the vessel chirp is scanned, the comparator contents are compared to each comparator of threshold T, the magnitude of which is equal to the absolute integer of the initial lobe of the vessel chirp. If the contents of the accumulator that require the sign bit of the current GCR signal to be zero for accumulation exceed this threshold T after the initial lobe of the vessel bundle, then the comparator associated with the accumulator is decomposed to the filter coefficient computer 55. Supply 1 to indicate presence. Conversely, if the contents of the accumulator that require the sign bit of the current GCR signal to be 1 for accumulation exceed this threshold T after the initial lobe of the Bessel chirp, the comparator associated with the accumulator will return the ETR to filter coefficient computer 55. Supply 1 to indicate the presence of the signal. If the contents of one of these accumulators after the initial lobe of the Bessel chirp do not exceed this threshold T, the associated comparators do not have an ETP or ETR signal present in the television signal the apparatus of FIG. 2 attempts to suppress ghosts. Supply 0 to computer 55 to determine that. In an improvement of this configuration, the threshold T is adjusted in response to noise and GCR signal amplitude conditions.

In the modification of the ghost suppression circuit in FIG. 2, when data is moved from the temporary scan line storage unit 60 to the scan line storage recorder in the filter coefficient computer 55, the addressing of the temporary scan line storage unit 60 and the scan line storage recorder moved is performed by the pixel counter 72. Instead occurs within computer 55.

The multiplexer under the control of decoder 58 or computer 55 may apply addresses to temporary scan line storage 60 while selecting addresses from pixel counter 72 during the 19th line of each field or selecting addresses from those provided by computer 55. . Also, variants of the ghost suppression circuit of FIG. 2 are used to enable the computer 55 such that multiple temporary scanline stores update the coefficients of filters 51, 52 and 53 on eight fieldcycles instead of a single temporary scanline store 60. Make it possible.

In another variation of the ghost suppression circuit of FIG. 2, the transient single scan line storage 60 may be replaced by a temporary two-scanning storage and the decoder 58 is the 19th-20th scanning line which conditions the multiplexer 59 loading the temporary two-scanning storage. It can be replaced by a decoder to detect the presence of them. Alternatively, the temporary single scan line storage 60 may be replaced by a temporary 3-scan line storage and the decoder 58 may be replaced by a decoder for detecting the presence of the 19th to 21st scan lines that condition the multiplexer 59 loading the temporary 3-scan line storage. Can be. These arrangements facilitate a bidirectional combination of VBI intervals that include antiphase GCR signals and other reference signals in phase with the in order to suppress longer delayed macroghosts.

Another variant of the ghost suppression circuit of FIG. 2 is the accumulation of the 19th scan lines in the temporary scan line storage 60 in 16 consecutive fields instead of eight. This further correlates the separated vessel information and improves the signal-to-noise ratio when fed to the filter coefficient computer 55. In this variant, modulo-8 field counter 57 is replaced with modulo-16 field counter. Further accumulation, ie the additional accumulation of the 19th scan line in 24 consecutive fields, shows a slight improvement in the signal-to-noise ratio of the separated Bessel chirp information supplied to the filter coefficient computer 55.

4 shows a flowchart of a procedure for setting operating parameters of the filters 51, 52 and 53, which is performed by the filter coefficient computer 55. As shown in FIG. Initiation of Procedure The start of state 81 is done when the power is turned on at the TV receiver, when a new channel is tuned, or after a certain time has elapsed since the last ghost suppression procedure. The process 82 of resetting all ghost suppression filters sets the filter coefficients at filters 51, 52 and 53 to predetermined values for the channel stored in the memory addressed to the TV receiver 20 and the channel addressed. Optionally, the filter coefficients at filters 51, 52, and 53 during power up or re-tuning can be values associated with the ghost free signal, and previous values of the filter coefficients are maintained during reset during periodic ghost suppression.

The data capture process 83 is then completed after the number of fields that the computer 55 has to wait for accumulation in the temporary scan line storage unit 60 to be completed has elapsed in order to generate a separate GCR signal which is suitable input data to the computer 55. Although not shown in FIG. 4, the data capture process 83 correlates whether the computer 55 determines whether the input received from the temporary scan line storage 60 during field 000 is an ETP signal, an ETR signal, or does not correlate with an ETP or ETR signal. Includes substeps.

Channel characteristic process 84 is then performed. In the data supplied to the computer 55, the position at the time of the dominant response is detected, where each position in successively smaller time during a very large ghost response can be suppressed by the filter 52 and the number of post ghosts that can be suppressed by the filter 51. It is up to the number of preghosts that are present. The predominant responses in the data supplied to the computer 55 and each position in time of the multiplex responses are calculated to be used as a reference for programming the bulk delay lines interspersed between the taps in the IIR filter 51. The relative strengths of the predominant responses and the multiplex responses in the data supplied to the computer 55 are calculated to be used as a reference for assigning weights to the taps of the IIR filter 51 and the taps of the FIR filter 52.

The process 85 of updating the IIR coefficients is performed next after the channel characteristic process 84 is performed, in which the programmable delays and the nonzero weighting coefficients of the IIR filter 51 are updated.

The process 86 of updating the FIR coefficients is performed after the process 85 of updating the IIR coefficients, and non-zero weighting coefficients of the FIR filter 52 are updated. After steps 85 and 86 of updating the IIR coefficients and the FIR coefficients are performed, a decision process 87 is performed in which the ghost is lower than the threshold level. If the decision is no, not all ghosts are cleared even though filters 51 and 52 have the ability to be further adjusted to suppress at least one or more ghosts or to improve ghost cancellation already suppressed, and the procedure returns to data capture process 83. . A threshold level of 30 dB down from the dominant image was used in process 87. If the determination is YES, then all ghosts are canceled or have no ability to be further adjusted to cause filters 51 and 52 to cancel at least one or more ghosts, and the procedure goes to equalization process 88 where the weighting factors for amplitude equalization filter 53 are calculated. Continue. Updating the initial one of the dependent ghost cancellation filters, which is a post ghost filter 51, causes a type of false ghost that cannot be suppressed by the last filter of these filters. Since the weighting factors calculated in the channel characterization process 84 do not attract these ghosts to the account, the initial one weighting factors of the dependent ghost cancellation filters must be recalculated to represent the compensation ghosts that will reduce the ghosts in the initial filter response. do. Since this reduction cannot be perfect, it is advisable to recalculate the weighting factor of the last one of the dependent ghost cancellation filters. The decision loop from 83 to 86 performs these recalculations.

Equalization process 88 can be performed by taking the Discrete Fourier Transform (DFT) of the cascaded responses of filters 51, 52, and 53 as the correlator response, then dividing it by the DFT of the ideal correlator response stored in the memory of computer 55. This gives a reference for calculating the necessary adjustment in the tap weighting of the FIR filter 53. Since the number of taps for FIR filter 53 is typically only 32, the number of spectrometers in the DFT is reasonably small, but the DFT calculation tends to be long. One of the fastest ways to calculate the equalization coefficients is the minimum to adjust the weighting factors of the filter 53 so that the response of the cascaded connections of filters 51 to 53 accumulated in the temporary scan line storage 60 best matches the ideal response stored in the computer 55. Is to use the unit method.

Following the equalization process 88, the procedure of FIG. 3 reaches step 89, the end of the situation. Since the high-level ghosts generated in the IIR filtering can be accounted for before the FIR filtering coefficients are calculated, the updating of the FIR coefficients 86 and the equalizing process 88 are preferably performed after the updating of the IIR coefficients 85 is performed. The FIR filtering coefficients can then be estimated to suppress their high ghosts.

The combo of FIG. 5 includes the TV antenna of FIG. 1 with a video tape device 10, a TV receiver front end 20 for that device, and an adaptive ghost suppression circuit similar to the adaptive ghost suppression circuit 40 of FIG. This ghost suppression circuit includes a filter coefficient computer 90, a GCR signal acquisition circuit 91 which requires a GCR signal from the front end of the TV receiver 20 to the computer 90, and a ghost suppression composite video signal for recording to the videotape device 10. And a ghost suppression filter 92 for ghost suppressing the composite video signal from the front end of the TV receiver. Video tape device 10 receives a sound signal for recording from a sound detector at the front end of the TV receiver.

The combo of FIG. 5 is another TV receiver front 93 where the TV antenna is selectively connected by an input selector switch 94, and another GCR signal acquisition circuit 95 that requires a GCR signal for the computer 90 from the TV receiver front 93; Another ghost suppression filter 96 for ghost suppressing the composite video signal from the TV receiver front end 93 is further included. Another GCR signal acquisition circuit 95 is similar to the GCR signal acquisition circuit 91, which may be similar to that formed by elements 56 to 78. Ghost suppressor filters 92 and 96 have cascaded connections of respective filters, similar to cascaded connections of filter 51 to 53 shown in FIG.

The ghost suppressed composite image from the ghost suppression filter 96 is fed to a luma / chroma separator 97 responsive to the ghost suppressed composite image to generate separate luminance and color signals. Chroma demodulator circuit 98 responds to the separated color signal to generate a well-known pair of color difference signals. The I and Q chrominance signals for application to the color matrix circuit 99 come with separate luminance signals. The color matrix circuit 99 generates red (R), green (G) and blue (B) color signals for application to the TV monitor 100. The TV monitor 100 responds to the R, G, and B signals from the color matrix circuitry 99, and the sound signal, horizontal sync signal and vertical sync signal from the front end of the TV receiver 93 to generate a TV image with sound for human observers. Respond to the signal.

A feature of the FIG. 5 combo is that the single filter coefficient computer 90 performs the dual function of calculating the filter coefficients for the ghost suppression filters 92 and 96 on a time division multiplex basis. Here, avoiding the use of a separate computer to calculate the filter coefficients for filters 92 and 96 is a significant cost savings. There is also some technical advantage to reduce power consumption, one of them as well.

The combos of the fifth, sixth and seventh degrees respectively are similar in that they estimate the filter coefficients for the ghost suppression filters 92 and 96 on a time division multiplex basis. The difference between these combos is that the playback sound and the playback picture are sent to the TV monitor 100.

In the FIG. 5 combo, a color-under signal recovered from a recorded video tape to generate a quadrature amplitude modulated (QAM) reproduction color signal is up-converted in the reproduction electronic element of the video tape device 10. The circuit 101 additionally combines the color signal with the reproduction luminance signal recovered from the recorded video tape to generate a reproduction composite image signal in which the composite image signal is modulated with a low power high frequency image carrier in an amplitude modulator 102. The sound signal recovered from the recorded video tape is modulated into a low power high frequency sound carrier in frequency modulator 103. Circuit 104 additionally combines the output signals from modulators 102 and 103 to generate a low power TV signal, wherein the low power TV signal is output from the TV antenna 30 when the input selection switch 94 is reproduced from the recorded video tape. It is a signal that can be selected as an input signal to the front end of the TV receiver rather than as a signal, and can also be applied to other TV receivers.

In the combo of FIG. 6, the TV receiver front end 93 includes a front portion and a multi-stage intermediate frequency amplifier 107, a sound detector 108, a video detector 109, and a synchronous separation circuit 110, which include a high frequency amplifier 105 and a down frequency converter 106. It is divided into a rear portion that includes. The selector switch 111 selects the input signal to the intermediate frequency amplifier 107 with the output signal from the downlink frequency converter 106 selected when the TV monitor 100 is used to represent the signal received via the antenna 30 or the alternative wire. The color undersignal recovered from the recorded videotape is upconverted in the playback electronics of the videotape device 10 to generate quadrature amplitude modulation (QAM) reproduction color signals. The circuit 101 additionally combines this color signal with a reproduction luminance signal recovered from the recorded video tape to produce a reproduction composite image signal in which the composite image signal is modulated to a low power intermediate frequency image carrier in an amplitude modulator 112. The sound signal recovered from the recorded video tape is modulated by the frequency modulator 113 into a low power intermediate frequency sound carrier. The circuit 114 additionally combines the output signals from the modulators 112 and 113 to generate a low power TV signal which is selected as an input signal to the intermediate frequency amplifier 107 during playback from the recorded videotape device to the TV monitor 100 by the selector switch 111. do. Upconverter 115 may be commercially available to generate a low power TV signal for application to another TV receiver. This upconverter 115 is shown to be located immediately after the additional coupling circuit 114, but it can optionally be located after the intermediate frequency amplifier 107 and thus acts as a residual sideband filter.

In the combo of FIG. 7 the TV receiver front end 93 is modulated such that the selector switch 116 selects the input signal applied to the synchronous separation circuit. During playback of the video tape on which the composite video signal from the video tape device 10 is recorded, the composite video signal from the video detector 109 is selected as an input signal to the sync separation circuit 110. During playback of the recorded videotape, an additional selector switch 117 selects the color signal from video tape device 10 to chroma demodulator circuit 98 as its input signal, or selector switch 117 outputs the color signal from luma / chroma separator 97 to circuit 98. Select as input signal. During playback of the recorded video tape, another selector switch 118 selects the luminance signal applied to the color matrix circuit 99 from the video tape device 10 as its luminance signal input, or the selector switch 118 is a luma / chroma separator. The luminance signal from 97 is selected as the luminance signal input of the circuit 99. During playback of the recorded video tape, another selector switch 119 selects the sound signal to be applied from the video tape device 10 to the TV monitor 100, or selector switch 119 selects the sound signal to be applied to the TV monitor 100 at the front end of the TV receiver. Choose.

Recorded video tape responsive to ghosted high frequency TV signals using the devices shown in FIGS. 1, 5, 6 and 7 are conventionally used to supply ghostless high frequency TV signals to conventional TV receivers without ghost suppression circuitry. Can be played on a videotape player, so there will be no ghosts in the recovered TV images. Images without ghosts will make the combinations shown in Figures 1, 5, 6 and 7 commercially more valuable.

Claims (11)

An image signal processing apparatus comprising: a front end of a television receiver comprising a sound detector for supplying a sound signal, a video detector for supplying a first composite video signal having accompanying ghosts, and associated elements; A ghost suppression circuit connected to receive the first composite video signal and generating a second composite video signal as a ghost suppression response to the first composite video signal; And a video tape device having a recording capability to receive for recording the sound signal and the second composite image input signal from the sound detector. A video signal processing apparatus comprising: a sound detector for supplying a sound signal, a video detector for supplying a first composite video signal having ghosts in the form of a continuous analog signal, and a separation from the first composite video signal A front end of a television receiver including a horizontal synchronous separator for supplying the horizontal synchronous pulse, and a vertical synchronous separator and related elements for supplying a vertical synchronous pulse separated from the first composite video signal; A video tape device having a recording capability for receiving a second composite video input signal in the form of an analog signal continuous with the sound signal of the sound detector and including recording electronic elements; A ghost suppression response to the first composite video signal in the form of a continuous analog signal, comprising a ghost suppression circuit for generating the second composite video signal in the form of a continuous analog signal, the ghost suppression circuit comprising: a ghost suppression circuit; A converter for generating the first composite video signal in the form of a continuous analog signal as a first composite video signal in the form of sampled data; A filtering parameter that can be adjusted according to a weighting factor temporarily stored in an internally contained register, and converting the second composite video signal in sampled data form in response to the first composite video signal in sampled data form. With a filter for generating; A converter for generating the second composite video signal in a continuous analog signal form in response to the second composite video signal in sampled data form; Means for storing a ghost pre ghost cancellation reference signal, the means for generating the weighting factors associated with the ghost pre ghost cancellation reference signal and responsive to ghost ghost cancellation reference signals, A filter coefficient computer for generating the weighting coefficients for temporary storage in a register; A scan line counter counting a horizontal synchronous pulse supplied from the horizontal synchronous separator to generate a scan line count, and periodically reset to an initial value of the scan line count in response to a vertical synchronous pulse supplied from the vertical synchronous separator; And means for responding to said scan line count reaching a predetermined value to capture a current scan line of said first composite video signal as a component of said ghost erase reference signal. 3. The apparatus of claim 2, further comprising: a field counter for counting, in units of vertical sync pulses, supplied from the vertical sync separator to generate a field count; Means for accumulating scan lines of the first composite video signal captured as a component of the ghosted ghost cancellation reference signal in individual pixel units until a zero return signal is received; Means for responding to a scan line count and a field count reaching a first predetermined value and a second predetermined value, respectively, to generate the zero return signal; And means for transmitting said ghosted ghost cancellation reference signal to said filter coefficient computer after accumulation in said accumulation means. 4. The apparatus of claim 3, further comprising: a control oscillator for generating sinusoidal oscillations at a frequency controlled by a generator control signal; A burst gate generator responsive to horizontal synchronous pulses supplied from the horizontal synchronous separator to generate burst gate pulses each delayed by a predetermined time period from one pulse corresponding to the horizontal synchronous pulse; Means for separating a color burst portion of said first composite video signal responsive to said burst gate pulses; And means for generating the oscillator control signal responsive to the separated color burst portions of the first composite video signal. 4. The apparatus of claim 3, further comprising: a control oscillator for generating sinusoidal oscillations at a frequency controlled by an oscillator control signal; Means for generating the oscillator control signal responsive to horizontal sync pulses supplied from the horizontal sync separator; And means for responding to the sinusoids to determine the timing of sampled data generated by the converter for generating a first composite video signal in sampled data form. 3. The apparatus of claim 2, further comprising: a control oscillator for generating sinusoidal oscillations at a frequency controlled by an oscillator control signal; A burst gate generator responsive to the horizontal sync pulses supplied from the horizontal sync separator to generate burst gate pulses each delayed by a predetermined time period from a corresponding one of the horizontal sync pulses; Means for separating color burst portions of the first composite video signal responsive to the burst gate pulses; Means for generating the oscillator control signal responsive to the separated color burst portions of the first composite video signal; And means for responding to the sine oscillations for determining the timing of the sampled data generated by the converter for generating the first composite video signal in sampled data form. 3. The apparatus of claim 2, further comprising: a control oscillator for generating sinusoidal oscillations at a frequency controlled by an oscillator control signal; Means for generating the oscillator control signal responsive to horizontal sync pulses supplied from the horizontal sync separator; And means for responding to the sinusoids for determining the timing of sampled data generated by the converter for generating a first composite video signal in sampled data form. 3. The filter of claim 2, wherein the filter for generating the second composite video signal in sampled data form as a response to the first composite video signal in sampled data form is a digital filter, and the sampled data form. And the converter for generating a first composite video signal is an analog / digital converter and generates the second composite video signal in a continuous analog signal form in response to the second composite video signal in the form of sampled data. And said transducer is a digital / analog converter. An image signal processing apparatus comprising: a first sound of vertical synchronization pulses from respective sound detectors for supplying a first sound signal and respective video detectors for supplying a first composite video signal and first composite video signals; A first television receiver front end for receiving a first high-frequency television signal, each vertical synchronous separator for separating a set and each horizontal synchronous separator for separating a first set of horizontal synchronous pulses from a first composite video signal; ; Each vertical detector for separating a second set of vertical sync pulses from each sound detector for supplying a second sound signal and each video detector for supplying a second composite image signal and the second composite image signal, including associated elements. A second television receiver front end for receiving a second high frequency television signal, each horizontal synchronous separator for separating a second set of horizontal synchronous pulses from the synchronous separator and the second composite video signal; A first filter for generating a third composite video signal in response to the first composite video signal, which is a filtering parameter of a filter that can be adjusted according to a weighting factor temporarily stored in an internal register; A second filter for generating a fourth composite video signal in response to the second composite video signal, which is a filtering parameter of a filter that can be adjusted according to a weighting factor temporarily stored in an internal register; Means for storing a ghost-free ghost cancellation reference signal, for generating said weighting coefficient of said first filter responsive to a first ghosted ghost cancellation reference signal correlated with said ghost-free ghost cancellation reference signal; Arranged to generate the weighting coefficient of the second filter responsive to a second ghosted ghost erasing reference signal correlated with the ghost-free ghost erasing reference signal and included in the first and second filters. A filter coefficient computer for generating a weighting coefficient for temporarily storing in the register; Recording electronic elements for receiving the first sound signal and the third composite video signal in sampled data form for recording on a video tape, reproducing a reproduction sound signal and a reproduction composite video signal from the recorded video tape; Video electronic equipment for reproducing and recording capability, wherein said electronic components are provided for reproducing the reproduction sound signal and said reproduction composite video signal for each carrier to generate a third high frequency television signal; Means for selecting one of the accompanying first, second or third high frequency television signals as the first high frequency television signal; A luma / chroma separator responsive to the fourth composite video signal for generating a separated luminance signal and a separated color signal; A chroma demodulation circuit responsive to the separated color signal to generate first and second color difference image signals; A color matrix circuit responsive to said first and second color difference image signals and said separated luminance signal for generating red, green and blue color image signals; A display device for receiving the second set of vertical sync pulses and the second set of horizontal sync pulses, an audio portion receiving a third sound signal, and the red, green and blue image signals for application to the display device A television monitor comprising a video portion for receiving the sound; A first scan line counter that is periodically reset to an initial value of a first scan line count responsive to a vertical sync pulse in the first set, and counts pulses in the first set of horizontal sync pulses to generate a first scan line count Wow; Means for responding to said first scan line count reaching a predetermined value for capturing a current scan line of said first composite video signal as a component of said first ghosted erase reference signal; A second scan line counter that is periodically reset to an initial value of a second scan line count responsive to a vertical sync pulse in the second set, and counts pulses in the second set of horizontal sync pulses to generate a second scan line count Wow; And means for responding to said second scan line count reaching a predetermined value for capturing a current scan line of said second composite video signal as a component of said second ghosted ghost cancellation reference signal. An image signal processing apparatus comprising: a converter and a high frequency amplifier responsive to a first high frequency television signal for generating a first intermediate frequency signal, wherein the intermediate frequency amplifier and one of the first intermediate frequency signal or the second intermediate frequency signal Means for selecting as an input signal to the intermediate frequency amplifier and each video detector including the intermediate frequency amplifier, comprising elements, each sound detector for supplying a first sound signal and the first composite video signal And each vertical synchronous separator for separating the first set of vertical synchronous pulses from the first composite video signal and each horizontal synchronous separator for separating the first set of horizontal synchronous pulses from the first composite video signal. A first television receiver front end for receiving a high frequency television signal; Each of the sound detectors for supplying a second sound signal and each video detector for supplying a second composite video signal and each vertical for separating a second set of vertical sync pulses from the second composite video signal. Each horizontal synchronous separator for separating a second set of horizontal synchronous pulses from the synchronous separator and the second composite video signal, the second high frequency television signal carrying the first high frequency television signal and the first high frequency television signal. A second television receiver front end for receiving; A first filter for generating a third composite video signal in response to the first composite video signal, which is a filtering parameter of a filter that can be adjusted according to a weighting factor temporarily stored in an internal register; A second filter for generating a fourth composite video signal in response to the second composite video signal, which is a filtering parameter of a filter that can be adjusted according to a weighting factor temporarily stored in a register included therein; Means for storing a ghost-free ghost cancellation reference signal, for generating said weighting coefficient of said first filter responsive to a first ghosted ghost cancellation reference signal correlated with said ghost-free ghost cancellation reference signal; Arranged to generate the weighting coefficient of the second filter responsive to a second ghosted ghost cancellation reference signal correlated with the ghost-free ghost cancellation reference signal and included in the first and second filters. A filter coefficient computer for generating a weighting coefficient for temporarily storing in the register; Recording electronic elements for receiving the first sound signal and the third composite video signal in sampled data form for recording on a video tape, reproducing a reproduction sound signal and a reproduction composite video signal from the recorded video tape; A video tape device having reproducing and reproducing capability, comprising: reproducing electronic elements for modulating the reproducing sound signal and the reproducing composite video signal on each carrier to generate the second intermediate frequency signal; A luma / chroma separator responsive to the fourth composite video signal for generating a separated luminance signal and a separated color signal; A chroma demodulation circuit responsive to the separated color signal to generate first and second color difference image signals; A color matrix circuit responsive to said first and second color difference image signals and said separated luminance signal for generating red, green and blue color image signals; A display device for receiving the second set of vertical sync pulses and the second set of horizontal sync pulses, an audio portion receiving a third sound signal, and the red, green and blue image signals for application to the display device A television monitor comprising a video portion for receiving the sound; A first scan line counter that is periodically reset to an initial value of a first scan line count responsive to a vertical sync pulse in the first set, and counts pulses in the first set of horizontal sync pulses to generate a first scan line count Wow; Means for responding to said first scan line count reaching a predetermined value for capturing a current scan line of said first composite video signal as a component of said first ghosted erase reference signal; A second scan line counter that is periodically reset to an initial value of a second scan line count responsive to a vertical sync pulse in the second set, and counts pulses in the second set of horizontal sync pulses to generate a second scan line count Wow; And means for responding to said second scan line count reaching a predetermined value for capturing a current scan line of said second composite video signal as a component of said second ghosted ghost cancellation reference signal. An image signal processing apparatus comprising: related elements, each sound detector for supplying a first sound signal, each video detector for supplying a first composite video signal, and one of the first composite video signal or a third composite video signal Means for selecting a second composite video signal corresponding to one of the plurality of vertical sync pulses for separating a first set of vertical sync pulses from the second composite video signal, and a horizontal sync pulse from the second composite video signal. A first television receiver front end for receiving a first high frequency television signal, each horizontal synchronous separator for separating a first set; Each vertical detector for separating a second set of vertical sync pulses from each sound detector for supplying a second sound signal and each video detector for supplying a fourth composite video signal and the fourth composite video signal, including associated elements. A second high frequency television signal carrying the first high frequency television signal and the first high frequency television signal, each horizontal synchronization separator for separating a second set of horizontal sync pulses from the synchronous separator and the fourth composite video signal. A second television receiver front end for receiving a; A first filter for generating a fifth composite video signal in response to the first composite video signal, which is a filtering parameter of a filter that can be adjusted according to a weighting factor temporarily stored in an internal register; A second filter for generating a sixth composite video signal in response to the fourth composite video signal, which is a filtering parameter of a filter that can be adjusted according to a weighting factor temporarily stored in an internal register; Means for storing a ghost-free ghost cancellation reference signal, for generating said weighting coefficient of said first filter responsive to a first ghosted ghost cancellation reference signal correlated with said ghost-free ghost cancellation reference signal; Arranged to generate the weighting coefficient of the second filter responsive to a second ghosted ghost cancellation reference signal correlated with the ghost-free ghost cancellation reference signal and included in the first and second filters. A filter coefficient computer for generating a weighting coefficient for temporarily storing in the register; Recording electronic elements for receiving the first sound signal and the fifth composite video signal in the form of sampled data for recording on a video tape, the reproduction sound signal and the third composite video signal being a third sound signal; A video tape device including a reproduction electronic element for reproducing a phosphorus reproduced composite video signal from a recorded video tape, the recording and reproducing capability having a recording and reproducing capability; Means for selecting one of the second or third sound signals as a fourth sound signal; A luma / chroma separator responsive to the sixth composite video signal for generating a separated luminance signal and a separated color signal; A chroma demodulation circuit responsive to the separated color signal to generate first and second color difference image signals; A color matrix circuit responsive to said first and second color difference image signals and said separated luminance signal for generating red, green and blue color image signals; A display device receiving said second set of vertical sync pulses and said second set of horizontal sync pulses, an audio portion receiving said fourth sound signal, and said red, green and blue color image for application to said display device A television monitor comprising a video portion for receiving signals; A first scan line counter periodically for initializing a first scan line count responsive to a vertical sync pulse in said first set, and counting pulses in said first set of horizontal sync pulses to generate a first scan line count Wow; Means for responding to said first scan line count reaching a predetermined value for capturing a current scan line of said first composite video signal as a component of said first ghosted erase reference signal; A second scan line counter for periodically resetting to an initial value of a second scan line count responsive to a vertical sync pulse in said second set, and counting pulses in said second set of horizontal sync pulses to generate a second scan line count Wow; And means for responding to said second scan line count reaching a predetermined value for capturing a current scan line of said second composite video signal as a component of said second ghosted ghost cancellation reference signal.