RU2132110C1 - Method for optimal vector control of induction electric motor and electric drive which implements said method - Google Patents

Abstract

FIELD: electrical engineering, in particular, electric drives of petroleum bore pumps, electric motor cars, machine-tool construction, chemically aggressive environment. SUBSTANCE: method for vector regulation of in-phase and orthophase currents U_sα,U_sβ in function of measured two-phase stator voltage

involves calculation phase φ_s and rotor interlinkage amplitude φ_r with respect to measured stator voltage levels U_sα, U_sβ and current levels i_sα, i_sβ, and alteration of given in-phase current amplitude I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ in proportion to orthophase current amplitude I_ort which is measured with respect to calculated rotor interlinkage phase φ_s(U_sα, U_sβ, i_sα, i_sβ, t),, regulation of slip frequency in proportion to alteration of measured orthophase current amplitude I_ort and calculated rotor interlinkage amplitude φ_r(U_sα, U_sβ, i_sα, i_sβ, t).. Corresponding electric drive has phase voltage detector 32, which measures instantaneous value of rotor interlinkage phase which provides reference for shifting of stator current vector

by optimal angle due to conversion of measured orthophase current amplitude I_ort by unit 32 into optimal given in-phase current level I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ . Inputs of slip frequency calculation unit 33 are connected to output of reverse Cartesian coordinates converter 30 and output of rotor interlinkage amplitude calculation unit. EFFECT: minimal losses of electric power upon regulation of momentum and speed without speed detector. 3 cl, 12 dwg

Description

 The invention relates to electrical engineering, in particular to variable AC electric drives, and can be used to minimize energy losses when feeding an induction motor from a converter, as well as to regulate the moment and speed of induction motors, including for machines and mechanisms equipped with asynchronous motors without sensors on the motor shaft: in oil production for rocking machines and deep submersible electric pumps of oil wells, for pumps and fans, To EVs electrospindle for machine tools, in weaving looms and in chemically active environments.

 There is a method of vector control of an induction motor, in which the stator windings of the induction motor are supplied with alternating current equal to the difference of the stator currents synchronously changed with the synchronization frequency of the cosine (in-phase) and sinusoidal (orthophase) currents, the amplitudes of which vary depending on a given moment from the phase shift of the current vector stator relative to the synchronization phase by an angle equal to the arc tangent of the ratio of the amplitudes of the common-mode and orthophase current [1] (RF Patent N 1458951, V.A Mishchenko, N.M. Mishchenko, priority 03/26/1984).

 The electric drive for implementing this method contains an inverter, the power outputs of which are connected via phase current sensors to the stator windings of the asynchronous electric motor, and the control inputs are connected to the outputs of the direct converter of two-phase-three-phase coordinates through the unit for generating control pulses and the block of PWM current regulators connected to the phase current sensors the outputs of which are connected to the outputs of the direct Cartesian coordinates transformer [1].

 The disadvantages of this technical solution are high losses of electricity in an induction motor, low accuracy of torque and speed control and a small range of regulation of an induction motor without the use of sensors on the motor, which does not allow the effective use of torque and speed control for many machines and mechanisms in which it is difficult or impossible the use of sensors structurally associated with an induction motor (for example, in the oil and chemical industries sti).

 The closest (prototype) is a technical solution that represents a method of optimal vector control of an induction motor, in which the stator windings of the asynchronous motor are supplied with alternating current equal to the difference of the stator currents synchronously changed with the synchronization frequency, the sinusoidal (in-phase) and sinusoidal (orthophase) currents, whose amplitudes vary in depending on a given moment from the condition of the optimum phase angle angle of the stator current vector relative to the flow vector rotor Lenia preferably at a level close to ± π / 4 [2] (RF Patent N 1515322, VA Mishenko, priority of 11.05.1984).

 The electric drive for implementing this optimal vector control method comprises an inverter, the power outputs of which are connected to the stator windings of the induction motor through phase current sensors, and the control inputs are connected to the outputs of the direct converter two-phase by means of a control pulse generation unit and a block of PWM current regulators connected to the phase current sensors -three-phase coordinates, the inputs of which are connected to the outputs of the direct converter of Cartesian coordinates, while the orthophase and in-phase the inputs of the direct Cartesian coordinates converter are connected to the outputs of the orthophase current controller and the common-mode current controller, respectively, the input of the orthophase current controller is connected to the output of the adaptive torque controller, the input of the torque reference of which is connected to the output of the speed controller, the input connected to the speed reference unit [2].

 The disadvantages of this technical solution are the impossibility of minimizing energy losses and regulating the moment and speed of an induction motor without a speed sensor on the motor shaft, which is caused by the dependence of the parameters of the stator current vector on the measured value of the rotation speed obtained at the output of the speed sensor on the electric motor.

 The disadvantage is the limited use of the electric drive in the oil, chemical industry, in electric vehicles, in electric spindles, where, under various operating conditions, it is impossible or difficult to use any sensors structurally associated with the electric motor. For example, a deep-well submersible asynchronous electric motor of a centrifugal electric pump for oil production is located in an oil well at a depth of 2–3 km and is powered by a special three-phase submersible power cable from the ground frequency converter, which cannot transmit information from a deep-seated submersible speed sensor and, therefore, it is impossible to use the optimal vector control of an induction motor according to the technical solution according to the prototype [2].

 The task of minimizing the loss of electricity in an induction motor due to the optimal vector control of an induction motor without sensors on the motor, which is first posed in the present invention, is new, relevant and highly efficient for the electric power industry, since half of the electricity produced in the world is consumed by asynchronous motors without sensors on the electric motor and the created standard solid-state power modules of frequency converters become commensurable in cost, overall am and reliability with protective shut-off equipment, universally used for connecting asynchronous electric motors to an industrial network in quantities of 50-70 million units.

 The aim of the invention is to minimize the loss of electricity in an induction motor without the use of sensors on the motor, improving the accuracy of optimal control modes taking into account the saturation of the magnetic circuit of the motor while expanding the range of torque and speed control.

 The solution of this problem, the invention is aimed at, and the goal of the invention is achieved by a technical solution, namely, in the method of optimal vector control of an asynchronous electric motor, in which the stator windings of the asynchronous electric motor are supplied with alternating current equal to the difference synchronously changed with the synchronization frequency of the cosine ( in-phase) and sinusoidal (orthophase) stator currents, the amplitudes of which vary depending on a given moment from The optimality of the angle of the phase shift of the stator current vector relative to the rotor flux vector mainly at a level close to ± π / 4, the two-phase stator voltage and the two-phase stator current are additionally measured, then the rotor flux link amplitude is calculated by integrating the difference of the measured voltages and currents, in addition, the magnitude sine and cosine functions of the rotor flux linkage phase, proportionally to which the sine (orthophase) and cosine (in-phase) stator currents are regulated respectively introducing predetermined and measured relative to the indicated sine and cosine functions of the rotor flux linkage phase of the stator orthophase and in-phase current amplitudes, the predetermined common-mode current amplitude is proportional to the measured orthophase current amplitude, the value of which is set proportionally to the ratio of the given moment to the calculated rotor flux-coupling amplitude, the slip frequency is regulated proportionally the ratio of the measured amplitude of the orthophase current and the amplitude of the flux linkage of the rotor, and the rotation speed I regulate by comparing the set rotation speed with the current rotation speed, which is calculated by summing the slip frequency with the synchronization frequency equal to the frequency of the sine and cosine functions of the rotor flux linkage phase.

 To implement the method of optimal vector control in an electric drive containing an inverter, the power outputs of which are connected via phase current sensors to the stator windings of an asynchronous electric motor, and the control inputs are connected to the outputs of the direct converter via a block of control pulse generation and phase sensors connected to the phase current sensors two-phase-three-phase coordinates, the inputs of which are connected to the outputs of the direct converter of Cartesian coordinates, while orthophase and in-phase the inputs of the direct Cartesian coordinates converter are connected to the outputs of the orthophase current controller, the input of the orthophase current controller is connected to the output of the adaptive torque controller, the input of the torque reference of which is connected to the output of the speed controller, the input of which is connected to the speed reference unit, in addition to the power outputs of the inverter phase voltages, two outputs of which are directly and through an adder of phase voltages connected to the inputs of two adders, the second inputs two adders are connected to the outputs of the inverter of the two-phase-three-phase coordinates, inputs connected to the phase current sensors, the outputs of two adders are connected to the inputs of two integrators, the outputs are connected to the inputs of two other adders, the second inputs of which are connected to the outputs of the inverter of the two-phase three-phase coordinates, the outputs adders are connected to the first inputs of two dividers and at the same time to two inputs of the rotor flux link amplitude calculation unit, the output of which is connected to The combined second inputs of the dividers, the outputs of the dividers are connected to the sine and cosine inputs of the forward and reverse Cartesian coordinates, the two inputs of the inverse of the Cartesian coordinates are connected to the outputs of the inverter of the two-phase-three-phase coordinates, the orthophase and common-mode inputs of the inverse of the Cartesian coordinates are connected to the feedback inputs, respectively orthophase and common-mode current regulators, in addition, the orthophase output of the inverse converter of the Cartesian coor inat is connected through the module selection unit to the input of the optimal common-mode current control unit, the output is connected to the master input of the common-mode current controller, the second input of the common-mode current control unit is connected to the second output of the adaptive torque controller, while the orthophase output of the inverse Cartesian coordinate converter is connected to the first input a slip frequency calculation unit, the second input of which is connected to the output of the rotor flux linkage amplitude calculation unit, a sine and cosine input The forward and reverse converters of the Cartesian coordinates are connected to the inputs of the synchronization frequency calculation unit, the output connected to the first input of the frequency adder, the second input of the frequency adder is connected to the output of the slip frequency calculation unit, the output of the frequency adder is connected to the feedback input of the speed controller and simultaneously to the second input adaptive torque controller, the third input of which is connected to the output of the rotor flux link amplitude calculation unit.

 In addition, a digital signal coprocessor, a digital signal processor, a programmable read-only memory, a serial port, a program control and diagnostics panel are introduced into the microprocessor-based electric drive, while the control pulse generating unit on the control inputs of the inverter is made in the form of a driver chip with an additional PWM resolution input, the driver microcircuit inputs are connected to the outputs of the digital signal coprocessor formed by the outputs of the three-phase time block pulse-width modulation of a digital signal coprocessor, the analog inputs of which are connected to the outputs of the phase voltage sensor and the phase current sensor, are formed by the inputs of the integrated multiplex analog-to-digital converter of the digital signal coprocessor, which also contains an integrated serial command programmer, an integrated vector conversion unit, a block register control, interconnected and with the specified built-in multiplex analog-to-digital converter and th phase timer pulse width modulation data bus and address bus, which are coupled to the digital signal processor, a programmable read only memory, a serial port and the software control and diagnosis unit.

 The proposed technical solution is illustrated by the figures of the drawings and diagrams (Fig. 1 - 7), which depict: in Fig. 1 is a diagram of an electric drive with optimal vector control; in FIG. 2 is a diagram of a microprocessor electric drive with optimal vector control; in FIG. 3 - vector diagrams of optimal vector control; in FIG. 4 - graphs of stator current changes in static modes and current minimum mode; in FIG. 5 are graphs of changes in rotor sweating and in-phase and orthophase stator current amplitudes with optimal vector control in static drive modes; in FIG. 6 - graphs of the change in torque in dynamic modes with a limited stator current and the optimal mode of maximum torque; in FIG. 7 - oscillograms of the dynamic process of optimal vector control of the motor moment at start-up.

The electric drive according to the circuit of FIG. 1 contains an inverter 1 (Fig. 1), the power inputs of which are connected through phase sensors 2, 3 to the stator windings of an asynchronous electric motor 4. The control inputs of inverter 1 through a block 5 generating control pulses and connected to the sensors 2, 3 of the phase current PWM block 6 - current regulators are connected to the outputs of the direct converter of two-phase-three-phase coordinates. Inputs i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ a direct transducer 7 of two-phase-three-phase coordinates are connected to the outputs of a direct transducer 8 of Cartesian coordinates, the orthophase and common mode inputs of which are connected to the outputs of the orthophase current controller 9 and the common mode controller 10, respectively. Input I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ the orthophase current regulator 9 is connected to the output of the adaptive moment regulator 11, the moment input M ^{* of} which is connected to the output of the speed regulator 12. The input input ω ^{* of the} speed controller 12 is connected to the speed setting unit 13.

In addition, to the power outputs of the inverter 1 is connected to the sensor 14 phase voltage with two outputs U _sa , U _sb connected to the inputs of the adder 15 phase voltage.

The combination of the sensor 14 phase voltage and the adder 15 phase voltage forms a two-phase voltage sensor u _sα , u _sβ with a phase shift of u _sβ relative to u _sα by an angle of ± 90 ^o . The first output U _sa of the phase voltage sensor 15, forming the first output u _sα of the two-phase voltage sensor, is connected to the input of the first adder 16. The output of the adder 15 phase voltages, forming the second output u _sβ of the two-phase voltage sensor u _sα , u _sβ , is connected to the input of the second adder 17. The second inputs of the first and second adders 16, 17 are connected respectively to the first output i _sα and the second output i _sβ of the inverter 18 of two-phase-three-phase coordinates, two inputs of which are connected to the outputs i _sa , i _{sb of the} sensors 2, 3 of the phase current. The outputs of the sensors 2, 3 of the phase current i _sa , i _{sb are} connected to the inputs of the adder 19 phase currents and are simultaneously connected to two feedback inputs of the block 6 of the PWM current controllers. The third input i _sc feedback block 6 of the PWM current controllers is connected to the output of the adder 19 phase currents. The outputs of two adders 16, 17 are connected to the inputs of two integrators 20, 21, the outputs are connected to the inputs of two other adders 22, 23, the second inputs of which are connected to the outputs i _sα , i _sβ of the inverter 18 of two-phase-three-phase coordinates. The outputs of the two second adders 22, 23 are connected to the first inputs of the two dividers 24, 25 and are simultaneously connected to the two inputs of the rotor flux link amplitude calculation unit 26, which contains two multipliers 27, 28 at the input and a quadrature computer 29 at the output. The outputs of the multipliers 27, 28 connected to the inputs of the quadrature computer 29, the output of which forms the output of the rotor flux link amplitude calculation unit 26, connected to the second combined inputs of the dividers 24, 25. The outputs sin, cos of the dividers 24, 25 are connected respectively to the sine and braids through the inputs of the forward and reverse transducers 8, 30 Cartesian coordinates. Two inputs i _sα , i _{sβ of the} inverse transducer 30 of Cartesian coordinates are connected to the outputs i _sα , i _{sβ of the} inverse transducer 18 of two-phase-three-phase coordinates. The orthophase and common mode outputs I _ort , I _{syn of the} inverse transformer of the Cartesian coordinates are connected to the feedback inputs of the orthophase current regulator 9 and the common mode current regulator 10, respectively. In addition, the orthophase output I _{ort of the} inverse transformer of the Cartesian coordinates 30 is connected via the module selection unit 31 to the input of the optimal common-mode current control unit 32. The output of the block 32 of the optimal common-mode current control is connected to the reference input I _{syn of the} in-phase current controller 10. The second input of the block 31 of the optimal common-mode current control is connected to the output of the adaptive regulator 11 of the moment. The orthophase output I _{ort of the} Cartesian coordinate inverter 30 is also connected to the first input of the slip frequency calculation unit 33, the second input of which is connected to the output of the rotor flux link amplitude calculation unit 26. The sine and cosine inputs sin, cos of the direct and inverse transducers 8, 30 of the Cartesian coordinates are connected to the inputs of the synchronization frequency calculation unit 34. The output of the synchronization frequency calculation unit 34 is connected to the first input of the frequency adder 35, the second input of the frequency adder 35 is connected to the output of the slip frequency calculator 33. The output of the adder 35 frequencies is connected to the feedback input of the speed controller 12 and to the second input of the adaptive torque controller 11. The third input of the adaptive moment controller 11 is connected to the output of the rotor flux link amplitude calculation unit 26.

The electric drive according to the circuit of FIG. 1 works as follows. The inverter 1 through the sensors 2, 3 phase currents supplies the stator windings of the induction motor 4 with pulse-width modulated power voltage pulses, the duration of which is determined by the control pulses coming from the output of the pulse forming unit 5. Block 6 PWM current controllers controls the duration of the control pulses by pulse-width modulation of the mismatch of the measured phase currents i _sa , i _sb , i _sc and the given phase currents i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sa}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sb}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sc}}$ which are formed by a direct converter of two-phase-three-phase coordinates depending on the input values of the two-phase stator current setting i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ shifted relative to each other by a phase angle of 90 ^o . Setting two-phase stator current i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ is formed at the output of the direct transformer 8 Cartesian coordinates using the orthophase current regulator 9 and the in-phase current regulator 10. To the control input of the regulator 9 orthophase current receives the orthophase current I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ from the output of the adaptive regulator 11 of the moment, the input value of the job moment M ^* for which is formed at the output of the speed controller 12. The speed controller 12 carries out astatic regulation of the rotor speed of the induction motor according to the proportional-integral law depending on the mismatch of the given rotation speed ω ^* and the calculated current rotation speed ω. The rotation speed ω ^* is set using the speed setting unit 13.

The pulse-width modulated power phase voltages U _sa , U _sb , U _sc generated at the power outputs of the inverter 1 as a result of pulse-width modulation of the mismatches of the set and measured phase currents are measured at the output of the inverter 1 using the sensor 14 phase voltage. At the outputs of the sensor 14 phase voltages, signals of the measured phase voltages U _sa , U _sb are generated, which are fed to the adder 15 phase voltages. Phase "a" of inverter 1 is used as the reference phase of symmetric three-phase voltage systems U _sa , U _sb , U _sc , and currents i _sa , i _sb , i _sc , which is realized directly by supplying a signal of the measured phase voltage U _{sa of} phase "a" of inverter 1 from the output of the sensor 14 phase voltages to the input u _{sα of the} adder 16. At the output of the adder 15 phase voltages, a signal of the measured voltage u _{sβ is formed} , lagging behind or ahead of the voltage u _sα = U _sa by a phase angle of ± 90 ^o . The signal of the measured voltage u _{sβ is} fed to the input of the adder 17. The adders 16, 17 form at the outputs two differences of the measured two-phase voltages u _sα , U _sβ and the corresponding measured two-phase currents i _sα , i _sβ .
The signals of the measured two-phase currents i _sα , i _sβ are generated at the two outputs i _sα , i _sβ of the inverter 18 of two-phase-three-phase coordinates, converting the three-phase system of measured currents i _sa , i _sb , i _sc with a phase shift of 120 ^o into a two-phase system of measured currents i _sα , i _sβ with a phase shift of ± 90 ^o . For this, the signals of the measured phase currents i _sa , i _sb from the outputs of the sensors 2, 3 of the phase current are supplied to the two inputs of the inverter 18 of the two-phase-three-phase coordinates. To isolate the feedback signal (block 6 PWM current regulators) by phase current i _{sc the} signals of two measured phase currents i _sa , i _sb from the outputs of sensors 2, 3 of the phase current are fed to the inputs of the adder 19 phase currents, the output signal of which is fed to the input channel "c" block 6 PWM current regulators.

асинхронного электродвигателя 4. С помощью умножителей 27, 28, которые умножают входные величины ψ_rα,ψ_rβ на эти же входные величины ψ_rα,ψ_rβ, производится вычисление квадратов величин ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rα}}$ ,ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rβ}}$ , преобразуемых в блоке 26 вычисления амплитуды потокосцепления ротора с помощью операции извлечения корня квадратного из суммы квадратов ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rα}}$ ,ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rβ}}$ . В результате выполнения указанной вычислительной операции на выоде блока 26 вычисления амплитуды потокосцепления ротора образуется сигнал амплитуды потокосцепления ротора ψ_r, соответствующий модулю вектора потокосцепления ротора

Этот сигнал косвенного измерения амплитуды потокосцепления ротора асинхронного электродвигателя 4 образован из сигналов измеренных фазных токов и фазных напряжений на выходе инвертора 1 без применения датчиков, конструктивно связанных с асинхронным электродвигателем 4. При этом нелинейные процессы магнитного насыщения магнитопровода асинхронного электродвигателя 4, пространственного и зубцовых искажений формы распределения магнитного потока в асинхронном двигателе интегрированно с наибольшей точностью по сравнению с прямым измерением ψ_r отражаются на взаимном изменении фазных напряжений и фазных токов, а следовательно, на полученном на выходе блока 26 сигнале амплитуды потокосцепления ротора ψ_r.
Сигнал вычисленной (косвенно измеренной) амплитуды потокосцепления ротора ψ_r с выхода блока 26 поступает на объединенные вторые входы делителей 24, 25. В результате деления сигналов ψ_rα,ψ_rβ на величину амплитуды потокосцепления ротора ψ_r на выходах делителей 24, 25 образуются нормированные (с единичной амплитудой) гармонические сигналы cosφ_s,sinφ_s, характеризующие функции фазы φ_s вектора потокосцепления ротора ψ_r относительно неподвижной опорной оси "α", совпадающей с осью опорной фазы "a" статора асинхронного электродвигателя 4. Управление углом фазового сдвига ε_φ вектора тока статора

относительно вектора потокосцепления ротора

осуществляется путем синхронизации управления и измерения вектора тока

в координатах вектора потокосцепления ротора

за счет подачи одновременно на синусные и косинусные входы прямого и обратного преобразователей 8, 30 декартовых координат нормированных функций фазы потокосцепления ротора 1•sinφ_s,1•cosφ_s, полученных с выходов делителей 24, 25.The integrators 20, 21 perform the operation of integrating the differences of the measured voltages u _sα , u _sβ and the measured currents i _sα , i _sβ . The signals obtained at the outputs of the integrators 20, 21 are fed to the inputs of two other adders 22, 23, to the second inputs of which the output signals i _sα , i _{sβ of the} inverse transducer 18 of two-phase-three-phase coordinates are supplied. From the outputs of the adders 22, 23, the signals ψ _rα , ψ _rβ , proportional to the flux linkage of the rotor, are fed to the first inputs of two dividers 24, 25 and to the two inputs of the rotor flux link amplitude calculation unit 26. Received at the outputs of adders 22, 23, the signals of the two-phase rotor flux _clamp ψ _rα , ψ _{rβ are} shifted relative to each other by a phase angle of ± 90 ^o and represent the quadrature components of the rotor flux link vector

asynchronous electric motor 4. Using multipliers 27, 28, which multiply the input quantities ψ _rα , ψ _rβ by the same input values ψ _rα , ψ _rβ , the squares of the quantities ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rα}}$ , ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rβ}}$ transformed in block 26 calculating the amplitude of the flux linkage of the rotor using the operation of extracting the square root of the sum of squares ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rα}}$ , ψ $\genfrac{}{}{0ex}{}{\text{2}}{\text{rβ}}$ . As a result of performing the indicated computational operation, at the output of the rotor flux link amplitude calculation unit 26, a rotor flux link amplitude signal ψ _r corresponding to the rotor flux link vector module

This indirect signal measuring the amplitude of the flux linkage of the rotor of the induction motor 4 is formed from the signals of the measured phase currents and phase voltages at the output of the inverter 1 without the use of sensors structurally associated with the asynchronous motor 4. In this case, the nonlinear processes of magnetic saturation of the magnetic circuit of the asynchronous electric motor 4, spatial and gear distortions magnetic flux distribution in an induction motor integrated with the greatest accuracy compared to direct measurement ψ _r are reflected in the mutual change of phase voltages and phase currents, and therefore, on the signal received at the output of block 26 of the amplitude of the flux linkage of the rotor ψ _r .
The signal of the calculated (indirectly measured) amplitude of the rotor flux linkage ψ _r from the output of block 26 goes to the combined second inputs of the dividers 24, 25. As a result of dividing the signals ψ _rα , ψ _rβ by the magnitude of the rotor flux coupling amplitude ψ _r , the normalized ones are formed at the outputs of the dividers 24, 25 ( with unit amplitude) harmonic signals cosφ _s, sinφ _s, characterizing the phase function φ _s of the rotor flux linkage ψ _r relative to the fixed reference axis "α", which coincides with the axis of the phase reference "a" of the stator of the induction motor 4. control y th phase shift ε _φ stator current vector

relative to the rotor flux linkage vector

carried out by synchronizing control and measuring the current vector

in the coordinates of the rotor flux link vector

by supplying simultaneously the sine and cosine inputs of the direct and inverse converters 8, 30 Cartesian coordinates of the normalized functions of the rotor flux coupling phase 1 • sinφ _s , 1 • cosφ _s obtained from the outputs of the dividers 24, 25.

в неподвижных декартовых координатах α,β, одна из осей которых (ось "α" ) совпадает с осью фазы "a" статорной обмотки трехфазного асинхронного электродвигателя 4. С помощью полученных нормированных функций 1•sinφ_s,1•cosφ_s производится синхронизация измерения двухфазного тока i_sα,i_sβ относительно гармонических функций sinφ_s,cosφ_s фазы потокосцепления ротора.The signals of the measured two-phase current i _sα , i _sβ supplied to the two inputs of the inverse transformer 30 Cartesian coordinates are projections of the stator current vector

in fixed Cartesian coordinates α, β, one of whose axes (axis “α”) coincides with the phase axis “a” of the stator winding of a three-phase asynchronous electric motor 4. Using the obtained normalized functions 1 • sinφ _s , 1 • cosφ _s , the two-phase measurement is synchronized current i _sα , i _sβ with respect to harmonic functions sinφ _s , cosφ _{s of} the rotor flux linkage phase.

во вращающейся декартовой системе координат x, y синхронизируются по управлению путем векторного преобразования входных проекций x, y относительно фазы φ_s вектора потокосцепления ротора ψ_r с помощью полученных нормированных функций 1•sinφ_s,1•cosφ_s, подаваемых на синусный и косинусный входы прямого преобразователя 8 декартовых координат.Similarly, the signals for setting the projections of the stator current vector coming to the two inputs of a direct transformer of 8 Cartesian coordinates

in a rotating Cartesian coordinate system x, y are synchronized by control by vector transformation of input projections x, y relative to phase φ _s of the rotor flux link vector ψ _r using the obtained normalized functions 1 • sinφ _s , 1 • cosφ _s applied to the sine and cosine inputs of the direct transducer 8 Cartesian coordinates.

в координатах вектора потокосцепления ротора

а на выходах обратного преобразователя 30 декартовых координат образуются сигналы проекций i_sx = I_syn, i_sy = I_ort вектора тока статора

в координатах вектора потокосцепления ротора

в зависимости от входных сигналов измеренного двухфазного тока i_sα,i_sβ. При этом фаза φ_s вектора потокосцепления ротора

косвенно измеряется по измеренным фазным токам и напряжениям с помощью измерительного преобразователя, образуемого совокупностью блоков 16, 17, 20, 21, 22, 23, 24, 25, 26, относительно неподвижной оси "α" опорной фазы "a" статорной обмотки асинхронного электродвигателя 4.As a result of vector transformations into the functions sinφ _s , cosφ _s , the signals of a given two-phase current i are generated at the output of the direct transducer 8 of Cartesian coordinates $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ defined by the input projections i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sx}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sy}}$ stator current vectors

in the coordinates of the rotor flux link vector

and at the outputs of the inverse transformer 30 Cartesian coordinates, projection signals i _sx = I _syn , i _sy = I _ort of the stator current vector

in the coordinates of the rotor flux link vector

depending on the input signals of the measured two-phase current i _sα , i _sβ . The phase φ _s of the rotor flux linkage vector

indirectly measured by the measured phase currents and voltages using a measuring transducer formed by a set of blocks 16, 17, 20, 21, 22, 23, 24, 25, 26, relative to the fixed axis "α" of the reference phase "a" of the stator winding of the induction motor 4 .

на ось x, совпадающую с осью вектора потокосцепления ротора, пропорционален косинусной функции фазы φ_s потокосцепления ротора и совпадает по фазе с вектором потокосцепления ротора, в связи с чем образуется синфазный (косинусоидальный) ток статора, например, для фазы "a" i_{syn a} = I_syn • cosφ_s.
На втором выходе обратного преобразователя декартовых координат образуется сигнал i_sy = I_ort проекции вектора тока статора

ортогональной по отношению к вектору потокосцепления ротора

Проекция i_sy характеризует ток статора, опережающий по фазе в двигательном режиме и отстающий по фазе в режиме торможения на угол ± 90^o относительно фазы φ_s потокосцепления ротора ψ_r, в связи с чем образует ортофазный (синусоидальный) ток статора, например, для фазы "a" i_{ort a} = ± I_ort • sinφ_s. Результирующий ток статора асинхронного электродвигателя 4 регулируется равным разности синфазного (косинусоидального) и ортофазного (синусоидального) токов статора, например, для фазы "a" i_sa = i_syn - i_ort.The projection signal i _sx of the stator current vector received from the first output of the inverse transformer 30 Cartesian coordinates

on the x axis, which coincides with the axis of the rotor flux linkage vector, is proportional to the cosine function of the rotor flux linkage phase φ _s and coincides in phase with the rotor flux linkage vector, and therefore the in-phase (cosine) stator current is generated, for example, for phase "a" i _{syn a} = I _syn • cosφ _s .
At the second output of the inverse Cartesian coordinate converter, a signal i _sy = I _{ort of the} projection of the stator current vector is formed

orthogonal to the rotor flux linkage vector

The projection i _sy characterizes the stator current, outstripping in phase in the motor mode and lagging in phase in braking mode by an angle of ± 90 ^o relative to the phase φ _{s of the} rotor flux linkage ψ _r , and therefore forms an orthophasic (sinusoidal) stator current, for example, for phase "a" i _{ort a} = ± I _ort • sinφ _s . The resulting stator current of the asynchronous electric motor 4 is regulated equal to the difference in-phase (cosine) and orthophase (sinusoidal) stator currents, for example, for phase "a" i _sa = i _syn - i _ort .

Thus, the signals at the outputs of the inverter 30 Cartesian coordinates characterize the amplitude of the common-mode current I _syn and the orthophase current I _ort .

на вход блока 32 оптимального управления синфазным током. В динамических и статических режимах изменения измеренной амплитуды ортофазного тока I_ort приводит к оптимальному изменению амплитуды синфазного тока I_syn за счет изменения выходного сигнала блока 32 оптимального управления синафазным током, поступающего на задающий вход регулятора 10 синфазного тока. Регулятор 10 синфазного тока отслеживает заданный блоком 32 оптимальный по энергетическим критериям режим изменения амплитуды синфазного тока I_syn = F(I_ort) и определяемый изменением амплитуды синфазного тока оптимальный закон изменения амплитуды потокосцепления ротора ψ_r в зависимости от момента M асинхронного электродвигателя.In the dynamic modes of the electric drive, deviations ΔI _ort , ΔI _{sin of the} measured amplitudes I _ort , I _{syn of the} orthophase and in-phase currents obtained at the output of the inverter 30 Cartesian coordinates relative to the given amplitudes I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ , I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ orthophase and common mode currents at the master inputs of regulators 9, 10 orthophase and common mode currents. The astatic regulation of the amplitudes of the orthophase and common-mode currents is carried out by proportional-integral control of the orthophase and common-mode currents due to the arrival of the signals I _ort , I _syn from the outputs of the inverse transformer 30 Cartesian coordinates to the feedback inputs of the regulators 9, 10 of the orthophase and common-mode currents. At the same time, the signal of the measured amplitude of the orthophase current I _ort from the output of the inverse transducer 30 Cartesian coordinates is received through the module selection unit 31

to the input of block 32 of the optimal common-mode current control. In dynamic and static modes, the change in the measured amplitude of the orthophase current I _ort leads to an optimal change in the amplitude of the common-mode current I _syn due to a change in the output signal of the optimal _syn -phase current control unit 32 supplied to the input of the common-mode current controller 10. The common-mode current controller 10 monitors the mode of changing the common-mode current amplitude I _syn = F (I _ort ) specified by block 32 according to the energy criteria and the optimal law of changing the rotor flux coupling amplitude ψ _r determined by the change in common-mode amplitude depending on the moment M of the induction motor.

относительно ротора асинхронного электродвигателя 4 определяется в зависимости от отношения действительной (измеренной) амплитуды ортофазного тока I_ort к действительной (косвенно измеренной) амплитуде потокосцепления ротора ψ_r, что реализуется подачей сигнала измеренной амплитуды ортофазного тока I_ort с выхода обратного преобразователя 30 декартовых координат и сигнала косвенно измеренной амплитуды потокосцепления ротора ψ_r на два входа блока 33 вычисления частоты скольжения.The given moment M ^* , the value of which is generated at the output of the proportional-integral speed controller 12, is converted in the adaptive moment controller 11 into a signal of a given amplitude of the orthophase current I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ , which in dynamics differs from the measured amplitude of the orthophase current I _ort by the mismatch ΔI _ort . The actual value of the slip frequency Δω characterizing the relative angular velocity of rotation of the rotor flux linkage vector

relative to the rotor of the induction motor 4 is determined depending on the ratio of the actual (measured) amplitude of the orthophase current I _ort to the actual (indirectly measured) amplitude of the flux linkage of the rotor ψ _r , which is realized by supplying a signal of the measured amplitude of the orthophase current I _ort from the output of the inverse transformer 30 Cartesian coordinates and signal the indirectly measured amplitude of the flux linkage of the rotor ψ _r to the two inputs of the slip frequency calculation unit 33.

относительно неподвижной оси "α" статора однозначно определяется с помощью блока 34 вычисления частоты синхронизации производной фазы φ_s(t) вектора потокосцепления ротора

полученной на выходах делителей 24, 25 в виде нормированных функций фазы синхронизации φ_s прямого и обратного преобразования декартовых координат и векторного регулирования тока статора

Нормированные гармонические сигналы 1•sinφ_s,1•cosφ_s поступают на входы блока 34 вычисления частоты синхронизации, в котором вычисляется фаза синхронизации φ_s и ее производная

Образуемый на выходе блока 34 вычисления частоты синхронизации сигнал частоты синхронизации ω_s, пропорциональной производной фазы синхронизации, суммируется в сумматоре 35 частот с сигналом частоты скольжения Δω, поступающим с выхода блока 34 вычисления частоты синхронизации. На выходе сумматора 35 частот образуется сигнал, пропорциональный мгновенной текущей величине скорости вращения ротора асинхронного электродвигателя 4. Выходной сигнал сумматора 35 частот поступает на вход обратной связи регулятора 12 скорости и на второй вход адаптивного регулятора 11 момента. Регулятор 12 скорости осуществляет астатическое регулирование скорости вращения асинхронного электродвигателя 4 в пропорционально-интегральной зависимости от динамического рассогласования мгновенных величин заданной скорости вращения ω^* (t) и текущей скорости вращения ω (t), полученной в результате измерения напряжения и тока на выходе инвертора 1 и преобразования сигналов измеренных напряжений и токов с помощью блоков 14 - 34.The instantaneous value and direction of the angular velocity of rotation ω _s of the rotor flux linkage vector

relative to the fixed axis "α" of the stator is uniquely determined using block 34 for calculating the synchronization frequency of the derived phase φ _s (t) of the rotor flux linkage vector

obtained at the outputs of the dividers 24, 25 in the form of normalized functions of the synchronization phase φ _s direct and inverse transformation of the Cartesian coordinates and vector control of the stator current

The normalized harmonic signals 1 • sinφ _s , 1 • cosφ _s are fed to the inputs of the synchronization frequency calculation unit 34, in which the synchronization phase φ _s and its derivative are calculated

Generated at the output of the synchronization frequency calculation unit 34, the signal of the synchronization frequency ω _s proportional to the derivative of the synchronization phase is summed in the frequency adder 35 with the slip frequency signal Δω coming from the output of the synchronization frequency calculation unit 34. The output of the adder 35 frequencies produces a signal proportional to the instantaneous current value of the rotational speed of the rotor of the induction motor 4. The output signal of the adder 35 frequencies is fed to the feedback input of the speed controller 12 and to the second input of the adaptive torque controller 11. The speed controller 12 performs astatic regulation of the rotation speed of the induction motor 4 in proportion to the dynamic mismatch of the instantaneous values of the given rotation speed ω ^* (t) and the current rotation speed ω (t) obtained by measuring the voltage and current at the output of the inverter 1 and converting the signals of the measured voltages and currents using blocks 14 to 34.

относительно вектора потокосцепления ротора

в зависимости от трех входных величин: M^*, ω , ψ_r .As a result of the astatic regulation of the rotation speed at the output of the speed controller 12, a signal of a given moment M ^* is generated, which is necessary for maximum speed of restoring the current rotation speed ω (t) to a value equal to a given rotation speed ω ^* . The signal of a given moment M ^* from the output of the speed controller 12, supplied to the first input of the adaptive regulator 11 of the moment, is converted in the adaptive controller 11 of the moment into a signal of a given amplitude of the orthophase current as a result of dividing the input value of a given moment M ^* by the value of the indirectly measured amplitude of the rotor flux linkage ψ _r the signal of which is supplied to the second input of the adaptive moment controller 11 from the output of the rotor flux link amplitude calculation unit 29. The third input of the adaptive moment controller 11 receives a signal of the current rotation speed ω from the output of the adder 35 frequencies. The adaptive moment controller 11 performs the operation of switching the optimal laws of regulation of the amplitudes I _ort , I _{syn of the} orthophase and in-phase currents and the angle ε _{φ of the} phase shift of the stator current vector

relative to the rotor flux linkage vector

depending on three input quantities: M ^* , ω, ψ _r .

Three-zone optimal vector control of an induction motor to minimize energy losses, maximum speed and maximum ratio of moment M to current i _s in static and dynamic modes of operation of the electric drive is carried out using block 32 for optimal common-mode current control and adaptive torque regulator 11.

In the first regulation zone, for small values of the given moment M ^* and the actual moment M ≤ 0.25 M _{n, the} block for optimal common-mode current control 32 sets a small initial value of the common-mode current amplitude I _syn ≤ 0.5 I _{syn n} , which creates an initial the rotor flux linkage ψ _ro , which is necessary for optimal performance in working out in the dynamics of the jump in the maximum moment M _max with a limited amplitude of the stator current i _{s add.} = 1.5 - 5 i _sn .

относительно потокосцепления ротора

астатически регулируется на оптимиальном уровне, равном ±π/4, что обеспечивает оптимизацию режимов асинхронного электродвигателя 4 по максимуму отношения момента M к току i_s и по минимуму потерь электроэнергии в системе "инвертор - силовой кабель - асинхронный электродвигатель".In the main second control zone, at a rotation speed less than the boundary value ω _gr established in the adaptive moment controller 11, the optimum common-mode current control unit 32 generates an output signal of a given common-mode current amplitude I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ in proportion to the measured amplitude of the orthophase current I _ort obtained at the output of the inverse transducer 30 Cartesian coordinates. In this main regulation zone, as a result of the astatic equalization of the amplitudes of the in-phase and orthophase currents, I _syn = I _ort the phase shift angle ε _φ of the stator current vector

regarding rotor flux linkage

It is statistically regulated at the optimum level equal to ± π / 4, which ensures optimization of the modes of the asynchronous electric motor 4 to the maximum ratio of the moment M to the current i _s and to the minimum energy loss in the "inverter - power cable - asynchronous electric motor" system.

For static modes in this main regulation zone, the implemented optimization of the phase shift angle adjustment ε _φ = ± π / 4 with respect to the maximum torque to current ratio is manifested in minimizing the stator current i _{s min} for a given load moment M _s in minimizing the total power loss in an induction motor. At the same time, the power losses in the inverter 1 and the voltage losses in the power cable connecting the power outputs of the inverter 1 with an induction motor are minimized, which minimizes the loss of electricity in the electric drive.

For dynamic starting, braking, rotor locking and rapid rotation speed changes, the implemented optimization of the phase shift angle control ε _φ = ± π / 4 is manifested in reaching the maximum dynamic moment M _max when the maximum value for inverter 1 is set in block 6 of PWM current regulators, phase current i _s ≤ i _{s add.} , which leads to minimization of energy losses in the dynamics and creates a mode of maximum speed changes in torque and speed.

In a high-speed electric drive with high values of the permissible stator current i _{s add.} > 3 i _{sн the} optimal process of forcing the amplitude of the rotor flux linkage at ε _φ = ± π / 4 occurs in the zone of high magnetic saturation of the magnetic circuit of the induction motor to ψ _{r opt} ≥ 1.3, which is taken into account in the block 32 for optimal control of the common-mode current by reducing the proportionality coefficient K < 1 preset common-mode current amplitude I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ _opt = K ₁ I _ort , starting with the boundary value I _ort .

When the current rotation speed ω increases above the boundary value ω _gr specified in the adaptive moment controller 11, the optimal vector control passes to the third control zone as a result of a decrease in the value of the given common-mode current amplitude I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ (ω) with an increase in the current speed ω> ω _gr , which is realized with the help of the adaptive regulator 11 of the moment due to the input of the signal of the dependent attenuation of the common-mode current from the second output of the adaptive regulator 11 of the moment to the second input of the block 32 for optimal control of the common-mode current.

In all three control zones, the adaptive moment controller 11 generates an output signal for setting the amplitude of the orthophase current I _ort inversely proportional to the indirectly measured amplitude of the rotor flux link ψ _r obtained from the measured voltages and currents at the inverter 1 output, which automatically takes into account with increased accuracy the magnetic saturation of the asynchronous magnetic circuit motor physical connection due to the measured phase voltage and phase currents of the stator.

относительно синусной и косинусной функций фазы φ_s потокосцепления ротора и регулирования амплитуды синфазного тока, пропорционально измеренной амплитуде ортофазного тока.When the connection of the output of the adder of 35 frequencies with the feedback input of the speed controller 12 is opened, the electric drive is converted from a speed-controlled drive to a torque drive, in which the first input M ^{* of the} adaptive torque controller 11 forms an input for setting the moment of the torque drive, for example, a traction electric drive of an electric vehicle. In this case, all the described processes of optimal vector control are performed in three regulation zones. The basic condition for the optimal vector control of the phase shift angle ε _φ of the stator current vector is satisfied due to the forced orientation of the stator current vector

relative to the sine and cosine functions of the phase φ _{s of the} rotor flux linkage and regulation of the common-mode current amplitude, in proportion to the measured orthophase current amplitude.

 The electric drive according to the circuit of FIG. 1 can be implemented on discrete analog and digital-analog elements similar to the elements used in the prototype [2].

 The method of optimal vector control of an asynchronous electric motor carried out in the electric drive of FIG. 1 can be implemented in a microprocessor electric drive according to the circuit of FIG. 2.

 A microprocessor-based electric drive with optimal vector control of an asynchronous electric motor (Fig. 2) contains an inverter 1 (Fig. 2), the power outputs of which are connected to the stator windings of the asynchronous motor 4 through the sensors 2, 3 of the phase current.

 The control inputs of the inverter 1 are connected to the control pulse generation unit 5, made in the form of a driver chip with an additional PWM resolution input. The inputs of the driver microcircuit are connected to the outputs of the microprocessor optimal vector control device.

 The microprocessor vector control device comprises a digital signal coprocessor 36, a digital signal processor 37, a read-only memory unit 38, a serial port 39, a control and diagnostic console 40. The digital signal coprocessor 36 contains the following blocks built into the coprocessor: a three-phase pulse width modulation timer unit 41, an integrated multiplex analog-to-digital converter 42, an integrated serial command programmer 43, an integrated vector conversion unit 44, and a register control unit 45. In addition, the microprocessor vector control device and the digital signal coprocessor 36 includes a data bus 46 and an address bus 47 connecting the device units.

 The four inputs of the microprocessor vector control device formed by the inputs of the integrated multiplex analog-to-digital converter 42 of the digital signal coprocessor 36 are connected to two outputs of the phase voltage sensor 14 and, through the galvanic isolation unit 48, to two outputs of the phase current sensors 2, 3. The phase voltage sensor 14 is connected to the power outputs of the inverter 1. In the microprocessor optimal vector control device, the digital signal processor 37 is connected by inputs / outputs to the inputs / outputs of the serial port 39, through which information is exchanged with the program control and diagnostics panel 40.

On the address bus 47, address commands are received from the output of the digital signal processor 37 to the input of the digital signal coprocessor 36 and to the input of the integrated multiplex analog-to-digital converter 42 of the digital signal coprocessor 36. Analog signals of the values of the measured phase voltages U _sa , U _sb and the measured phase currents i _sa , i _sb come from the outputs of the sensor 14 phase voltages and block 48 galvanic isolation to the analog inputs of the integrated multiplex analog-to-digital Converter 42 digital signal copro Essor 36.

и векторе тока

по шине данных 46 поступает с выхода встроенного мультиплексного аналого-цифрового преобразователя 42 на входы встроенного блока 44 векторных преобразований, в котором происходит прямое и обратное преобразование двухфазно-трехфазных координат α,β - a, b, c и прямое и обратное преобразование декартовых координат x, y - α,β. Широтно-импульсная модуляция цифровых кодов (например, 12-разрядных кодов), рассогласований заданных и измеренных токов производится в блоке 41 трехфазного таймера широтно-импульсной модуляции, связанного шиной данных 46 с встроенным блоком 44 векторных преобразований.Via the address bus 47, the digital inputs of the integrated multiplex analog-to-digital converter 42 receive commands for sequential analog-to-digital conversion of the analog signals U _sa , U _sb , i _sa , i _sb into a digital n-bit code, for example, an 11-bit code. Received digital voltage vector measurement information

and current vector

via the data bus 46, it comes from the output of the built-in multiplex analog-to-digital converter 42 to the inputs of the built-in vector transform unit 44, in which the direct and inverse transforms of the two-phase-three-phase coordinates α, β - a, b, c and the direct and inverse transforms of the Cartesian coordinates x , y - α, β. Pulse-width modulation of digital codes (for example, 12-bit codes), mismatch of set and measured currents is performed in block 41 of a three-phase pulse-width modulation timer connected by data bus 46 with an integrated block 44 of vector transformations.

 The sequence of command execution and computational operations with digital codes of optimal vector control values is generated and transmitted via address bus 47 by a digital signal processor 37 and by priority by an integrated serial programmer 43 commands, which receives current information about the operations performed from the data bus 46. By instructions received via the bus address 46 from the output of the digital signal processor 37 to the input of the block 38 of read-only memory.

 On the data bus 46, the read-only memory unit 38, when accessing the memory via the address bus 47, transmits information to the inputs of the digital signal processor 37 and the digital signal coprocessor 36. The register control unit 45 and the integrated serial program programmer 43 control the operations of the vector converter performed in the built-in unit 44 vector transformations, and in block 41 of a three-phase pulse width modulation timer. The results of the computational processes are stored in the registers of the digital signal coprocessor 36 controlled by the register control unit 45.

 Using the remote control 41 program control and diagnostics through a serial port (for example, RS 232 or RS 485) exchanges the master and control information between the remote control 41, the processor 37 and the coprocessor 36. Information about the excess of the measured and calculated parameters is displayed on the remote control and diagnostics panel 41 electric drives of established permissible values for current, voltage, torque, speed, acceleration.

 The remote control 41 of program control and diagnostics, which may include a touch keyboard, additional processor and digital indicator, sets the control mode “torque electric drive”, “high-speed electric drive”, the rate of change of moment or speed, the path of change of moment and speed in time, the limit and maximum speed, the value of the set speed, the value of the set moment, the "stop under load" mode. Remote 41 program control and diagnostics at the same time can perform system functions of control and regulation of machines and mechanisms, including the adjustment of the angular position or other technological parameter for the purpose of the machine or mechanism.

 The microprocessor-based electric drive according to the circuit of FIG. 2 can be performed on standard type elements. Inverter 1 can be made in the form of a typical solid-state six-key IGBT module on insulated gate bipolar transistors (IGBT) with reverse diodes (IGBT-key-IGBT circuit is shown in Fig. 2 in the circuit denoting inverter 1).

 As a driver microcircuit (pulse forming unit 5), a typical driver microcircuit for IGBT modules can be used.

 As a digital signal coprocessor 36, an LSI digital signal motion coprocessor ADM200 having a speed of 12.5 MHz or an ADMS201 coprocessor can be used.

 As a digital signal processor 37, a typical digital signal processor (DSP) can be used, for example, types ADSP-2115 or TMS 320C20, TMS 320C25, TMS 320C25-50.

 In block 38 of read-only memory, a typical programmable read-only memory chip may be used, for example with ultraviolet erasure.

 As the serial port 38, typical serial interface microcircuits, for example RS232, RS485, can be used.

 The remote control 40 software control and diagnostics can be performed on a typical digital signal processor (DSP), for example types TMS 320C20, TMS 320C25, TMS 320C25-50 or ASSP-2115.

 An optimal vector control microprocessor drive according to the circuit of FIG. 2 works as follows.

The semiconductor switches of the inverter 1 are switched with a constant high switching frequency f _k , depending on the speed and heat dissipation of the keys or IGBT modules, in the frequency range 3 - 10 KHz, which determines the period T _{t of the} pulse-width modulation of the inverter I and the time T _m of the microprocessor cycle control devices T _m <T _t . As a result of switching the keys, the inverter 1 through the sensors 2, 3 of the phase current supplies the induction motor 4 with a pulse-width modulated sequence of power voltage pulses, the duration of which is determined by the duration of the control pulses coming from the output of the pulse forming unit 5 (driver chip) in the case of a PWM enable signal with the output of the digital signal coprocessor 36 to the additional input of the driver chip 5.

 The signals of the measured phase currents from the outputs of the sensors 2, 3 of the phase currents are fed through the galvanic isolation unit 48 to the analog inputs of the digital signal coprocessor 36. The signals of the measured phase voltages from the outputs of the sensor 14 of the phase voltage are supplied to the other analog inputs of the digital signal coprocessor 36. When the phase currents are exceeded or phase voltages of the maximum allowable values programmatically set in the digital signal coprocessor 36, there is no PWM resolution signal at the additional input of the microcircuits Driver 5 and in inverter 1 have all the semiconductor switches locked, an overload signal for current or voltage is supplied via serial port 39 to the control and diagnostic panel.

If there is a PWM enable signal at the additional input of the driver microcircuit, the moments of sampling of the measurement information i _sa , i _sb , U _sa , U _sb within the clock period T _{t of the} PWM are set by the built-in multiplex analog-to-digital converter 42 depending on the information about the beginning of the next period T _t clock PWM formed in block 41 of a three-phase pulse width modulation timer. The interruption frequency of the analog-to-digital signal converter i _sa , i _sb , U _sa , U _sb is determined by a sampling frequency of the order of 10 KHz. The sampling time for four channels is about 15 μs.

Information about the completion of the next sampling of digital codes i _sa , i _sb , U _sa , U _sb enters the digital signal processor 37, which gives the command to start the next cycle of vector transformations performed by the built-in block 44 of vector transformations.

In the digital signal coprocessor 36, vector information processing and dual-circuit regulation of stator currents are performed. The internal circuit for regulating the instantaneous phase currents i _sa , i _sb , i _sc is implemented in block 41 of the three-phase PWM timer by pulse-width modulation of digital mismatch codes of the set and measured phase currents. The external control loop of the orthophase and common mode currents is implemented by the built-in block 44 of vector transformations. The optimal vector torque and speed vector control algorithm is implemented by a digital signal processor 37 and a read-only memory unit 38 according to a previously entered program, which is the software of a microprocessor-based optimal vector control device.

Initially, the drive operates in the first zone of common-mode current regulation I _syn = I _{syn 0} at zero amplitude of the orthophase current I _ort = 0 and zero initial synchronization phase φ _so = 0, which is provided by the initial conditions introduced into the program of the digital signal processor 37 when setting remote control 40 control and diagnostics of zero initial data M ^* = 0, ω ^* = 0 and commands to turn on the drive.

i_sc =

регулируемыми цифровым сигнальным сопроцессором 36 путем широтно-импульсной модуляции с частотой тактирования ШИМ цифровых кодов рассогласований заданных и измеренных постоянных токов.The phase windings of the stator of the induction motor 4 are powered by direct currents i _sa = I _{syn 0} ; i _sb =

i _sc =

adjustable by a digital signal coprocessor 36 by pulse-width modulation with a clock frequency of the PWM digital mismatch codes of the given and measured constant currents.

совпадает с трехфазной системой начальных фазных токов, в связи с чем начальный вектор потокосцепления ротора ψ_r совпадает с осью "α" опорной фазы "a" асинхронного электродвигателя и осью α неподвижной декартовой системы координат α,β. Установившиеся постоянные фазные напряжения U_sa = U_so, U_sb =

U_sc =

измеряются датчиком 14 фазных напряжений.In an induction motor 4, a constant magnetic field is excited, characterized by a constant initial flux linkage of the rotor ψ _ro . Steady-state three-phase system of initial phase rotor flux linkages

coincides with the three-phase system of initial phase currents, and therefore the initial rotor flux linkage vector ψ _r coincides with the axis α of the reference phase a of the asynchronous electric motor and the axis α of the fixed Cartesian coordinate system α, β. Steady-state constant phase voltages U _sa = U _so , U _sb =

U _sc =

measured by the sensor 14 phase voltage.

The built-in multiplex analog-to-digital converter 442 transmits samples of digital codes of measured values i _sa , i _sb , U _sa , U _sb via data bus 46 to the built-in vector block 44 of the coprocessor 36 and to the input of the digital signal processor 37.

The calculated values of the normalized sine and cosine functions of the initial phase φ _so = 0 are cos φ _so = 1, sin φ _so = 0. The rotor of the asynchronous electric motor 4 is stationary, since there is no orthophase current of the stator (I _ort = 0) and the moment is zero.

относительно вектора потокосцепления ротора

за время меньшее 0,05 M_с устанавливается на оптимальном уровне ±π/4, определяемом соотношением амплитуд I_ort, I_syn ортофазного и синфазного токов. Вследствие сдвига фазы вектора тока статора

на угол π/4 относительно вектора потокосцепления ротора, происходит максимальное приращение момента M асинхронного электродвигателя 4 для заданного приращения амплитуды фазного тока статора

Образуемый динамический момент M приводит к угловому перемещению ротора асинхронного электродвигателя 4, что вызывает изменения угла и величины вектора потокосцепления ротора

и соотношения фазных напряжений U_sa, U_sb, обеспечивающих автоматическое регулирование изменившихся фазных токов i_sa, i_sb.Then, after the signal from the control panel 40 and the diagnostics of the speed reference signal ω ^* or the moment reference signal M ^{* is} supplied, the digital signal processor 37 performs the operation of dividing the predetermined moment signal M ^* by the calculated value of the rotor flux coupling amplitude ψ _ro . The resulting digital increment of the specified orthophase current amplitude via the data bus 46 is supplied to the built-in block 44 of vector transformations of the coprocessor 36, operating in the sequential interrupt mode controlled by the built-in serial programmer 43 of the commands and the block 45 of register control of the coprocessor 36. Block 41 of the three-phase timer pulse-width modulation of the coprocessor 36 changes the ratio of the durations of the open and closed state of the inverter switch 1 is switched from a constant high switching frequency _to f 3 KHz output pulses changes moments levels chip driver 5. As a result of changing the ratio of the durations of the open state of the inverter 1 are changed key phase voltages U _sa, U _sb, U _sc and thereupon vary currents i _sa, i _sb, i _sc thus that the phase shift angle ε _φ of the stator current vector

relative to the rotor flux linkage vector

for a time shorter than 0.05 M _{s, it is} set at the optimal level ± π / 4, determined by the ratio of the amplitudes I _ort , I _{syn of the} orthophase and in-phase currents. Due to the phase shift of the stator current vector

at an angle π / 4 relative to the rotor flux linkage vector, the maximum increment of the moment M of the induction motor 4 occurs for a given increment of the stator phase current amplitude

The resulting dynamic moment M leads to the angular displacement of the rotor of the induction motor 4, which causes changes in the angle and magnitude of the rotor flux linkage vector

and the ratio of phase voltages U _sa , U _sb , providing automatic control of the changed phase currents i _sa , i _sb .

The changed measurement information from the outputs of the integrated multiplex analog-to-digital converter is supplied to the built-in vector conversion unit 44, in which the values of i _sβ , i _sβ , u _sα , u _sβ , I _ort , I _syn are calculated.

In the digital signal processor 37, the increment of the values of the normalized harmonic functions cosφ _s , sinφ _{s is} calculated, which are supplied to the built-in block 44 of vector transformations and specify the increment of the stator current phase. Using the block 38 of read-only memory in the digital signal processor 37, the optimal values of the common-mode current I are calculated $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ _onm , subsequent increment of the orthophase current I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ , increments of the slip frequency Δω, synchronization frequency ω _s and rotation speed ω according to previously calculated increments of cosφ _s , sinφ _s , I _ort , ψ _r .
The current rotor speed ω increases to a predetermined rotation speed ω ^* and is statically adjusted at a given level by transforming the increment of the mismatch of the current speed ω relative to a given speed ω ^* according to the proportional-integral law implemented in the digital signal processor 37.

Adaptive three-zone torque control depending on the values of I entered in the constant memory unit 38 $\genfrac{}{}{0ex}{}{\text{*}}{\text{so}}$ , ω _gr , i _{s max} , M $\genfrac{}{}{0ex}{}{\text{*}}{\text{max}}$ , ω $\genfrac{}{}{0ex}{}{\text{*}}{\text{max}}$ is performed in the digital signal processor 37 according to the optimal laws of changing the phase shift angle ε _φ and the given common-mode amplitude I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ _onm as a function of the current measured amplitude of the orthophase current.

The optimization of the electric drive to minimize energy losses is achieved in the electric drive by adjusting the in-phase current amplitude I _syn mainly in proportion to the measured orthophase current amplitude I _ort .

 The speed and accuracy of torque and speed control are increased by indirectly measuring the amplitude and phase of the rotor flux linkage from the measured phase voltages and phase currents, including in strong nonlinear magnetic saturation modes of the magnetic circuit of an induction motor and by calculating the current increment of the slip frequency in proportion to the increment of the ratio of the measured amplitude orthophase current and the indirectly measured amplitude of the flux linkage of the rotor.

 In the static operating modes of the electric drive in the main control zone, the stator current consumption is minimized to create a moment equal to the load moment, which reduces the total power loss in the inverter – power cable – asynchronous electric motor system, reduces voltage losses in the power cable, for example, in the deep submersible cable in oil production, and in the case of the use of a battery at the inverter input, for example, in an electric car, increases the energy resource of the machine and the path length of the electric car .

 Electric drives according to the schemes of FIG. 1, 2 differ in the design of the optimal vector control system, the structures of which depend on the types of integrated circuits used, VLSI, microprocessors. When using known Intel or Motorola microprocessors, the drive circuit may have structural differences from the circuitry of FIG. 1, 2. Common to various embodiments of an electric drive that solves the problem and realizes the purpose of the invention is the method of optimal vector control of an induction motor, implemented in the described electric drives of FIG. 12.

An optimal vector control method in addition to FIG. 1, 2 is illustrated by drawings in FIG. 3 - 7. The features of the applied control method that distinguish it from the known methods of frequency, frequency-current and vector control are, as can be seen from the diagrams in FIG. 1, 2, the absence of a channel for setting the frequency of the current or voltage, the absence of a channel for controlling the sliding frequency, the absence of sensors on the electric motor, the absence of a link to simulate the saturation of the magnetic circuit or taking into account the magnetization curve of an asynchronous electric motor when the phenomenon of the strong magnetic saturation induction motor at high torque M ≥ M _μ mode equality of amplitudes B and ortofaznogo phase currents.

и вектора напряжения

(модуль, фазовый угол, частота вращения вектора) в функции измеренных параметров этих же векторов

по замкнутым циклам в зависимости от заданного момента и связанного с ним оптимального задания амплитуды ортофазного тока.The essence of the method of optimal vector control of an induction motor is the optimal interconnected regulation of all parameters of the current vector

and stress vectors

(module, phase angle, vector rotation speed) as a function of the measured parameters of the same vectors

in closed cycles, depending on a given moment and associated optimal specification of the amplitude of the orthophase current.

и напряжения

происходит в результате реакции асинхронного электродвигателя при возникновении действительного момента M асинхронного электродвигателя и приращении действительной скорости вращения ω ротора асинхронного электродвигателя. Реакция асинхронного электродвигателя по закону электромагнитной индукции создает автоматическое задание и изменение частоты и фазы тока статора. Векторная обработка этой реакции асинхронного электродвигателя производится такой последовательностью операций и таким взаимным изменением параметров векторов

чтобы выполнялись критерии оптимальности режимов асинхронного электродвигателя:
по минимуму электрических потерь в асинхронном двигателе на возбуждение магнитного поля в режимах холостого хода и малых величинах нагрузки M ≤ 0,25 M_н (i_so = i_{so min}) при заданных начальных условиях для динамики;
по минимуму тока статора в статических режимах электропривода при заданном моменте нагрузки M_с в основной зоне регулирования (i_s = i_{s min} при M_с ≥ 0,25 M_н);
по минимуму потерь мощности в асинхронном электродвигателе при регулировании скорости в диапазоне ω = 0-ω_гр;
по максимуму выходной механической мощности P_м = M • ω = P_{м max} при регулировании скорости вверх от граничной скорости ω_гр при ограниченном напряжении статора;
по максимуму момента электродвигателя M = M_max при ограниченном токе статора is = i_{s max} в динамических режимах пуска, торможения, изменения скорости и компенсации возмущений со стороны нагрузки;
по максимуму быстродействия отработки приращений заданного момента M^* и заданной скорости вращения ω^* в динамических режимах;
по минимуму ошибки между заданными и действительными величинами момента и скорости вращения в статических и динамических режимах асинхронного электродвигателя при расширенных диапазонах регулирования момента M = 0 до M = 5-8 M_н и ω скорости от ω = 0 до ω = 4ω_н в обоих направлениях.Mutual change of current vector parameters

and voltage

occurs as a result of the reaction of the induction motor when a real moment M of the induction motor occurs and the actual rotation speed ω of the rotor of the induction motor increases. The reaction of an induction motor according to the law of electromagnetic induction creates an automatic setting and change of the frequency and phase of the stator current. Vector processing of this reaction of an induction motor is carried out by such a sequence of operations and such a mutual change in the parameters of the vectors

to satisfy the optimality criteria of the modes of an induction motor:
to minimize electric losses in an induction motor to excite a magnetic field in idle conditions and small load values M ≤ 0.25 M _n (i _so = i _{so min} ) under given initial conditions for dynamics;
minimum stator current in static modes of the electric drive at a given load moment M _s in the main control zone (i _s = i _{s min} at M _s ≥ 0.25 M _n );
to minimize power losses in an induction motor when controlling speed in the range of ω = 0-ω _gr ;
at the maximum output mechanical power P _m = M • ω = P _{m max} when controlling the speed upwards from the boundary speed ω _g with a limited stator voltage;
to the maximum of the motor moment M = M _max with a limited stator current is = i _{s max} in dynamic modes of starting, braking, changing the speed and compensating for disturbances from the load side;
to maximize the speed of working out the increments of a given moment M ^* and a given rotation speed ω ^* in dynamic modes;
to minimize errors between the set and actual values of the moment and speed of rotation in the static and dynamic modes of the induction motor with extended ranges of torque control M = 0 to M = 5-8 M _n and ω speeds from ω = 0 to ω = 4ω _n in both directions .

 The integrally listed optimality criteria are expressed as minimizing energy losses while increasing the speed, accuracy and range of regulation of the moment, speed and magnetic flux of an induction motor without the use of sensors structurally associated with an asynchronous motor.

 The minimization of the electric energy consumed by an induction motor in the proposed optimal vector control method is the main essence of the method for asynchronous electric motors that are not adjustable in speed and moment, instead of electric drives with known protective-disconnecting equipment, for example, with magnetic starters.

 The control sequence and interrelated actions on the control object (asynchronous electric motor) are illustrated by mathematical equations and diagrams in FIG. 3 - 7.

 The axis of one of the phases of the stator, for example, the axis of phase "a" (Fig. 3), is taken as the supporting axis α of the vector control in the stationary three-phase stator coordinates a, b, c.

производится относительно неподвижной опорной оси α = a в положительном направлении против часовой стрелки (фиг. 3). Это условие реализуется тем, что измеренный фазный ток i_sa и измеренное фазное напряжение U_sa в опорной фазе "a" используются при обработке измерительной информации в качестве измеренных величин в двухфазной (декартовой) координатной системе α,β:i_sα = i_sa, u_sα= u_sa. Физически при осуществлении способа это осуществляется тем, что мгновенные величины всех переменных состояния асинхронного электродвигателя в фазе "a" используются в качестве проекций векторов состояния асинхронного электродвигателя

на неподвижную ось α = a и выражаются косинусными функциями угла соответствующего вектора относительно оси фазы "a".Counting angles of vectors

is produced relative to the fixed reference axis α = a in the positive direction counterclockwise (Fig. 3). This condition is realized in that the measured phase current i _sa and the measured phase voltage U _sa in the reference phase "a" are used in the processing of measurement information as measured values in the two-phase (Cartesian) coordinate system α, β: i _sα = i _sa , u _sα = u _sa . Physically, when implementing the method, this is achieved by the fact that the instantaneous values of all state variables of the induction motor in phase "a" are used as projections of the state vectors of the induction motor

to the fixed axis α = a and are expressed by the cosine functions of the angle of the corresponding vector relative to the phase axis "a".

где φ_s(t) - фаза синхронизации, равная фазе потокосцепления ротора

относительно неподвижной опорной оси "α" (фиг. 3);
ε_φonm - оптимальный угол фазового сдвига вектора тока статора относительно вектора потокосцепления ротора

Формирование мгновенных величин фазных токов i_sa, i_sb, i_sc по закону (1) и оптимального по минимуму потерь электроэнергии закона регулирования угла фазового сдвига ε_φonm вектора тока статора

относительно вектора потокосцепления ротора

осуществляются в способе по измеренным величинам приращений напряжений u_sα, u_sβ (двухфазного напряжения) и тока i_sα,i_sβ (двухфазного тока) с использованием электромагнитной реакции асинхронного электродвигателя на эти приращения согласно уравнениям асинхронного двигателя в двухфазной системе координат α,β:
для вектора потокосцепления статора

для вектора потокосцепления ротора

:

где R_s - активное сопротивление фазы статорной цепи (фазной обмотки статора);

переходная индуктивность статорной цепи (фазной обмотки статора), примерно равная сумме индуктивностей рассеяния статора и ротора L_σs+L_σr;
K_р - коэффициент связи ротора, близкий к единице.The stator phase currents are formed according to the cosine law

where φ _s (t) is the synchronization phase equal to the rotor flux linkage phase

relatively stationary reference axis "α" (Fig. 3);
ε _φonm is the optimal phase angle of the stator current vector relative to the rotor flux linkage vector

The formation of instantaneous values of phase currents i _sa , i _sb , i _sc according to the law (1) and the optimal law of regulation of the phase shift angle ε _φonm of the stator current vector

relative to the rotor flux linkage vector

are carried out in the method according to the measured values of the voltage increments u _sα , u _sβ (two-phase voltage) and current i _sα , i _sβ (two-phase current) using the electromagnetic reaction of an induction motor to these increments according to the equations of an induction motor in a two-phase coordinate system α, β:
for stator flux linkage vector

for rotor flux linkage vector

:

where R _s is the active resistance of the phase of the stator circuit (phase winding of the stator);

transient inductance of the stator circuit (phase stator winding), approximately equal to the sum of the stator and rotor leakage inductances L _σs + L _σr ;
K _p - rotor coupling coefficient close to unity.

в этих координатных системах, в том числе проекции по осям α,β, описываемые уравнениями (2), (3).In FIG. 32a), 3b), 3d), 3e) shows the relative position of the two-phase coordinate system α, β relative to the three-phase coordinate system a, b, c and in FIG. 3d) shows projections of vectors

in these coordinate systems, including projections along the α, β axes, described by equations (2), (3).

 Physically, electromagnetic processes in an asynchronous electric motor (HELL) are manifested in the interaction of state vectors shown in vector diagrams to the right of the vertical bar in FIG. 4 (AD region) in FIG. 3c) and 3d).

 The vector control processes in the vector control system (VCA) are shown in FIG. 3 to the left of the dashed line (area of the VCA): in the forward direction of actions to the induction motor - in FIG. 3a), 3b), in the opposite direction from the induction motor - in FIG. 3d), 3e). The arrows in FIG. Figure 3 shows the sequence of operations and processes of optimal vector control, including processes in an asynchronous electric motor, participating in the vector control method.

The control actions in FIG. 3 are indicated with an index ( ^* ), the actual variables of the induction motor are expressed by signs without an index.

асинхронного электродвигателя, описываемый уравнениями (1) и представленный векторами

на фиг. 3а), 3в), регулируется равным разности двух периодических токов - косинусоидального (синфазного) тока

и синусоидального (ортофазного) тока

амплитуды которых I_syn, I_ort являются проекциями вектора тока статора

на ортогональные оси вращающейся декартовой системы координат x, y (фиг. 3а), а фаза φ_s равна фазе вектора потокосцепления ротора

относительно неподвижной опорной оси

Ось "x" вращающейся декартовой системы координат x, y совпадает с направлением вектора

(фиг. 3д). Такое регулирование фазных токов статора согласно способу описывается уравнениями:

где синфазные токи регулируются по косинусоидальному закону:

а ортофазные токи регулируются по синусоиадальному закону:

Фаза φ_s(t) является фазой вектора потокосцепления ротора

который изменяется по закону:

Из сравнения уравнений (5), (7) следует, что согласно способу управления вектор синфазного тока

совпадает по направлению с ориентирующей осью "x" и с вектором потокосцепления ротора ψ_r (фиг. 3а), 3д) и изменяется синфазно с ориентирующей осью "x" и потокосцепления ротора, что отражено в названии этого тока, так как модуль вектора

(амплитуда ψ_r ) всегда положителен, то вектор i_syn однонаправлен в положительном направлении ориентирующей оси "x".Stator alternating current

asynchronous motor described by equations (1) and represented by vectors

in FIG. 3a), 3c), it is regulated equal to the difference of two periodic currents - cosine (common-mode) current

and sinusoidal (orthophase) current

whose amplitudes I _syn , I _ort are projections of the stator current vector

on the orthogonal axis of the rotating Cartesian coordinate system x, y (Fig. 3a), and the phase φ _s is equal to the phase of the rotor flux linkage vector

relatively motionless supporting axis

The x axis of the rotating Cartesian coordinate system x, y coincides with the direction of the vector

(Fig. 3d). Such a regulation of the stator phase currents according to the method is described by the equations:

where common-mode currents are regulated according to the cosine law:

and orthophase currents are regulated according to the sinusoidal law:

The phase φ _s (t) is the phase of the rotor flux linkage vector

which changes by law:

From a comparison of equations (5), (7) it follows that according to the control method, the common-mode current vector

coincides in direction with the orienting axis "x" and with the rotor flux vector ψ _r (Fig. 3a), 3d) and changes in phase with the orienting axis "x" and rotor flux links, which is reflected in the name of this current, since the vector module

(amplitude ψ _r ) is always positive, then the vector i _{syn is} unidirectional in the positive direction of the orienting axis "x".

ортогонален по отношению к вектору

что отражено в названии ортофазного тока.Sinusoidal (orthophasic) current lags or is ahead of the common mode current according to equations (6) by 90 ^o , vector

orthogonal to the vector

which is reflected in the name of the orthophase current.

на ось "y" (фиг. 3а) определяется знаком "+" или "-" в уравнениях (6).Projection direction

on the axis "y" (Fig. 3A) is determined by the sign "+" or "-" in equations (6).

согласно уравнениям (2).The rotor flux linkages are excited in an induction motor according to equations (3) as a result of the stator flux excitation of the stator flux links

according to equations (2).

The stator flux linkage increments Δψ _s for the time interval Δt according to Eqs. (2) occur along the α, β axes when there is an increment in the difference Δu _α , Δu _β between the stator voltages u _sα , u _sβ and the stator voltage drops proportional to the stator currents i _sα , i _sβ that is the physical basis of the proposed method for optimal vector control of an asynchronous electric motor without sensors structurally associated with the electric motor.

 By measuring increments and converting voltage differences, mode optimization is achieved.

Ориентация вектора производится по вектору

косвенно измеряемому по измеренным величинам напряжения u_sα,u_sβ и тока i_sa, i_sβ статора на основании уравнений (2), (3).The increment of the relative angular velocity Δω of the magnetic field with respect to the rotor (slip frequency) is detected in the method based on the differential equations of the rotor chain, the main parameters of which are the projections of the rotor flux linkage vector

The orientation of the vector is made by vector

indirectly measured from the measured values of voltage u _sα , u _sβ and current i _sa , i _{sβ of the} stator based on equations (2), (3).

в декартовой координатной системе x, y, ориентированной осью "x" по вектору

согласно способу управления устанавливается в следующем порядке: измеряют двухфазное напряжение u_sα,u_sβ и двухфазный ток i_sα,i_sβ по величинам которых вычисляют двухфазные потокосцепления статора ψ_sα,ψ_sβ согласно уравнениям (2).The sequence of operations of the vector orientation of the stator current

in the Cartesian coordinate system x, y, oriented by the x axis along the vector

according to the control method, it is established in the following order: two-phase voltage u _sα , u _sβ and two-phase current i _sα , i _{sβ are measured} by the values of which two-phase stator flux _linkages ψ _sα , ψ _{sβ are calculated} according to equations (2).

в двухфазной (неподвижной) декартовой системе координат α,β (фиг. 3д)). Затем вычисляют по уравнениям (3) косвенно измеренные величины двухфазного потокосцепления ротора ψ_rα,ψ_rβ (фиг. 3д)). После этого на основании ортогональности осей α,β вычисляют амплитуду потокосцепления ротора ψ_r по формуле

Затем вычисляют по косвенно измеренным величинам двухфазного потокосцепления ротора ψ_rα,ψ_rβ и вычисленной амплитуде потокосцепления ротора ψ_r нормированные косинусную и синусную функции фазы φ_s потокосцепления ротора согласно выражениям

Полученные нормированные (с единичной амплитудой) косинусную и синусную функции фазы φ_s потокосцепления ротора характеризуют единичный вектор

во вращающейся декартовой системе координат x, y, ориентированной осью "x" по направлению вектора

(фиг. 3д)), благодаря чему фаза потокосцепления ротора одновременно образует равную ей фазу синхронизации φ_s регулируемых синфазного и ортофазного токов

(фиг. 3е)).The obtained values of ψ _sα , ψ _sβ form indirectly measured projections of the stator flux linkage vector

in a two-phase (fixed) Cartesian coordinate system α, β (Fig. 3d)). Then, indirectly measured values of the two-phase flux linkage of the rotor ψ _rα , ψ _rβ are calculated according to equations (3) (Fig. 3d)). After that, on the basis of the orthogonality of the axes α, β, the amplitude of the flux linkage of the rotor ψ _{r is} calculated by the formula

Then, the indirectly measured values of the two-phase rotor flux linkage ψ _rα , ψ _rβ and the calculated rotor flux linkage amplitude ψ _{r are} calculated, the normalized cosine and sine functions of the rotor flux linkage phase φ _s according to the expressions

The obtained normalized (with unit amplitude) cosine and sine phase functions φ _{s of the} rotor flux linkage characterize the unit vector

in a rotating Cartesian coordinate system x, y, oriented by the x-axis in the direction of the vector

(Fig. 3d)), due to which the rotor flux coupling phase simultaneously forms the synchronization phase φ _{s of the} adjustable in-phase and orthophase currents equal to it

(Fig. 3e)).

как показано на фиг. 3а), б), в соответствии с уравнениями прямого и обратного преобразований координат.In the synchronization phase function φ _s, according to the expressions (9) obtained, the normalized cosine and sine functions of the synchronization phase determine and regulate the in-phase and orthophase currents

as shown in FIG. 3a), b), in accordance with the equations of direct and inverse transformations of coordinates.

где I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ - заданная амплитуда синфазного тока;
I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ - заданная амплитуда ортофазного тока;
φ_s - фаза синхронизации.The operation of direct transformation of the Cartesian coordinates x, y α, β (Fig. 3a)) is performed according to the formulas:

where i $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ - a given amplitude of the common-mode current;
I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ - a given amplitude of the orthophase current;
φ _s is the phase of synchronization.

Обратное преобразование двухфазно-трехфазных координат a, b, c ⇒ α, β (фиг. 3д)) выполняется согласно выражениям:

Операция обратного преобразования декартовых координат α,β x, y выполняется в соответствии с формулами

Последовательность выполнения операций над токами по шагам производится от операции (10), затем (11) к операции (12) и затем (13), как показано стрелками на фиг. 3.Direct conversion of two-phase-three-phase coordinates α, β a, b, c (Fig. 3b)) is performed according to the formulas

The inverse transformation of the two-phase-three-phase coordinates a, b, c ⇒ α, β (Fig. 3d)) is performed according to the expressions:

The inverse transformation of the Cartesian coordinates α, β x, y is performed in accordance with the formulas

The sequence of operations on currents in steps is carried out from operation (10), then (11) to operation (12) and then (13), as shown by arrows in FIG. 3.

Operations (10), (11) are performed on control quantities ( ^* ), operations (12), (13) are performed on the obtained measured values i _sa , i _sb .

As a result of a closed control loop, in steps (10) - (12) and (12) - (10), the stator phase currents i _sa , i _sb , i _{sc are} regulated equal to the differences in-phase and orthophase currents according to equations (1), (4), (5), (6).

производится в функции фазы синхронизации φ_s, образованной косвенно измеренными функциями cosφ_s,sinφ_s фазы φ_s потокосцепления ротора

в соответствии с уравнениями (9), (8), (3), (2) по шагам в последовательности операций от измеренных двухфазных величин i_sa, i_sb, U_sa, U_sb к измереным приращениям ψ_sα,ψ_sβ,ψ_rα,ψ_rβ,ψ_r,cosφ_s,sinφ_s и далее по шагам операций (10) - (13).Current regulation

is performed as a function of the synchronization phase φ _s , formed indirectly by the measured functions cosφ _s , sinφ _{s of the} phase φ _s rotor flux linkage

in accordance with equations (9), (8), (3), (2) in steps in the sequence of operations from the measured two-phase quantities i _sa , i _sb , U _sa , U _sb to the measured increments ψ _sα , ψ _sβ , ψ _rα , ψ _rβ , ψ _r , cosφ _s , sinφ _s and further in the steps of operations (10) - (13).

уменьшаются на величины падений напряжения u_sα,u_sβ и согласно уравнениям (2) определяют приращения двухфазного потокосцепления статора R_si_sα,R_si_sβ за интервал времени шага Δψ_sα,Δψ_sβ как интеграл разности напряжения и соответствующего падения напряжения за интервал времени Δt
Сущность способа оптимального векторного управления заключается в том, что измеренная описанным выше способом амплитуда ортофазного тока I_ort, полученная в результате двух замкнутых последовательностей операций над измеренными величинами фазных токов и фазных напряжений, задает оптимальный процесс изменения амплитуды потокосцепления ротора Δt. и частоты скольжения ψ_{r onm} путем пропорционального амплитуде I_ort изменения амплитуды синфазного тока и частоты скольжения Δω_onm из условия выполнения оптимизации режимов асинхронного электродвигателя, характеризуемых минимумом тока, минимумом потерь электроэнергии, максимумом момента и максимумом быстродействия, как показано на фиг. 4 - 7.The inverse transformations of the measured phase voltages U _sa , U _{sb are} performed similarly to (12)

The values of two-phase voltage measured according to (14)

decrease by the magnitude of the voltage drops u _sα , u _sβ and, according to Eqs. (2), determine the increments of the two-phase stator flux linkage R _s i _sα , R _s i _sβ over the step time interval _{Δψ sα} , _{Δψ sβ} as the integral of the voltage difference and the corresponding voltage drop over the time interval Δt
The essence of the optimal vector control method is that the orthophase current amplitude I _ort measured by the method described above, obtained as a result of two closed sequences of operations on the measured values of phase currents and phase voltages, sets the optimal process for changing the rotor flux coupling amplitude Δt. and the slip frequency ψ _{r onm} by proportional to the amplitude I _{ort of the} change in the amplitude of the common-mode current and the slip frequency Δω _onm from the condition for optimizing the modes of the induction motor, characterized by a minimum of current, a minimum of power losses, a maximum of torque and a maximum of speed, as shown in FIG. 4 - 7.

Optimal-controlled electromagnetic reaction of an induction motor to the interaction of currents and voltages described above is used in the method to optimize motor modes by interconnecting actions on the in-phase and orthophase current amplitudes I _syn , I _ort , as a result of which a typical asynchronous motor not equipped with sensors acquires new more efficient energy and dynamic properties (Figs. 4-7) both in an electric drive that is adjustable in moment or speed, and in an unregulated electric drive during operation with constant speed and variable load torque.

скорость вращения координат x, y,
ω_к - скорость вращения ротора,
в результате выполнения описанных выше операций принудительной ориентации и регулирования вектора тока статора ω - относительно вектора потокосцепления ротора

преобразуется к следующему виду, соответствующему достигнутой ориентации синфазного тока

оси "x" декартовой системы координат x, y вдоль вектора потокосцепления ротора

во всех статических и динамических режимах асинхронного электродвигателя

где

- амплитуда потокосцепления ротора,
ψ_r - угловая частота потокосцепления ротора,
ω_s - текущая скорость вращения ротора,
ω - угловая частота скольжения.Electromagnetic transient of a typical asynchronous electric motor, described by typical differential equations for a rotor circuit
Δω
where R _r , L _r - resistance and inductance of the rotor,
L _m - mutual inductance,

coordinate rotation speed x, y,
ω _to - the speed of rotation of the rotor,
as a result of performing the above operations of forced orientation and regulation of the stator current vector ω - relative to the rotor flux linkage vector

is converted to the following form corresponding to the achieved common-mode current orientation

x-axis of the Cartesian x, y coordinate system along the rotor flux linkage vector

in all static and dynamic modes of an induction motor

Where

- the amplitude of the flux linkage of the rotor,
ψ _r is the angular frequency of the flux linkage of the rotor,
ω _s is the current rotor speed,
ω is the angular slip frequency.

вычисляется как отношение измеренной амплитуды ортофазного тока I_ort к косвенно измеренной амплитуде потокосцепления ротора Δω согласно второму уравнению (16).According to the method, setting the common-mode current amplitude I _syn determines the rotor flux coupling amplitudes Δω according to the exponential law in accordance with the first equation (16) of an asynchronous electric motor with a time constant T _{r of} the flux coupling amplitude change
ψ _r
and the current actual value of the slip frequency

is calculated as the ratio of the measured amplitude of the orthophase current I _ort to the indirectly measured amplitude of the rotor flux linkage Δω according to the second equation (16).

и амплитуды потокосцепления ротора Δω как показано на фиг. 5. Для каждого момента M электродвигателя существует такое сочетание величин I_sin, I_ort, ψ_r, при котором достигается режим минимума тока i_s = i_{s min}. Естественная характеристика "ток - момент - скольжение" для неуправляемого асинхронного электродвигателя показана на фиг. 4 штриховой линией, сплошная линия, соединяющая точки минимума тока, соответствует режиму оптимального векторного управления согласно способу управления.In static modes of an induction motor, the amplitude of the rotor flux linkage is constant and equal to
ψ _r
the motor moment M is equal to the static load moment M _c , and the stator current amplitude ψ _r = L _m I _syn , (18) to create the moment M varies according to U-shaped characteristics depending on the values of the slip frequency

and the rotor flux coupling amplitudes Δω as shown in FIG. 5. For each moment M of the electric motor there is such a combination of the values of I _sin , I _ort , ψ _r , at which the minimum current mode i _s = i _{s min is} achieved. The natural current-moment-slip characteristic for an uncontrolled induction motor is shown in FIG. 4 by a dashed line, a solid line connecting the points of minimum current corresponds to the optimal vector control mode according to the control method.

As shown in FIG. 5, the optimal vector control mode is carried out by setting the moment of change in the main zone M = 0.25 - 2.0 M _{n to} set the common-mode current amplitude I _syn equal to the measured orthophase current amplitude I _ort , and this is also an optimality condition for higher values of the moment predominantly maintained with some slight deviation, as shown in FIG. 5.

The optimal definition of the common-mode current amplitude Δω, ψ _r proportional to the measured orthophase current amplitude I _ort according to the method stably holds the asynchronous motor mode at the current minimum points (Fig. 5), since the dynamic deviation of the common-mode current I _syn towards large values I _syn > I _{syn opt} moves the mode point to the area to the left of the optimal point in FIG. 5. In this regime region, the orthophase current decreases, which reduces the measured value of the orthophase current I _ort relative to a given value I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ _onm = I _{syn onm} and according to the method proportionally reduces the common-mode current I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ Following this, proportional-integral regulation of the amplitudes of the in-phase and orthophase currents restores the equality of the amplitudes of the in-phase and orthophase currents to the equality of the given and measured values and equality between themselves. Similar operations to compensate for deviations are also performed during increments in the right region from the optimum point.

Slip Frequency I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ . at an optimal level corresponding to the optimum ratio of the moment M to current i _s , according to the method, is changed automatically in accordance with equation (16) depending on the measured phase voltages and phase currents by calculating the indirectly measured amplitude of the flux linkage of the rotor Δω and measuring the amplitude of the orthophase current.

According to the method, at the beginning of the operation of the electric motor, the initial common-mode current amplitude I _{syn o is set} (Fig. 6) and the initial flux coupling of the rotor ψ _{r is} excited in the induction motor. Then, the moment M ^{* is set} .

изменяют пропорционально моменту M (фиг. 5)

При возрастании задания момента амплитуда ортофазного тока I_ort возрастает до равенства заданной начальной амплитуды синфазного тока I_syn, после чего амплитуду синфазного тока изменяют пропорционально измеренному ортофазному току I_ort, а угол фазового сдвига регулируют на постоянном уровне ε_φ= K_ε•M. (фиг. 5).In the case of a small value of the set moment M ^* <0.25 M _n (for example, in idle mode), the amplitude of the orthophase current I _{ort is} regulated proportionally to the set moment M ^* at a constant amplitude of the common-mode current I _{syn o} , and the phase shift angle ψ _ro . stator current vector ε _φ relative to the rotor flux link vector

change proportionally to the moment M (Fig. 5)

As the torque reference increases, the amplitude of the orthophase current I _ort increases to equal the specified initial amplitude of the common-mode current I _syn , after which the amplitude of the common-mode current is changed proportionally to the measured orthophase current I _ort , and the phase shift angle is controlled at a constant level ε _φ = K _ε • M. (Fig. 5).

увеличивают с увеличением момента M путем увеличения амплитуды синфазного тока I_syn как показано на фиг. 5. При этом нелинейное явление магнитного насыщения магнитопровода асинхронного электродвигателя в зоне сильного насыщения, необходимое для оптимизации тока при больших значениях момента M > 2M_н, в способе управления не моделируется, а точно измеряется путем измерения соотношений фазных напряжений и фазных токов, естественным путем связанных реальной характеристикой намагничивания асинхронного электродвигателя.The amplitude of the flux linkage of the rotor

increase with increasing moment M by increasing the in-phase current amplitude I _syn as shown in FIG. 5. In this case, the nonlinear phenomenon of magnetic saturation of the magnetic circuit of an asynchronous electric motor in the zone of strong saturation, which is necessary for optimizing the current at large values of the moment M> 2M _n , is not modeled in the control method, but is accurately measured by measuring the ratios of phase voltages and phase currents, naturally connected real characteristic of the magnetization of an induction motor.

In dynamic modes of operation of an induction motor (start, braking, a significant increase in speed, overload in time) according to the method limit the maximum value of the stator current i _{s max} at the level of permissible value i _{s extra} for the inverter supplying the motor, and perform the operations of optimal vector control described above by the criterion of the maximum moment M = M _max at i _{s max} = i _{s extra} , as shown in FIG. 6. The optimal value of the amplitude of the flux linkage of the rotor ψ _{r onm is} set the greater, the higher the permissible current i _{s add} . This is achieved by a corresponding optimal increase in the in-phase current amplitude I _{syn opt in} proportion to the measured orthophase current I _ort .

In the dynamic mode that occurs during a stepwise increment of the signal for setting the moment U _m , for example, at start-up (waveforms are shown in Fig. 7), the initial value of the rotor flux linkage ψ _r is small with respect to the optimal value ψ _ro for the maximum moment at the permissible current i _{s additional} . According to the method, the amplitude of the orthophase current ψ _{r onm is} proportional to the ratio of the given moment M ^* to the indirectly measured (calculated from the measured values) amplitude of the flux linkage of the rotor I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ variable according to the optimal law ψ _r , depending on the in-phase current amplitude I _{syn opt}
ψ _r = ψ _{r onm}
where Z _p - the number of pairs of poles of an induction motor.

, пропорционального заданному моменту M^* (фиг. 7), задание амплитуды ортофазного тока I_ort скачком увеличивают согласно уравнению (18), после чего пропорционально скачком увеличивают синфазный ток I_syn, в результате чего амплитуда потокосцепления ротора u $\genfrac{}{}{0ex}{}{\text{*}}{\text{м}}$ начинает резко возрастать в связи с форсировкой амплитуды синфазного тока. Косвенно измеренная нарастающая амплитуда потокосцепления ротора ψ_r(t) согласно уравнению по закону (18) снижает величину заданной амплитуды ортофазного тока ψ_r(t) и вслед за этим снижается амплитуда синфазного тока, как показано на фиг. 7.From time t _{0, the} supply of a jump in the control action

proportional to a given moment M ^* (Fig. 7), the specification of the amplitude of the orthophase current I _{ort is} jumpwise increased according to equation (18), after which the in-phase current I _{syn is} proportionally jumped, as a result of which the rotor flux coupling amplitude u $\genfrac{}{}{0ex}{}{\text{*}}{\text{m}}$ begins to increase sharply due to the boosting of the common-mode current amplitude. Indirectly measured increasing amplitude of the rotor flux linkage ψ _r (t) according to the equation according to the law (18) reduces the value of the specified amplitude of the orthophase current ψ _r (t) and then the common-mode amplitude decreases, as shown in FIG. 7.

Постоянный и максимальный момент M асинхронного электродвигателя в динамике вызывает постоянное и максимальное ускорение, в результате чего скорость вращения M(t) = M_max= M^*(t) = K_м•u_м(t). (20) изменяется с максимальным быстродействием линейно с максимальным постоянным ускорением (фиг. 7).Slip Frequency I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ also forced at the beginning of the transition process and then reduced to a steady level Δω (t) (Fig. 7). As a result of the dynamic forcing of the common-mode current I _syn (t) and the slip frequency Δω _onm , the speed of the increase in the rotor flux linkage amplitude Δω (t) increases, which increases with the dynamic time constant T _dyne two times smaller than the time constant T _r , determined with a constant increment of the common-mode current according to (19)
ψ _r (t),
With this method of optimal vector control of the dynamic mode, the actual moment of the induction motor M is proportional to the given moment M ^* and is maximum for the permissible stator current i _{s extra}

The constant and maximum moment M of the induction motor in the dynamics causes constant and maximum acceleration, as a result of which the rotation speed M (t) = M _max = M ^* (t) = K _m • u _m (t). (20) varies with maximum speed linearly with maximum constant acceleration (Fig. 7).

выше граничной скорости ω в статических и динамических режимах условие оптимальности по максимуму входной механической мощности (21) согласно способу осуществляется путем снижения величины заданной амплитуды синфазного тока ω_гр из условия режима работы на максимальном напряжении или на максимальном токе i_{s доп}.When the speed increases above a predetermined boundary value of the rotation speed ω (t), the maximum moment M ^* decreases inversely with the current rotation speed ω _gr from the condition of the maximum output dynamic power P _max
ω> ω _gr
under the following conditions, the restrictions on the maximum stator voltage and on the maximum stator current
P = M _max (ω) • Z _p • ω = P _max (21)
With increasing speed

above the boundary speed ω in static and dynamic modes, the optimality condition for the maximum input mechanical power (21) according to the method is carried out by reducing the value of the given common-mode amplitude ω _g from the condition of the operating mode at maximum voltage or at maximum current i _{s add} .

Three-zone regulation of the torque and speed of an induction motor according to the method is carried out by optimal setting and regulation of the amplitude of the common mode current:
in the first zone I _{syn onm} (ω, I _ort )
in the second zone I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ = I _{syn o} ;
in the third zone I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ = K ₁ • I _ont ;
Three-zone optimal regulation of the common-mode current amplitude and, therefore, the magnitude of the flux linkage amplitude of the rotor I $\genfrac{}{}{0ex}{}{\text{*}}{\text{syn}}$ = (K ₁ I _ort , K ₂ ω). minimum energy losses are achieved in static and dynamic modes, the expansion of the torque and speed control ranges above the nominal values of M _n , ψ _{r onm} , and an increase in the output mechanical power of the asynchronous electric motor ω _n when voltage and current restrictions are met according to condition (22).

 The object of the invention, which is to minimize energy losses in an induction motor due to optimal vector control of an induction motor without sensors on the motor, is solved in the proposed solution to the method of optimal control and electric drive for its implementation by the described sequence of operations and interactions over the currents and voltages of the stator of an induction motor. The sequence of operations is carried out in such a way that without measuring the rotation speed and magnetic flux and using only the reaction of the control object itself (asynchronous electric motor), which changes the ratio of phase currents and phase stator voltages, the asynchronous electric motor self-regulates according to the optimal laws of maximum torque to current ratio, minimum consumption electricity, maximum speed while providing the ability to obtain the limit values of the moment and power on the asyn shaft ronnogo motor.

The purpose of the invention is the minimization of energy losses, improving the accuracy of regulation of optimal conditions taking into account saturation of the magnetic circuit while expanding the range of regulation of the moment and speed of an asynchronous electric motor without sensors on the electric motor, is achieved by the fact that the maximum torque-to-current ratio M / i _{s max} , depending on the moment M, the increase in the amplitude of the flux linkage of the rotor (M • Z _p • ω) with increasing moment M occurs with a natural allowance for the actual saturation of the magnetic circuit of an induction motor according to the measured values of the phase voltages U _sa , U _sb and phase currents i _sa , i _sb similar to the typical measurement of the real magnetization characteristic of an asynchronous voltage-current electric motor, adopted in type tests of electric machines.

где J - момент инерции электропривода.The expansion of the regulation range of the "sensorless" electric drive is achieved by the fact that the direct drive channel for setting and controlling the frequency of the current or voltage of the stator is excluded in the drive, and the speed is set and controlled by self-regulation directly by the control object (asynchronous electric motor) according to the measured values of phase currents and phase voltages and given moment M ^* . Self-regulation of the current frequency is carried out by measuring the increments of the difference between the current magnitude and the stator voltage when an electric motor moment M occurs, resulting in an increment of the mechanical rotation speed ψ _{r onm} (M) according to the equation of motion

where J is the moment of inertia of the electric drive.

начиная от нулевой механической скорости вращения ω_мех, = 0 (в режиме "упора") до максимальной физически предельно достижимой величины ω_мех, ограниченной максимальным напряжением U_{s max}.The increment of the mechanical rotation speed (MM _c ) Δt = JΔω _{mech of the} control object creates an increment of the measured values of the difference in phase voltages and phase currents affecting the parameters of the current vectors and the rotor flux link vector Δω _mech, which increases the accuracy and range of regulation of the mechanical rotational speed

starting from zero mechanical rotation speed ω _mech , = 0 (in the “stop” mode) to the maximum physically maximum achievable value ω _mech , limited by the maximum voltage U _{s max} .

The dynamic and static accuracy of speed control is increased by compensating for disturbances on the load side by measuring the response of an induction motor to increments of the load moment and rotation speed, which increases the rigidity of the mechanical characteristics ω _mech = ω _{mech max} (M _s ) and decreases the control error ω _mech Dynamic control error ω ^* -ω. It also decreases due to the achievement of the maximum maximum dynamic moment for a given maximum current i _{s extra} and maximum acceleration of the electric drive, with a minimum time compensating for the mismatch between the set and current rotation speed.

The technical result and the effectiveness of technical solutions in various fields of technology are to obtain a set of positive technical and economic effects:
minimization of energy losses in a typical asynchronous electric motor without sensors on the electric motor by reducing electric losses by 20-25% relative to known electric drives by reducing losses on the excitation of the magnetic field at low load moments and the optimal increase in magnetic flux with increasing torque, providing maximum torque to current ratio;
with increasing control ranges, increasing the accuracy of controlling the optimal magnetic flux, minimum current and maximum torque modes by increasing the accuracy of accounting for the nonlinear magnetic saturation of the magnetic circuit of an asynchronous electric motor based on the interaction of voltages and currents;
universality of adjusting properties and applications of an asynchronous electric drive due to the exclusion of sensors structurally connected to the electric motor and due to the simultaneous increase in the range of speed control under load down, up to zero speed in the stop mode and up, up to 2-4 times the nominal speed
reduction of stator current at rated load by about 10% and total power loss by 15%;
reduction of voltage losses in power cables, including for long supply lines of an asynchronous electric motor, for example, for deep submersible electric pumps for oil production, by 30-40% in start-up, braking and overload modes;
the possibility of using the electric drive in high-precision and optimized torque electric drive modes, for example, in electric vehicles, for electric vehicles, without using sensors on the engine;
the possibility of using an electric drive solely for the purpose of saving electricity, including for unregulated mechanisms.

Industrial applicability with the greatest economic benefit for consumers of the proposed technical solution instead of the known ones is feasible in the following machines and mechanisms:
electric drives of rocking machines and deep submersible electric pumps of oil wells (regulating forces and speed, saving energy, reducing voltage losses in the power cable, increasing the success of exiting lock and overload modes);
electric drives of pumping and fan installations, including in municipal services (reduction of electric energy losses);
in elevators, hoists and cranes (increasing the ratio of torque to current, increasing the accuracy and smoothness of torque and speed regulation, the possibility of creating the "stop under active load"mode);
traction electric drive of electric transport, including electric drive of electric vehicle, electric locomotive (the possibility of smooth regulation of torque, minimum current, accuracy of acceleration, maximum torque during overloads);
electric drives of the main movement of machines, an electrospindle (the maximum range of speed control without a sensor on the shaft, the maximum output of mechanical power, the minimum start and stop time);
Electromechanisms operating in radioactive, fire-explosive, liquid and chemically active environments, which use special high-protection asynchronous electric motors, excluding the use of sensors structurally associated with the electric motor.

 The versatility and effectiveness of the energy and regulatory advantages of the technical solution "The method of optimal vector control and electric drive for its implementation" and the distribution of asynchronous electric motors in various fields of technology allow its use in quantities exceeding 100,000 units per year.

Sources of information
1. RF patent N 1458951, cl. H 02 M 7/40 "A method for controlling a multiphase inverter and a device for its implementation, Mishchenko VA, Mishchenko NI Priority 03/26/1984. Published. 02/15/1989. Bull. N 6.

 2. RF patent N 1515322, cl. H 02 P 7/42 "AC electric drive", V. Mishchenko Priority 05/11/1984. Publ. 10/15/89. Bull. N 38.

Claims (3)

1. The vector control method of an induction motor, in which the stator windings are supplied with alternating current, the amplitude and phase angle of which are regulated relative to the rotor flux linkage vector, while the phase currents are equal to the difference synchronously changed with the synchronization frequency of the in-phase and orthophase currents, the set time is changed depending on the mismatch of the set and current speed, while measuring currents in the three phases of the stator, characterized in that they additionally measure the voltage in the three phases of the stator convert the currents and voltages measured in three phases into a two-phase system of in-phase and orthophase currents and voltages, then determine the actual amplitude of the in-phase and orthophase currents, and also calculate by integrating the difference of the converted voltages and the voltage drop proportional to the converted currents, the rotor flux linkage amplitude and the sine value and cosine functions of the rotor flux linkage phase, while depending on the actual amplitude of the orthophase current, the predetermined amplitude of the syn phase current from the condition of changing the phase angle of the stator current vector relative to the rotor flux vector in the range of more than 45 ^o , and the value of the specified orthophase current amplitude is changed proportionally to the ratio of the given moment and the calculated rotor flux link amplitude, the current rotation speed is calculated by summing the slip frequency with the synchronization frequency, equal to the normalized sine and cosine functions of the rotor flux linkage phase, in proportion to which the orthophase and common mode current are regulated and, the amplitude of which varies depending on the mismatch of a given and actual amplitudes of in-phase and orthophase currents.  2. A device for vector control of an asynchronous electric motor, comprising an inverter, the power outputs of which are connected via phase current sensors to the stator windings of the asynchronous electric motor, a block of current regulators connected to phase current sensors, an adaptive torque regulator, the input of which torque is connected to the output of the speed controller, a reference input connected to the speed reference unit, characterized in that the control inputs of the inverter through the control pulse generation unit and the regulator block t the eyes are connected to the outputs of the direct converter of two-phase-three-phase coordinates, the inputs of which are connected to the outputs of the direct converter of Cartesian coordinates, the orthophase and common mode inputs of which are connected to the outputs of the orthophase and common mode current regulator, the input of the orthophase current regulator is connected to the output of the adaptive torque regulator, to the power outputs a phase current sensor, a phase voltage sensor is connected, the outputs of which are connected to the first inputs of two adders, the second inputs of which are connected to the outputs of the reverse a two-phase-three-phase coordinate converter, inputs connected to phase current sensors, the outputs of these adders are connected to the inputs of two integrators, the outputs are connected to the first inputs of two other adders, the second inputs of which are connected to the outputs of the inverse two-phase-three-phase coordinate converter, the outputs of the adders are connected to the first inputs two dividers and two inputs of the rotor calculation and flux linkage unit, the output of which is connected to the second inputs of the dividers, the outputs of the dividers are connected to the sine and the cosine inputs of the forward and reverse Cartesian coordinates transducers, the two inputs of the Cartesian coordinate inverters are connected to the outputs of the two-phase-three-phase coordinates inverter, and the orthophase and common-mode outputs are connected to the feedback inputs of the orthophase and common-mode current regulators, the orthophase output of the Cartesian coordinates inverse is connected a module isolation unit with an input of the optimal common-mode current control unit defining an optimal mode according to energy criteria and changes in the amplitude of the common-mode current depending on changes in the amplitude of the orthophase current, the output connected to the master input of the common-mode current controller, and the second input to the second output of the adaptive torque controller, while the orthophase output of the inverse Cartesian coordinate converter is connected to the first input of the slip frequency calculation unit, the second the input of which is connected to the output of the rotor flux linkage calculation unit, the sine and cosine inputs of the forward and reverse Cartesian coordinate converters are connected with the inputs of the synchronization frequency calculation unit, the output connected to the first input of the frequency adder, the second input of which is connected to the output of the slip frequency calculation unit, and the output is with the feedback input of the speed controller and the second input of the adaptive torque controller, the third input of which is connected to the output of the calculation unit the amplitude of the flux linkage of the rotor. 3. A device for vector control of an asynchronous electric motor, comprising an inverter, the power outputs of which are connected via phase current sensors to the stator windings of the asynchronous electric motor, and the control inputs are connected to a control pulse generation unit, characterized in that a phase voltage sensor is connected to the inverter power outputs digital signal coprocessor, digital signal processor, programmable digital memory, serial port, remote control and diagnostician and, in this case, the control pulse generation block is made in the form of a driver microcircuit with an additional pulse-width modulation resolution input, the inputs of which are connected to the output of a digital signal coprocessor, formed by the outputs of a timer, which performs pulse-width modulation of digital mismatch codes of given and measured currents, analog inputs which, connected to the outputs of the phase voltage and current sensors, are formed by the inputs of the integrated multiplex analog-to-digital digitizer a new signal coprocessor containing an integrated serial command programmer, an integrated vector conversion unit in which the direct and inverse transformation of two-phase-three-phase coordinates, direct and inverse transformation of Cartesian coordinates and regulation of common-mode and orthophase currents, and a register control unit, interconnected with the built-in multiplex analog-to-digital converter and timer, as well as with a digital signal processor and programmable read-only memory, data and the address bus, the digital signal processor is connected by inputs / outputs to the inputs / outputs of the serial port through which information is exchanged with the program control and diagnostics panel, while the digital signal processor implements an algorithm of optimal vector control of the motor torque and speed at the optimal energy change criteria ψ _r = ψ _ronm in accordance with the equations
I $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ = 2/3-L _r / Z _p L _m M ^* / ψ _ronm ,
L _r / R _r dψ _r / dt + ψ _r = L _m I _syn ,
Δω = ω _s -ω = R _r / L _r L _m I _ort / ψ _r ,
where i $\genfrac{}{}{0ex}{}{\text{*}}{\text{ort}}$ - a given amplitude of the orthophase current;
I _syn - common-mode current amplitude;
R _r and L _r - resistance and inductance of the rotor;
L _m is the mutual inductance;
Z _p - the number of pairs of poles of an induction motor;
M ^* is the given moment;
ψ _r is the amplitude of the flux linkage of the rotor;
ω is the current angular velocity of rotation of the rotor;
ω _s is the angular frequency of rotation of the flux linkage of the rotor;
Δω is the angular slip frequency.

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