RU2072525C1 - Directivity pattern shaping method - Google Patents

Abstract

FIELD: shaping multibeam directivity patterns of radar and sonar receiving antennas, including echo sounding for ocean investigations. SUBSTANCE: method involves reception of signal by means of planar antenna array, sampling of output signals from array antennas, and two-stage shaping of directivity pattern beams separately in bearing and elevation planes, setting of angles ψ and q in bearing and elevation planes, respectively; along with sampling, directivity pattern beam for angle j is shaped and antenna array is corrected for angle q, whereupon set of beams is shaped in elevation plane and then readings of complex signal envelope for each set of beams are separated. EFFECT: improved accuracy of directivity pattern shaping and reduced cost of computing operations. 2 cl, 11 dwg, 1 tbl

Description

 The invention relates to methods for generating a multi-beam pattern of receiving radar and sonar antennas and, in particular, is intended for use in oceanographic studies by the method of multi-beam echo sounder.

 In multipath echo sounders, it is necessary to ensure the possibility of receiving signals reflected from the bottom and hydrophysical inhomogeneities from a set of directions in space within a certain sector [1] for which the radiation pattern at reception should have the form of a fan of narrow rays.

 A known method of forming a radiation pattern of this kind [2, p. 295 296] which consists in receiving the reflected signal through the antenna array, the time delay of the output signals of the array elements and their subsequent summation. The disadvantage of this method is the difficulty of its practical implementation, due to the presence of technical problems, which are mainly associated with the need to implement a large number of stable, identical and tunable analog delay lines.

где Y_m результат формирования луча в азимутальной плоскости;
α_m•β_n= ω_mn весовая функция антенной решетки (амплитудное распределение);
τ₁= d_x•(C)^-1sinθ;, τ₂= d_y•(C)^-1sinΨ пространственные задержки фронта волны между элементами соответственно в угломестной и азимутальной плоскостях;
d_x расстояние между элементами антенной решетки в угломестной плоскости (ZOX на фиг. 1), d_y расстояние между элементами антенной решетки в азимутальной плоскости (ZOY на фиг. 1), С скорость распространения волны-переносчика сигнала S(t);
θ,Ψ- соответственно угол места и азимут для выбранного луча, т. е. углы между вертикальной осью (Z на фиг. 1) и проекциями вектора

, в направлении которого формируется луч, на угломестную и азимутальную плоскости.Also known is a method of forming a radiation pattern [3, p. 110] adopted as a prototype, which consists in receiving the signal S (t) (the carrier of which is an acoustic or electromagnetic wave) through a flat antenna array, sampling the output signals of the antenna array elements, two-stage beam formation separately in the azimuthal and elevation planes. According to this method, the output signal x _mn (t) of an arbitrary element with the number (m, n) of the antenna array of size M * N elements (see Fig. 1) by sampling with a frequency F _{d is} converted into a time series x _mn (lΔ) of samples (where l is the reference number), i.e., samples x _mn (t) taken with a sampling step Δ = (F _d ) ^-1 . The beam formation in the known method is based on the compensation of time delays between the moments of arrival of the front of the received wave on the grating elements separately in the azimuthal and elevation planes, by delaying the samples x _mn (lΔ) by a time multiple of Δ .. The output signal b (lΔ ) of any ray from the generated set of beams of the radiation pattern is also a time series, for an arbitrary l-th reference of which the relation

where Y _{m is the} result of beam formation in the azimuthal plane;
α _m • β _n = ω _mn weight function of the antenna array (amplitude distribution);
τ ₁ = d _x • (C) ^-1 sinθ ;, τ ₂ = d _y • (C) ^-1 sinΨ spatial delays of the wave front between elements in the elevation and azimuthal planes, respectively;
d _{x the} distance between the elements of the antenna array in the elevation plane (ZOX in Fig. 1), d _{y the} distance between the elements of the antenna array in the azimuthal plane (ZOY in Fig. 1), C the propagation speed of the signal carrier wave S (t);
θ, Ψ - respectively, elevation and azimuth for the selected beam, i.e., the angles between the vertical axis (Z in Fig. 1) and the projections of the vector

in the direction of which the beam is formed, on the elevation and azimuthal plane.

The disadvantage of this method is the limited accuracy of beamforming. The accuracy limitation is due to the fact that this method allows the formation of only "synchronous" beams, i.e. such for which the values of τ ₁ and τ ₂ are multiples of the sampling step Δ .. The decrease in the accuracy of the diagram formation is especially noticeable when the sampling is performed with a frequency substantially lower than the central frequency S (t) (the case of discretization of the complex envelope or subsampling, [3, 4] ) However, even if the sampling frequency exceeds the center frequency of the received signal (which requires unjustified computational costs when implementing the method), the decrease in accuracy cannot be neglected. In particular, if the sampling frequency is four times higher than the center frequency, and the distance between the elements of the antenna array is equal to half the wavelength, then the discreteness of the installation of the rays is approximately 30 deg.

 Thus, the fact that the known method does not allow the formation of radiation patterns in predetermined directions (if these rays are not “synchronous”) makes it practically not applicable in situations where it is necessary to smoothly scan the radiation pattern or compensate for changes in spatial orientation receiving antenna.

 The objective of the invention is to develop a method of forming a radiation pattern that allows you to create a radiation pattern in the form of a fan of narrow rays.

 The technical result from the use of the invention is to increase the accuracy of diagramming while reducing computational costs.

 This result is achieved by the fact that in the method of forming the radiation pattern, which consists in receiving a signal through a flat antenna array, sampling the output signals of the elements of the antenna array and two-stage beam formation of the radiation pattern separately in the azimuthal and elevation planes, some angles Ψ and θ, respectively, in the azimuthal and elevation planes, simultaneously with sampling, beamforming along the angle Ψ and antenna compensation are performed of the array along the q angle, for which, when sampling the output signals of adjacent elements of the antenna array, they are obtained with a time shift, the value of which is selected based on the angle Ψ for elements having the same coordinates in the elevation plane, and based on the angle q for elements having the same coordinates in the azimuthal plane, and form a vector Y of real discrete signals, each component of which is obtained by weighting the sum of the samples obtained from the output signals of those elements comrade antenna array that have the same coordinates in the elevation plane, then produce the formation of a beam set in the elevation plane, and then carried out the allocation of samples of the complex envelope signals received on each of the beam set.

 The specified technical result is achieved in addition to the fact that the formation of each of the set of rays in the elevation plane is carried out by summing the components of the vector Y with complex weighting factors.

 The essence of the invention consists, firstly, in eliminating the influence of the sampling frequency on the beamforming in the azimuthal plane and the accuracy of compensation of the antenna array in the elevation plane, and secondly, in the use of real samples to form a set of rays in the elevation plane.

 Moreover, setting angles J in the azimuthal plane and q in the elevation plane is necessary to ensure the correct spatial orientation of the radiation pattern being formed. In particular, when using the proposed method for forming the radiation pattern of the oceanographic complex, the angles J and θ can be respectively the angles of the keel and side rolling of the carrier vessel.

 Simultaneously with the sampling of beamforming along the angle J and compensation of the antenna array at the angle q, it is necessary to exclude the influence of the sampling frequency on the accuracy of the beamforming in the azimuthal plane and the accuracy of the compensation of the antenna array in the elevation plane (which, in particular, allows sampling without loss of accuracy with a low frequency).

 The execution of this operation can be based on obtaining samples from the output signals of adjacent elements of the antenna array with a time shift determined by the angles J and θ for elements that have the same coordinates, respectively, in the elevation and azimuthal planes, and the subsequent formation of the vector Y of real discrete signals, each component of which is the result of weighted summation of the samples obtained from the output signals of those elements of the antenna array whose coordinates are in the elevation plane are the same.

, которому соответствуют углы Ψ в азимутальной плоскости ZOY и q в угломестной плоскости ZOX. Если в некоторый момент времени t₀ 0 фронт волны достигает "углового" элемента решетки, имеющего номер (m_c, n_c), тогда при любых

элемента с произвольным номером (m, n) указанный фронт достигнет в момент времени

где τ₁= d_x•(C)^-1sinθ;τ₂= d_y•(C)^-1sinΨ подлежащие компенсации задержки распространения фронта волны соответственно в угломестной и азимутальной плоскостях;
d_x, d_y расстояние между элементами решетки соответственно в угломестной и азимутальной плоскостях;
С скорость распространения волны;

При этом задержка распространения фронта волны равна τ₂ для любых соседних элементов с номерами (m, n) и (m, n + 1) и τ₁ для любых соседних элементов с номерами (m, n) и (m + 1, n). Другими словами, указанная задержка определяется значением угла ψ для любых соседних элементов, имеющих одинаковые координаты в угломестной плоскости, и значением угла q для любых соседних элементов с одинаковыми координатами в азимутальной плоскости.Let a plane wave coming from the direction be received by means of a flat antenna array with the size of M * N elements (see Fig. 1)

corresponding to angles Ψ in the azimuthal plane ZOY and q in the elevation plane ZOX. If at some point in time t ₀ 0 the wave front reaches the "angular" lattice element having the number (m _c , n _c ), then for any

element with an arbitrary number (m, n), the specified edge will reach at time

where τ ₁ = d _x • (C) ^-1 sinθ; τ ₂ = d _y • (C) ^-1 sinΨ subject to compensation of the propagation delay of the wave front, respectively, in the elevation and azimuthal planes;
d _x , d _{y the} distance between the elements of the lattice, respectively, in elevation and azimuthal planes;
With the speed of wave propagation;

Moreover, the propagation delay of the wave front is τ ₂ for any neighboring elements with numbers (m, n) and (m, n + 1) and τ ₁ for any neighboring elements with numbers (m, n) and (m + 1, n) . In other words, this delay is determined by the value of the angle ψ for any neighboring elements having the same coordinates in the elevation plane, and the value of the angle q for any neighboring elements with the same coordinates in the azimuthal plane.

Представим последнее выражение в эквивалентном виде

где x_mn(t) выходной сигнал элемента антенной решетки, имеющего номер

β_n коэффициент амплитудного распределения в азимутальной плоскости, т.е. весовой коэффициент для х_mn(t);
Y_m выходной сигнал m-й (при θ>0) или (М-1-m)-й (при θ<0) линейной подрешетки;
δ(t) дельта-функция (функция Дирака).According to (1.a) in two-stage beamforming, the result of beam formation in the azimuthal plane Y _m is the output signal of the m-th linear sublattice, consisting of elements with numbers from (m, 0) to (m, N-1), those. having the same coordinates in the elevation plane. Moreover, for an arbitrary lth sample of the indicated signal (taking into account the compensation of the antenna array in the elevation plane by the angle q),

We represent the last expression in equivalent form

where x _mn (t) is the output signal of the antenna array element having the number

β _n the amplitude distribution coefficient in the azimuthal plane, i.e. weight coefficient for x _mn (t);
Y _{m the} output signal of the mth (for θ> 0) or (M-1-m) th (for θ <0) linear sublattices;
δ (t) delta function (Dirac function).

As follows from (2), if we summarize the weighted samples obtained from the output signals of an arbitrary linear sublattice with a time shift (so that the sampling time of the signal of an arbitrary element coincides with the moment it reaches the received wavefront), then the summation result at time t _m , which is distant from the time t _{0 of} taking a sample from the signal of the "angular" element for the time interval m • τ ₁ + (N-1) • τ ₂ , is a reference of the output signal Y _{m of} this sublattice, which is received from the beam formed in the azimuthal plane ty, from the direction given by the angle θ in the elevation plane. In this case, the shift between the sampling times for adjacent elements of any sublattice is equal to t ₂ (selected based on the value of the angle Ψ). For elements in which the azimuthal coordinates are the same and the elevation coordinates differ by d _x , i.e. located in neighboring sublattices, is t ₁ (i.e., it is selected based on the value of the angle θ).

в момент времени t_m. Поэтому и сам отсчет Y_m, как сумма выборок из синфазных сигналов, также представляет собой выборку из волнового фронта.Thus, each sample of the output signal Y _{m of the} corresponding (m-th or (M-1-m) -th) linear sublattice located in the azimuthal plane is the sum of weighted samples from the wavefront of the signal received by it from the direction

at time t _m . Therefore, the sample Y _{m itself} , as the sum of samples from in-phase signals, also represents a sample from the wavefront.

Therefore, the set of samples Y _m (lΔ), m = 0 ,,, M-1 at time t _v = t ₀ + (M-1) • τ ₁ + (N-1) • τ ₂ can be considered as a vector Y (lΔ) of l samples of the output signals of the elements of the equivalent linear sublattice, which is located in the elevation plane and compensated by the angle θ .. Such a sublattice consists of elements that are directive in the azimuthal plane and not directional in the elevation plane, while the formation of the vector Y reduces to storing (delaying) the samples of each of the discrete signals Y _m for the time interval (M-1-m) • τ ₁ ..

As follows from the foregoing, with the described combination of discretization with the formation of the beam pattern of the antenna array along the angle Ψ and its compensation along the angle q, both the accuracy of beam formation in the azimuthal plane and the accuracy of the compensation of the array in the elevation plane do not depend on the sampling step, but they are determined only by the accuracy of setting the shifts τ ₁ , τ ₂ between the moments of sampling from the output signals of the elements of the antenna array.

которое означает, что время распространения фронта волны, приходящей с направления, соответствующего этим углам, вдоль антенной решетки, не должно превышать шага дискретизации. Однако наличие данного ограничения в случае дискретизации принимаемого сигнала с низкой частотой (т.е. дискретизации комплексной огибающей) не приводит к существенному ограничению диапазона углов, в пределах которого возможно формирование диаграммы направленности. Например для квадратной периодической решетки, у которой N M, d_x d_y d и d равно половине длины волны λ, из (3) следует

где F_O, F_д соответственно центральная частота принимаемого сигнала S(t) и частота дискретизации;
δθ ширина луча диаграммы направленности, расположенного по нормали к раскрыву антенны, определенная в радианах, в соответствии с [5] как отношение l к апертуре антенны (М-1)•d.The only requirement that must be met in this case is to preserve the natural sequence of samples of the vector Y with a sampling step Δ. The simplest possible way to satisfy this requirement is to impose the following restriction on the maximum values J _max , θ _{max of the} angles Ψ and θ:

which means that the propagation time of the wave front coming from the direction corresponding to these angles along the antenna array should not exceed the sampling step. However, the presence of this limitation in the case of sampling the received signal with a low frequency (i.e., sampling the complex envelope) does not lead to a significant limitation of the range of angles within which beamforming is possible. For example, for a square periodic lattice in which NM, d _x d _y d and d is equal to half the wavelength λ, it follows from (3)

where F _O , F _d respectively the Central frequency of the received signal S (t) and the sampling frequency;
δθ the beam width of the radiation pattern normal to the aperture of the antenna, defined in radians, in accordance with [5] as the ratio of l to the aperture of the antenna (M-1) • d.

.In particular, if δθ = 0.1 rad. (≃5 ^° ), and the center frequency exceeds the sampling frequency by 10 times, then the radiation pattern can be formed in the range of angles

.

 The implementation of the operation of forming a set of rays in the elevation plane is necessary for the final formation of the radiation pattern of the receiving antenna array in the form of a fan of narrow rays.

 The implementation of this operation directly after the discretization operation is necessary so that regardless of whether the output signals of the antenna array elements are sampled at a high frequency or their complex envelopes are discretized (sub-sampling), the samples of the actual discrete signals should be used to form a set of rays.

 This operation can be based on the formation of each of the set of rays in the elevation plane by summing the components of the vector of real samples Y obtained as a result of discretization with complex weighting factors. The specified vector, as already noted, is essentially a sample vector of the output signals of the equivalent linear sublattice located in the elevation plane. Therefore, the operation of forming a set of beams of a flat antenna array in the elevation plane is completely equivalent to forming a set of beams by means of an equivalent linear sublattice.

где α_m амплитудный весовой коэффициент (амплитудное распределение в угломестной плоскости);
φ_k= 2πd(λ)^-1•sinθ_k фазовый множитель для k-го луча диаграммы направленности, d расстояние между элементами, λ длина волны, q_k- угол, соответствующий направлению, в котором формируется k-й луч диаграммы направленности, М число элементов;

l-й отсчет комплексной огибающей выходного сигнала m-го элемента (l любое целое в пределах от -∞ до +∞).A common way of forming a beam of a linear array radiation pattern when receiving narrowband signals is phase [6]. It consists in compensating for the phase differences of the output signals of its elements arising from the delay in the arrival of the front of the received wave to these elements (it is assumed that due to the narrow-band signals the delay negligibly affects the change in their complex envelopes). In this case, the formation of each of the rays is reduced to summing the samples of the complex envelope of the output signals of all elements with complex weighting coefficients, and the result of the chart formation (output signal of the beam) is a discrete signal in the form of a sequence of complex samples

where α _{m is the} amplitude weight coefficient (amplitude distribution in the elevation plane);
φ _k = 2πd (λ) ^-1 • sinθ _k phase factor for the kth beam of the radiation pattern, d distance between the elements, λ wavelength, q _k - angle corresponding to the direction in which the kth beam of the radiation pattern is formed, M number of elements;

lth sample of the complex envelope of the output signal of the mth element (l any integer ranging from -∞ to + ∞).

, которому соответствуют углы Ψ_k и θ_k в азимутальной и угломестной плоскостях, поступает плоская волна -переносчик узкополосного сигнала

или

где

комплексная огибающая S(t), * знак комплексного сопряжения, ω_o круговая частота, соответствующая центральной частоте F_oS(t).Let on a flat antenna array (see Fig. 1) from an arbitrary direction

corresponding to the angles Ψ _k and θ _k in the azimuthal and elevation planes, a plane wave arrives - a transmitter of a narrow-band signal

or

Where

complex envelope S (t), * sign of complex conjugation, ω _o circular frequency corresponding to the center frequency F _o S (t).

где

.Then, by virtue of (2), an arbitrary component Y _{m of the} vector Y is a sequence of samples from the narrow-band signal Y _m (t). This signal, taking into account the beam formation during sampling, in the azimuthal plane in the direction of the angle Ψ and compensation of the antenna array in the elevation plane in the direction of the angle q, according to (2) can be represented as follows:

Where

.

где

комплексная огибающая рассматриваемого сигнала; φ_k= ω_o•ε_k пространственный сдвиг фаз в угломестной плоскости;

комплексная константа (для фиксированного Ψ_k), модуль которой

, согласно [3] есть диаграмма направленности линейной подрешетки, расположенной в азимутальной плоскости, луч которой сформирован в направлении угла Ψ..Due to the narrowband S (t) (when the functions α (t) and ξ (t) are slowly changing, and according to the phase method of diagram formation for any m, n from the intervals 0≅m≅M-1, 0≅n≅N-1, we can take :
α (t + nν _k + mε _k ) ≃ α (t); ξ (t + nν _k + mε _k ) ≃ ξ (t) for y _m (t) holds:

Where

complex envelope of the considered signal; φ _k = ω _o • ε _k spatial phase shift in the elevation plane;

complex constant (for fixed Ψ _k ) whose module

, according to [3], there is a directivity pattern of a linear sublattice located in the azimuthal plane, the beam of which is formed in the direction of the angle Ψ ..

где l произвольное целое, Ω₀= ω₀•Δ дискретная угловая частота, а

является выборкой из комплексной огибающей

и, в случае θ_k= θ(φ_k= 0); Ψ_k= Ψ, с точностью до амплитуды совпадает с выборкой из комплексной огибающей S(t).Therefore, the discrete signal Y _m (lΔ) has the form:

where l is an arbitrary integer, Ω ₀ = ω ₀ • Δ is the discrete angular frequency, and

is a sample of the complex envelope

and, in the case θ _k = θ (φ _k = 0); Ψ _k = Ψ, up to amplitude, coincides with a sample from the complex envelope S (t).

суммирования всех М компонент вектора Y с комплексными весовыми коэффициентами ω_m,k= α_m•exp(-jmφ_k) выражается следующим образом:

где

и представляет собой сумму двух дискретных аналитических сигналов. Огибающая

первого из них (дискретный спектр которого сосредоточен в окрестности "поднесущей" Ω₀), согласно (4), есть выходной сигнал k-го луча, формируемого в угломестной плоскости фазовым способом. При этом

представляет собой последовательность отсчетов комплексной огибающей сигнала, принимаемого по этому лучу.In this case, the result

summation of all M components of the vector Y with complex weighting coefficients ω _{m, k} = α _m • exp (-jmφ _k ) is expressed as follows:

Where

and is the sum of two discrete analytical signals. Envelope

the first of them (the discrete spectrum of which is concentrated in the vicinity of the "subcarrier" Ω ₀ ), according to (4), is the output signal of the k-th beam formed in the elevation plane by the phase method. Wherein

is a sequence of samples of the complex envelope of the signal received on this beam.

 Therefore, the summation of the components of the vector Y with complex weights really provides the formation of a set of rays in the elevation plane. Note that in this case the multiplication of real, rather than complex, as required by (4), samples by complex weighting coefficients is performed, i.e., half the number of multiplication operations is necessary to form the rays. It should also be noted that since the antenna array is compensated during sampling in the elevation plane, no other computational operations are required to form a beam in the direction of the given angle θ except for the weight summation of the actual samples (the weights are real and constant). It also contributes to a certain reduction in the amount of computation during diagram formation and, in particular, allows you to organize a fast scan mode "central" from a set (fan) of rays, the position of which is determined by the given angles J and θ .. In addition, in the scan mode with the whole fan in the elevation plane preliminary (performed at sampling) full compensation of the antenna array by the "central" beam leads to its partial compensation and other rays. With phase diagram formation in the elevation plane, this allows one to reduce the distortion of the received signals due to the complex envelope filtering effect inherent in this method [6], since the delay between the moments of arrival of the wave front on various elements of the antenna array, the effect of which on the complex envelopes is neglected in this method, partially offset by sampling.

).The selection of samples of the complex envelope of the signals received for each of the set of rays is necessary, firstly, to extract from these signals only useful (necessary and sufficient for reconstruction) information, which is contained in their amplitude and phase modulation; and secondly, to suppress interfering components that may appear as a result of diagram formation (for example, a component

)

, в результате чего формируется сигнал

и последующей низкочастотной фильтрации в двух квадратурах, при которой подавляется высокочастотная компонента

. Заметим, что когда выполняется условие

q- любое положительное целое, каждая квадратурная компонента

представляет собой последовательность единичных и нулевых отсчетов, поэтому умножение на

практически не требует вычислительных затрат [4]
Следует также подчеркнуть, что по предлагаемому способу необходимо выделять отсчеты комплексной огибающей К сигналов (где К число лучей в наборе). В секторе -90^°≅θ≅90^° можно сформировать (см. [3]), максимум M + 1 луч (при этом лучи пересекаются на уровне -3дБ). Однако на практике, как правило, сектор обзора в угломестной плоскости меньше 180^o, уровень перекрытия лучей меньше -3дБ и К < M. При этом предлагаемый способ требует меньших вычислительных затрат на выделение отсчетов комплексной огибающей, чем те (например [6]), в которых эти отсчеты используются при диаграммоформировании и, следовательно, выделяются по крайней мере из М (или даже M•N) сигналов.This operation can be, in particular, performed by synchronous detection. So, if the signal obtained as a result of diagram-forming is described by expression (6), the detection is reduced [4] to its multiplication by a complex reference sequence

, resulting in a signal

and subsequent low-pass filtering in two quadratures, in which the high-frequency component is suppressed

. Note that when the condition is satisfied

q- any positive integer, each quadrature component

is a sequence of unit and zero samples, therefore, multiplying by

practically does not require computational costs [4]
It should also be emphasized that according to the proposed method, it is necessary to select the samples of the complex envelope of K signals (where K is the number of rays in the set). In the sector of -90 ^° ≅θ≅90 ^° it is possible to form (see [3]), the maximum is M + 1 beam (in this case, the rays intersect at the level of -3dB). However, in practice, as a rule, the viewing sector in the elevation plane is less than 180 ^o , the level of overlap of the rays is less than -3dB and K <M. Moreover, the proposed method requires less computational cost for extracting the complex envelope samples than those (for example [6]), in which these samples are used in the formation of diagrams and, therefore, stand out from at least M (or even M • N) signals.

 It should also be noted that in any locator in each receive channel (beam), as a rule, a coordinated filtering of the complex envelope is carried out [3], ie, low-pass filtering. Moreover, the equality of the number of separators of the samples of the complex envelope and the number of rays allows you to combine filtering when selecting these samples with a consistent one, which also reduces the overall computational cost when implementing the proposed method.

 The combination of the above differences is sufficient to achieve the objectives of the invention.

 The applicant is not aware of the previously described methods for forming a radiation pattern with the above set of features.

 In FIG. 2 shows a general block diagram of a device that implements a beamforming method, an embodiment; in FIG. 3 6 are structural diagrams of individual blocks included in the general structural diagram of FIG. 2 (in this case, the analog signal transmission buses are shown by thin lines, digital thickened, complex double); in FIG. 7 9 timing charts; in FIG. 10 spectral characteristics of the signals at some points of the device that implements the proposed method.

 The device of FIG. 2 contains a flat equidistant antenna array 1, consisting of M • N elements 2 (on the aperture of the array in the azimuthal plane there are N elements, in the elevation plane M), M sampling units (DB) 3, a pulse distributor 4, K (according to the number K simultaneously formed rays) of the beam forming units (BFL) 5, K of the blocks for the allocation of samples of the complex envelope (BVCO) 6 and the control unit (BU) 7.

 In this case, the outputs of any m-x (m 1.M) N elements having the same coordinates in the elevation plane are connected to information inputs, respectively, from the first to the N-th m-th database 3, so that the output of an arbitrary lattice element having the number ( m, n) is connected to the (nn + 1) th information input of the (mm + 1) th DB (Fig. 2 shows the connection of the first and Mx N elements, and the numbering of the elements corresponds to Fig. 1 and the expressions (1 ) (6)). The first control inputs of all database 3 are combined and connected to the first output of control unit 7, the second control inputs of all database 3 are combined and connected to the second output of control unit 7, the third control inputs of all database 3 are combined and connected to the third output of control unit 7, the fourth control inputs of all The DB 3 is also biased and connected to the sixth output of the BU 7, which, in addition, is connected to the start input of the pulse distributor 4 and the combined control inputs of all BVCO 6. Sample units 3 are connected to the outputs by their fifth control inputs I 4 pulses, respectively, the first to M-th, second to (M 1) -th. mth to (M + 1 - m) th. Mth to the first. The (M + 1) -th output of the pulse distributor 4 is connected to its reset input, and its clock input and mode input are connected respectively to the fourth and fifth outputs of the control unit 7. The output of an arbitrary m-th database 3 is connected to the combined m-information inputs of all BFL 5, the control inputs of which are combined with the second input of the control unit 7. The output of an arbitrary k-th (k 1.K) BFL 5 is connected to the information input of the corresponding k-th BVCO 6, the output of which is the k-th output of the device. The inputs of the device of FIG. 2 are the first and second inputs of the control unit 7, respectively, the inputs of the angles Ψ in the azimuthal and q in elevation planes.

 In FIG. 3 is a structural diagram of a database (sampling unit) 3. The block of FIG. 3 contains N (by the number of elements 2 located on the aperture of the lattice 1 in the azimuthal plane) of storage-sampling devices (SEC) 8, an analog N-input adder 9, an analog-to-digital converter (ADC) 10, a buffer register 11, a frequency divider with variable division coefficient (DPKD) 12 and a pulse distributor 13. In this case, the UVX 8, from the first to the Nth, are connected by their inputs to the information inputs of the DB 3, respectively, from the first to the Nth; their outputs to the inputs of the adder 9, respectively, from the first to the Nth. With their control inputs, the UVX 8 is connected to the outputs of the pulse distributor 13, respectively: the first to the Nth, the second to the (N 1) th. nth to (N + 1 n) th. Nth to the first exit. The output of the adder 9 is connected to the input of the ADC 10, the start input of which is connected to the (N + 1) -th output of the pulse distributor 13. The outputs of the data and the readiness of the ADC 10 are connected respectively to the data input of the buffer register 11 and the reset input of the pulse distributor 13. Buffer write input the register 11 is combined with the fourth input of the database 3 control, and its output is the output of the database 3. In this case, the first and second inputs of the database 3 control are combined respectively with the input of the coefficient setting and the clock input of the DPKD 12, the output of which is connected to the clock input of predelitelya pulses 13. The third and fifth control inputs of the database 3 are aligned respectively with the input pulse mode distributor 13 and connected DPKD input 12 and the input synchronization pulses 13 trigger the distributor.

As sampling-storage devices 8, serial chips KR1102SK2 described in [7, p. 445 447] can be used
Analog N-input adder 9 can be implemented on the basis of an operational amplifier in accordance with [8, p. 75 77]
An analog-to-digital converter 10 is known from [9, p. 194,229]
The buffer register 11 can be performed on standard microcircuits described in [10, p. 105 133]
The principles of constructing frequency dividers with an arbitrary division coefficient are described in [11, p. 576 577] in addition to DPKD 12 can be implemented using standard microcircuits known in [12, p. 272,273]
The pulse distributor 13 (as well as the pulse distributor 4 shown in Fig. 2) is known from [13, p. 268] and can be, in particular, implemented on the basis of reverse shift registers, which are described in [10, p. 105 133]
In FIG. 4 is a block diagram of a BFL (beam forming unit) 5. The block of FIG. 4 contains M integrated weighing devices (VHF) 14 (each of which includes a functional converter 15, multipliers 16 and 17) and M-input adders 18 and 19. Moreover, the first inputs of VHF 14 (from the first to Mth) are information inputs of the BFL 5, respectively, from the first to the Mth. The second inputs of all VHF 14 combined with the input control BFL. And the first and second outputs of an arbitrary m-th (m 1.M) VHF 14 are connected to the m-th inputs of adders 18 and 19, respectively, whose outputs form the output of the BFL 5 (i.e., the output for a complex reference). The first VHF input 14 is connected to the combined first inputs of the multipliers 16 and 17, the second VHF input 14 is combined with the input of the functional converter 15, the first and second outputs of which are connected to the second inputs of the multipliers 16, 17, respectively, and the outputs of these multipliers are the first and the second outputs of VHF 14.

 Functional converters 15 can be implemented in accordance with [14, p. 476 477] based on read-only memory with integrated address inputs.

Multipliers 16, 17 and adders 18, 19 can be performed using standard microcircuits described respectively in [12, p. 335 - 344] and [12, p. 344 346]
In FIG. 5 is a structural diagram of a BVCO (block for allocating samples of the complex envelope) 6. The block of FIG. 5 contains a digital switch of samples 20, sign inverters of samples 21 and 22, digital low-pass filters 23, 24, and frequency dividers by 2, 25, 26, and 27. In this case, the first and second inputs of digital switch 20 form the information input of the BVCO 6 (i.e. i.e. input for a complex sample), the first and second outputs of the digital switch 20 are connected to the first inputs of the inverters of the sample samples 21 and 22, respectively, the outputs of which are connected to the inputs of the digital filters 23, 24, respectively. The second inputs of the inverters of the sign 21 and 22 are connected to the outputs of the dividers frequencies at 2 s Responsibly 26 and 27, the inputs of which are combined with the control input of the digital switch 20 and the output of the frequency divider by 2, 25. Its input is the control input of the BVCO 6, the output of which, i.e. complex reference output, form the outputs of digital low-pass filters 23, 24.

Digital switch 20 in accordance with [11, p. 529 530] can be implemented in the form of 2 • Q (where Q is the number of bits of switched samples) of standard 2 x 1 multiplexers with combined address inputs, which are described in [10, p. 150 153]
Each of the inverters of the sign of samples 21, 22 can be made in the form of an inverter of the sign of the discharge of the corresponding sample and implemented on the logic circuit exclusive OR, which is described in [10, p. 56]
Digital low-pass filters 23, 24 are known from [15] and can be implemented in accordance with [14, p. 479,481]
Frequency dividers by 2, 25, 26 and 27 can be performed on the basis of the D-trigger in accordance with [11, p. 546]
In FIG. 6 is a structural diagram of a control unit (control unit) 7. The block of FIG. 6 contains functional converters 28 and 31, comparators of codes 29 and 32, a pulse generator 30, a frequency divider 33, and a frequency divider with a variable division ratio (DPKD) 34. The first input of the control unit 7 is connected to the combined inputs of the functional converter 28 and the code comparator 29 , the second input of the control unit 7 is connected to the combined inputs of the code comparator 32 and the functional converter 31, the output of which is connected to the input of the coefficient setting DPKD 34. The first output of the control unit 7 is the output of the functional converter 28; the second output of the control unit 7 is connected to the combined output of the pulse generator 30 and the clock inputs of the frequency divider 33 and DPKD 34; the third, fourth and fifth outputs of the control unit 7 are respectively the output of the code comparator 29, the output of the DPKD 34, the output of the code comparator 32; and the sixth output of the control unit 7 is connected to the combined output of the frequency divider 33 and the synchronization input DPKD 34.

 Functional converters 28, 31 can be performed according to [14, p. 476 477] on read-only memory devices.

Code comparators 29, 32 can be performed using standard microcircuits described in [10, p. 186 187]
The pulse generator 30 is known from [10, p. 188 196]
The frequency divider 33 can be performed in accordance with [11, p. 576 - 577] and DPKD 34 is implemented similarly to DPKD 12 database blocks 3.

In FIG. 7 (a-e) shows the timing diagrams of the following signals: a sequence (TD) of pulses at the sixth output of the control unit 7 — sampling frequency pulses (7, a); pulses at the fourth output of BU 7 (Tθ) (7, b); pulses at the first (Sθ ₁ ), second (Sθ ₂ ), Mth Sθ _M and (M + 1) -m (Sθ _R ) outputs of the pulse distributor 4, respectively (7, 7, e).

In FIG. Figure 8 (a-h) shows the timing diagrams of the following signals: a sequence (TD) of sampling pulses (8, a); pulses (Sθ _m ) at one of the outputs (arbitrary m-m) of the pulse distributor 4 (8.b); pulses (TΨ _m ) at the output of the DPKD 12 of the m-th database 3 (8, c); pulses at the first (SΨ _m1 ), second (SΨ _m2 ), Nth (SΨ _mN ) and (N + 1) -m (SΨ _mA ) outputs of the pulse distributor 13 of the mth DB 3, respectively (8, g 8, g ); and pulses (SA _m ) at the ADC ready output of the 10th mth database 3 (8, h).

 In FIG. Figure 9 (a) shows the timing diagrams of pulse sequences, respectively: at the control input of all BVCO 6 (9, a), and at the outputs of frequency dividers 25 (9, b), 26 (9, c), 27 (9, d) of any of these blocks.

 In FIG. 10 (a d) respectively: the spectrum S (ω) of the received signal S (t) (10, a); discrete spectrum Y (Ω) of the components of the vector Y (10, b); discrete spectrum E (Ω) of signals at the input of digital filters 23, 24 located in BVKO 6 (10, c); discrete spectrum of B (Ω) signals at the output of the BVCO 6 (10, g).

Consider the operation of the device of FIG. 2 6 as applied to the problem of spatial stabilization of the radiation pattern. Suppose that there is a coordinate system X1Y1Z1 (Fig. 11) and it is necessary to form a radiation pattern in the form of a fan of rays in the Y1OZ1 plane defined by a set of angles θ _k , whose maximum direction in the X1OZ1 plane coincides with the Z1 axis. Suppose that the spatial orientation of antenna array 1 has changed so that the perpendicular to its opening (i.e., the Z axis of the XYZ coordinate system associated with it) has angles -θ, -Ψ with the Z1 axis. Then, for this stabilization, it is necessary to form a beam of the diagram directivity of the antenna array along the angle Ψ in the azimuthal plane XOZ and ensure its compensation along the angle q in the elevation plane YOZ (a priori, we assume that constraint (3) holds for J and θ).

The device of FIG. 2 6 works as follows. The pulse generator 30 unit BU 7 (Fig. 6) generates a clock sequence of TT pulses of high frequency, the period T _t following which satisfies the condition
T _t ≪ d • (C) ^-1 sinδ,
where d is the distance between the elements of the antenna array, C is the wave propagation speed of the received signal S (t), δ is the maximum permissible error in the installation of the rays at the angles J, θ.

поэтому на выходах преобразователей 28 и 31 формируются коды соответственно

. Каждый компаратор кодов осуществляет сравнение своего входного кода D_вх с кодом нуля, и на его выходе могут присутствовать кодовые комбинации, например двухбитовые: A₊, если D_вх > 0; A₀, если D_вх 0; А_-, если D_вх < 0. Для рассматриваемого случая (фиг. 11) Ψ>0 и θ>0,, поэтому на выходах обоих компараторов 29, 32 будут сформированы кодовые комбинации A_ψ= A₊=A_θ= A₊..Codes J and defined by angles θ (e.g. roll and pitch sensors) to the inputs of functional converters 28, 31 and comparators codes 29, 32. Each function generator converts its input code D _Rin in accordance with the function

therefore, codes are generated at the outputs of the transducers 28 and 31, respectively

. Each code comparator compares its input code D _in with a zero code, and code combinations, for example, two-bit ones, can be present at its output: A ₊ , if D _in >0; A ₀ if D _in 0; A _- if D _Ix <0. For the case under consideration (Fig. 11) Ψ> 0 and θ> 0, therefore, code combinations A _ψ = A ₊ = A _θ = A ₊ will be generated at the outputs of both comparators 29, 32. .

где F₀,Δf соответственно центральная частота и ширина спектра принимаемого сигнала S(t) (фиг. 10,а); i любое положительное целое, при котором выполняется неравенство (8). Делитель с переменным коэффициентом деления 34 осуществляет деление частоты его входной последовательности в число раз, определяемое кодом, поступающим на его вход установки коэффициента с выхода функционального преобразователя 31. Поскольку вход синхронизации ДПКД 34 соединен с выходом делителя частоты 33, каждым импульсом TD производится установка ДПКД в исходное состояние. Поэтому его выходной сигнал Tθ (фиг. 7, б) в интервале между любыми импульсами дискретизации представляет собой пачку импульсов, начало которой по времени всегда совпадает с импульсом дискретизации. При этом период следования импульсов в пачке t₁ равен

.The TT sequence is fed to the second output of the control unit 7 and the clock inputs of the frequency divider 33 and the DPKD 34 located in it. The divider 33 has a constant division coefficient K _d such that the pulses of its output sequence are TD (Fig. 7, a; 8, a; 9 , a) follow with a sampling frequency F _d (T _t • K _d ) ^-1 , satisfying condition (4)

where F ₀ , Δf, respectively, the center frequency and spectrum width of the received signal S (t) (Fig. 10, a); i any positive integer for which inequality (8) holds. A divider with a variable division factor 34 divides the frequency of its input sequence by the number of times determined by the code supplied to its input of the coefficient setting from the output of the functional converter 31. Since the synchronization input DPKD 34 is connected to the output of the frequency divider 33, each pulse TD sets the DPKD to the initial state. Therefore, its output signal Tθ (Fig. 7, b) in the interval between any sampling pulses is a packet of pulses, the beginning of which always coincides with the sampling pulse in time. Moreover, the pulse repetition period in the t ₁ packet is

.

на первом выходе; последовательность ТТ импульсов с периодом Т_t на втором выходе; кодовая комбинация A_ψ (результат сравнения кода угла J с нулем) на третьем выходе; следующие с шагом дискретизации Δ = (F_д)^-1 когерентные пачки Tθ импульсов с периодом

их следования в пачке на четвертом выходе; кодовая комбинация A_θ (результат сравнения кода угла q с нулем) на пятом выходе; и последовательность TD импульсов дискретизации с частотой следования F_д (8) на шестом выходе.Thus, during operation of the control unit 7, code is continuously generated at its outputs

at the first exit; a sequence of TT pulses with a period of T _t at the second output; code combination A _ψ (the result of comparing the angle code J with zero) at the third output; following with a sampling step Δ = (F _d ) ^-1 coherent bursts Tθ of pulses with a period

their following in a pack at the fourth exit; code combination A _θ (the result of comparing the angle code q with zero) at the fifth output; and a sequence of TD sampling pulses with a repetition rate of F _d (8) at the sixth output.

Each pulse of the TD sequence starts the pulse distributor (RI) 4. Three modes of its operation are possible, depending on the code combination A _θ supplied to the mode input from the fifth output of the control unit (BU) 7:
if A _θ = A ₊ (i.e., for θ> 0), then the first output of RI 4 is excited synchronously with the TD pulse, and then sequentially in time with an interval equal to the period τ ₁ of the pulses in the Tθ packet arriving at the clock input RI 4 from the fourth output of BU 7, outputs 2, 3. M;
if A _θ = A _- (i.e., when θ <0), then the Mth output of the distributor 4 is excited synchronously with the TD pulse, and then its outputs M-1, M-2.1 with an interval of τ ₁ ;
if A _θ = A ₀ (i.e., when θ = 0), then simultaneously with the TD pulse, the outputs of RI 4 from the first through the Mth are excited simultaneously.

Since in the case under consideration θ> 0, the first output of RI 4 is excited first, and the waveform Sθ _{1 ... m} at the outputs of the distributor has the form, as in FIG. 7 (b-e). Upon excitation of all outputs of RI 4 from the first to the Mth, a reset pulse Sθ _R will be generated at its (M + 1) -th output, which, upon entering the reset input RI 4, will return the latter to its original state and make it insensitive to signals at clock input and mode input until the next sampling pulse arrives.

, т. е. абсолютной величине пространственной задержки в угломестной плоскости антенной решетки 1, определяемой для фронта волны, поступающего на нее с заданного направления, а очередность появления указанных импульсов на выходах РИ 4 определяется знаком угла θ..Thus, in the process of operation of the pulse distributor 4, sequences Sθ ₁ ,, Sθ _{N of} pulses are continuously formed at its outputs, the period of which is equal to the sampling step Δ = (F _d ) ^-1 .. The time shift between the pulses generated at the adjacent outputs is always equal

, i.e., the absolute value of the spatial delay in the elevation plane of the antenna array 1, determined for the front of the wave entering it from a given direction, and the sequence of occurrence of these pulses at the outputs of RI 4 is determined by the sign of the angle θ ..

с выходов РИ 4 поступают на пятые входы управления соответствующих блоков дискретизации (БД) 3. Рассмотрим работу этих блоков на примере произвольного m-го блока (фиг. 3, 8). Импульс Sθ_m с пятого входа управления этого БД 3 поступает на вход синхронизации делителя с переменным коэффициентом деления (ДПКД) 12 и вход запуска РИ 13. ДПКД 12 осуществляет деление частоты тактовой последовательности ТТ, поступающей на его тактовый вход с второго входа управления БД 3, в число раз, определяемое кодом D_ψ, подаваемым на его вход установки коэффициента с первого входа управления данного блока. В результате на выходе ДПКД 12 формируется пачка импульсов TΨ_m (фиг. 8, в), при этом период τ₂ следования импульсов в ней равен

. Так как каждым импульсом Sθ_m по входу синхронизации ДПКД 12 производится его установка в исходное состояние, начало указанной пачки всегда синфазно с импульсом последовательности Sθ_m. Импульсы TΨ_m с выхода ДПКД 12 поступают на тактовый вход РИ 13, который работает аналогично РИ 4 в режиме, задаваемом кодовой комбинацией на его входе режима. Данная комбинация (A_ψ) поступает на указанный вход с третьего входа управления БД 3 и для рассматриваемого случая равна А₊. Поэтому вначале возбуждается первый выход РИ 13 (причем синхронно с импульсом Sθ_m, поскольку последний запускает распределитель 13 по его входу запуска), а затем выходы 2.N. На выходах РИ 13 формируются импульсы SΨ_m1,SΨ_m2,,,SΨ_mN (фиг. 8,г-e), период следования каждого из которых совпадает с шагом дискретизации Δ. Временной же сдвиг между моментами формирования импульсов на соседних выходах указанного распределителя всегда равен

, т. е. абсолютной величине пространственной задержки в азимутальной плоскости антенной решетки 1, определяемой для фронта волны, поступающего на нее с заданного направления (заметим, что знаком Ψ определяется очередность появления импульсов на выходах РИ 13). Импульсы с выходов РИ 13 поступают на входы управления соответствующих УВХ 8 и последовательно во времени со сдвигом t₂ переводят их (начиная с N-го и кончая первым) из режима выборки в режим хранения. Входы УВХ данного БД 3 подключены к выходам тех элементов 2 антенной решетки 1, которые образуют (m m 1)-ю линейную подрешетку, расположенную в азимутальной плоскости (согласно фиг. 2, n-е УВХ данного m-го БД 3 соединено с элементом, имеющим номер (m m 1, n n 1)). Поэтому на выходах УВХ фиксируются аналоговые выборки, полученные со сдвигом τ₂ из выходных сигналов элементов указанной подрешетки. Причем для фронта волны, поступившего синхронно с импульсом Sθ_m на данную подрешетку (т. е. ее (N 1)-й элемент) с направления, заданного углами Ψ и θ,, момент выборки сигнала любого элемента совпадает с моментом его достижения указанным фронтом. Выходные сигналы УВХ 8 суммируются сумматором 9 с весами β_n, в качестве которых выступают либо коэффициенты передачи n-го УВХ, либо коэффициенты передачи сумматора по его n-му входу. Поэтому в момент t_m, когда все УВХ 8 переведены в режим хранения, выходной сигнал y_m сумматора 9 имеет вид

где x_mn(t) выходной сигнал элемента антенной решетки 1, подключенного к n-му входу данного m-го БД 3,
т. е. представляет собой выборку из выходного сигнала подключенной к информационным входам данного БД 3 линейной подрешетки, луч диаграммы направленности которой сформирован в направлении, заданном углом Ψ..Impulses

from the outputs of RI 4 go to the fifth control inputs of the corresponding blocks of discretization (DB) 3. Consider the operation of these blocks on the example of an arbitrary m-th block (Fig. 3, 8). The pulse Sθ _m from the fifth control input of this DB 3 is fed to the synchronization input of a divider with a variable division coefficient (DPKD) 12 and the start input of RI 13. DPKD 12 divides the frequency of the TT clock sequence arriving at its clock input from the second DB control input 3, the number of times determined by the code D _ψ supplied to its input of setting the coefficient from the first control input of this unit. As a result, a burst of pulses TΨ _m is formed at the output of the DPKD 12 (Fig. 8, c), while the period τ _{2 of the} following pulses in it is

. Since each pulse Sθ _m at the synchronization input of the DPKD 12 is set to its initial state, the beginning of this burst is always in phase with the pulse of the sequence Sθ _m . The pulses TΨ _m from the output of the DPKD 12 are fed to the clock input of RI 13, which works similarly to RI 4 in the mode specified by the code combination at its mode input. This combination (A _ψ ) is supplied to the specified input from the third input of the DB 3 control and for the case under consideration is equal to A ₊ . Therefore, at first, the first output of RI 13 is excited (and synchronously with the pulse Sθ _m , since the latter starts the distributor 13 at its start-up input), and then the outputs 2.N. At the outputs of RI 13, pulses SΨ _m1 , SΨ _m2 ,,, SΨ mN are _formed (Fig. 8, g-e), the period of each of which coincides with the sampling step Δ. The time shift between the moments of the formation of pulses at the adjacent outputs of the specified distributor is always equal

i.e., the absolute value of the spatial delay in the azimuthal plane of the antenna array 1, determined for the front of the wave arriving at it from a given direction (note that the sign Ψ determines the order of appearance of pulses at the outputs of RI 13). The pulses from the outputs of RI 13 are fed to the control inputs of the corresponding UVX 8 and sequentially in time with a shift of t ₂ transfer them (from the Nth to the first) from the sampling mode to the storage mode. The inputs of the I – V characteristic of this database 3 are connected to the outputs of those elements 2 of the antenna array 1 that form the (mm 1) -th linear sublattice located in the azimuthal plane (according to Fig. 2, the nth I – V characteristic of this m-th database 3 is connected to the element, having the number (mm 1, nn 1)). Therefore, analogue samples obtained with a shift of τ ₂ from the output signals of the elements of the specified sublattice are recorded at the outputs of the UVC. Moreover, for the front of the wave arriving synchronously with the pulse Sθ _m to this sublattice (i.e., its (N 1) th element) from the direction given by the angles Ψ and θ, the sampling time of the signal of any element coincides with the moment of its achievement by the indicated front . The output signals of the UVX 8 are summed by the adder 9 with the weights β _n , which are either the transmission coefficients of the nth UVX or the transmission coefficients of the adder at its nth input. Therefore, at the time t _m , when all the I / O 8 are transferred to the storage mode, the output signal y _{m of the} adder 9 has the form

where x _mn (t) is the output signal of the element of the antenna array 1 connected to the n-th input of this m-th database 3,
i.e., it is a sample of the linear sublattice connected to the information inputs of a given DB 3 from the output signal, the beam of which radiation pattern is formed in the direction given by the angle Ψ ..

At the time t _m (Fig. 8g), the signal SΨ _{mA of} the ADC start-up 10 is generated at the (N + 1) -m output of RI 13. At the end of the analog-to-digital conversion cycle, the digital equivalent (count) of the output signal is recorded in the output register of the ADC 10 y _m (t _m ) of the adder and the readiness output of the ADC generates a signal SA _m (Fig. 8, h) of readiness. This signal is fed to the reset input of RI 13 and returns to its original state, which is characterized by the fact that all the I / O are in sampling mode, the ADC retains its output code, and RI 13 is in standby mode for the next pulse Sθ _m ..

Since all sampling units 3 work identically, the digital equivalent of voltage of the form (9) is fixed at the output of the ADC 10 of any of them at a time spanning the time interval (N-1) • τ ₂ from the moment of arrival of the pulse Sθ to the fifth control input from the corresponding output of RI 4. As already noted, these pulses are shifted by t ₁ and therefore synchronous with the moments of arrival at the corresponding linear sublattices of the same wave front that entered the antenna array 1 simultaneously with the sampling pulse (at the moment of starting RI 4). Therefore, starting from the moment t _v , which is separated from the moment of starting RI 4 by the time interval (M-1) • τ ₁ + (N-1) • τ ₂ , the set of ADC output codes of all DB 3 is a vector of samples of the output signals of the elements compensated along the angle θ of the equivalent linear sublattice, which is located in the elevation plane, and the elements of this equivalent sublattice are linear sublattices located in the azimuthal plane, whose radiation pattern is formed in the direction of the angle J ..

Each pulse of the TD sequence (sampling pulse) is supplied to the fourth control inputs of all DB 3, and from them to the recording inputs of the buffer registers 11 of these blocks, which gate the output of the ADC 10. By virtue of (3), these pulses arrive at the inputs of the register entries with some delay δt≥0 relative to the moments t _v ; therefore, a voltage readout of the form (9) is fixed in the buffer register of each block. This ensures the formation of a vector [Y] of real samples that follow synchronously with the sampling step Δ .. Moreover, by virtue of (9) and the fact that the triggering times of RI 13 of any mth and (m + 1) th DB 3 are shifted by τ ₁ , for an arbitrary component Y _{m of the} indicated vector, i.e. output signal of the (m + 1) th DB 3, expression (2) takes place.

 Thus, as a result of the operation of the sampling units 3, at the same time, the output signals of the elements 2 of the antenna array 1 are sampled, its compensation in the elevation plane (by the angle θ), and beamforming in the azimuthal plane (with the main maximum in the direction of the angle J). Moreover, the output signal of each DB 3, up to the amplitude, is a sequence of samples of the received signal S (t), and therefore the shape of its discrete spectrum (Fig. 10, b) in the main interval of discrete frequencies coincides with the shape of the spectrum S (t) .

,
где m m 1; α_m коэффициент амплитудного распределения в угломестной плоскости; λ = c•(F₀)^-1 длина волны принимаемого колебания. Поэтому сигналы на выходах указанного УКВ представляют собой произведения дискретного сигнала, поступающего на m-е информационные входы всех БФЛ 5, и Р1_km, P2_km соответственно. Сумматоры 18, 19 осуществляют суммирование сигналов, поступающих соответственно с первых и вторых выходов всех УКВ. При этом их выходные сигналы согласно (4) (6) представляют собой действительную и мнимую части результата

формирования k-го луча в угломестной плоскости.The output signals of the OBD 3 are supplied to the corresponding information inputs of all the BFLs 5, the angle code q is supplied from the second input of the BU 7 to the control inputs. From the mth information input of each BFL connected to the output of the mth DB 3, a discrete signal is fed to the first input of the mth VHF 14 (Fig. 4), and from it to the first inputs of the multipliers 16, 17. The second inputs of these multipliers receive the converted code, respectively, from the first and second outputs of the functional converter 15. Transformation functions P1 _km , P2 _km for the first and second outputs of the mth VHF 14, nakh lasting in the composition of the k-th BFL 5, which is designed to form a beam in the direction of the angle θ _k in the elevation plane (Fig. 11) have the form

,
where mm 1; α _{m is the} coefficient of amplitude distribution in the elevation plane; λ = c • (F ₀ ) ^-1 the wavelength of the received oscillation. Therefore, the signals at the outputs of the specified VHF are the products of the discrete signal supplied to the mth information inputs of all BFL 5, and P1 _km , P2 _km, respectively. Adders 18, 19 summarize the signals coming from the first and second outputs of all VHF, respectively. Moreover, their output signals according to (4) (6) represent the real and imaginary parts of the result

the formation of the k-th ray in the elevation plane.

, на которую согласно (7) следует умножить входной сигнал

произвольного k-го БВКО, имеет вид

, т. е. ее действительная и мнимая части есть последовательности нулей и ± единиц. При этом действительная EC_k(lΔ) и мнимая ES_k(lΔ) части результата умножения связаны с действительной RC_k(lΔ) и мнимой RS_k(lΔ) частями

следующим образом (см. таблицу).Complex discrete signals from the outputs of the BFL 5 are fed to the information inputs of the corresponding BVCO 6, the operation of which is based on the allocation of the samples of the complex envelope by means of synchronous detection (7). Since the sampling rate satisfies (8), the reference sequence

which, according to (7), should multiply the input signal

arbitrary k-th BVKO, has the form

, i.e., its real and imaginary parts are sequences of zeros and ± ones. The real EC _k (lΔ) and imaginary ES _k (lΔ) parts of the multiplication result are related to the real RC _k (lΔ) and imaginary RS _k (lΔ) parts

as follows (see table).

 Therefore, multiplication by the reference sequence is reduced to switching samples and inverting their signs.

комплексных отсчетов, которая согласно приведенной выше таблице есть произведение входного сигнала БВКО и опорной последовательности. Действительная и мнимая части

поступают на входы цифровых фильтров 23 и 24, которые подавляют высокочастотные компоненты в спектре

(фиг. 10, в, г) и на выходе БВКО, образованном выходами указанных фильтров, формируется последовательность

отсчетов комплексной огибающей сигнала, принятого по k-му лучу диаграммы направленности.The samples of the real and imaginary parts of the input signal of an arbitrary k-th BVCO 6 come from its information input, respectively, to the first and second inputs of the digital switch 20 (Fig. 5). The control input of the switch from the output of the frequency divider 25 receives a sequence of pulses, the repetition rate of which is half the sampling frequency (Fig. 9, a). With a high signal level at the control input of the switch, the first and second outputs receive samples, respectively, from the first and second inputs. With a low signal level at the control input, the first output of the switch counts from the second input, and the second output from the first input. The samples from the first and second outputs of the switch 20 are received at the first inputs of the sign inverters 21 and 22, respectively. The second inputs of these inverters receive signals from the outputs of the frequency dividers by 2, 26 and 27, respectively. With a high level of signals at its second input (Fig. 9, c, d), the inverters change to the opposite signs of the samples received at their first inputs, t. e. samples with numbers l 1, 2, 5, 6. for inverter 21 and l 2, 3, 6, 7. for inverter 22. As a result, a sequence is formed at the outputs of inverters

complex samples, which according to the table above is the product of the input signal of the BVCO and the reference sequence. Real and imaginary parts

arrive at the inputs of digital filters 23 and 24, which suppress high-frequency components in the spectrum

(Fig. 10, c, d) and at the output of the BVCO formed by the outputs of these filters, a sequence is formed

samples of the complex envelope of the signal received by the k-th beam pattern.

 The proposed method of forming a radiation pattern is carried out in the sequential implementation of the following operations.

 1. Measure the wave field, ie signal reception by means of a flat antenna array.

 2. Specify some angles Ψ and θ, respectively, in the azimuthal and elevation planes of the antenna.

 3. Discretize the output signals of the elements of the antenna array and simultaneously with the sampling, the beam is formed by the radiation pattern along the angle компенса and the compensation of the antenna array by the angle q. For this, for example, samples of the output signals of adjacent elements of the antenna array are obtained with a time shift, the value of which is selected based on the value of the angle J for elements having the same coordinates in the elevation plane, and based on the value of the angle q for elements having the same coordinates in the azimuthal plane, then form a vector Y of real discrete signals, each component of which is obtained by weighting the sum of the samples obtained from the output signals of those elements of the antenna array, which have the same coordinates in the elevation plane.

To carry out this operation, in particular, in the device of FIG. 2 6, within each sampling step, samples are shifted by sequentially starting sampling blocks 3 from the pulse distributor 4 with an interval t ₁ ,, equal to the spatial delay in the elevation plane, and sequentially transferring to storage mode the sampling-storage devices 8 of each block 3 with an interval τ ₂ ,, equal to the spatial delay in the azimuthal plane;
obtaining the components of the vector Y by summing, with a multi-input analog adder, 9 output signals of all sample-storage devices of each sampling block;
the formation of the vector Y by simultaneously gating the output of the ADC 10 of each block discretization buffer registers 11.

 4. A set of rays is formed in the elevation plane, for example, by summing the components of the vector Y obtained as a result of the previous operation 3 with complex weight coefficients. To carry out this operation, in particular, in the device of FIG. 2 6 in each beam forming unit 5, the output samples of all sampling units are multiplied by the real (multiplier 16) and imaginary (multiplier 17) parts of the corresponding weight coefficient generated by the functional converters 15, and the real and imaginary parts of the multiplication results are summed by adders 18 and 19.

 5. Separate the samples of the complex envelope of the signals received for each of the set of rays. To carry out this operation, for example, in the device of FIG. 2 through 6 block synchronously detect the output signals of all beam forming units 5.

 This embodiment of the method of forming the radiation pattern allows in comparison with the prototype [3, p. 110] to increase the accuracy of chart formation and at the same time reduce the computational cost when implementing the method.

In the proposed method, in contrast to the prototype, the specified accuracy does not depend on the sampling step. In particular, when implementing the method using the device of FIG. 2 6 it is δ [rad] = arcsin [2 • (d • F _t ) ^-1 ] and for an equidistant lattice with a half-wave distance d between the elements is
δ = arcsin (2 • F ₀ • F $\genfrac{}{}{0ex}{}{\text{-one}}{\text{t}}$ ) (10)
where F _{o is the} central frequency of the received signal, F _t is the frequency of the clock generator 30 (Fig. 6).

When implementing the same prototype method, the specified accuracy is
δ = arcsin (2 • F ₀ • F $\genfrac{}{}{0ex}{}{\text{-one}}{\text{d}}$ ) (eleven)
where F _d is the sampling rate.

 Let us estimate the computational costs when implementing the proposed method and prototype in terms of the volume of computational operations, namely, the number of multiplications VM and additions VS per unit time, which is necessary for the formation of one beam of the radiation pattern of an antenna array of size N x N elements.

For the prototype VM ₁ = N • (N + 1) • F _d ; VS ₁ = (N + 1) • (N-1) • F _d . For the proposed method, when its operation 3 is performed by shifting the samples and generating an N-dimensional vector Y, operation 4 by summing the components of Y with complex weights, and operation 5 is implemented by synchronously detecting VM ₂ [2 (N + H) + 4] • F _d upon failure of condition (8) and VM ₂ 2 (N + H) • F _d upon its fulfillment; VS ₂ [2 (N + H) 4] • F _d (here H ≃ F _d (Δf) ^{-1 is the} duration of the impulse response of the digital filter, Δf is the width of the spectrum of the received signal).

Поскольку принимаемый сигнал всегда узкополосен F₀≫ Δf, при выбираемой исходя из ширины спектра частоте дискретизации обеспечиваемая прототипом точность установки лучей (11) недопустимо низка, тогда как в предложенном способе указанная точность настолько велика, насколько большой может быть выбрана тактовая частота F_t.According to the proposed method, the sampling frequency can be selected based on the width of the spectrum of the signal, namely F _d ≥4 • Δf, while the beams of the radiation pattern are sufficiently narrow, the aperture of the antenna array N • d should significantly exceed the wavelength. We take N> 10, HN, then even at the same sampling rate as the prototype, the proposed method provides a gain Q _v in the amount of calculations equal to

Since the received signal is always narrow-band F ₀ ≫ Δf, at the sampling frequency selected on the basis of the spectrum width, the beam alignment accuracy provided by the prototype (11) is unacceptably low, while in the proposed method the specified accuracy is as high as the clock frequency F _t can be chosen.

 The analysis shows the novelty and inventive step of the solution, and the technical study (commissioned by the GUNIO of the RF Ministry of Defense “Shagan-RVO”) confirms the possibility of its industrial applicability.

Literature Sources of information
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 4. Leshchinsky M.M. and Torgushin E.I. About one method of sampling narrowband signals. On Sat Methods and devices for primary signal processing in radio systems. Gorky: ed. GPI, 1985, pp. 7-12.

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Claims (2)

 1. The method of forming a radiation pattern, which consists in receiving signals through a flat antenna array, discretizing the output signals of the antenna array elements and two-stage beamforming separately in the azimuthal and elevation planes, characterized in that the angles Ψ and θ are set respectively in azimuthal and elevation planes, at the same time as sampling, they form a beam of the radiation pattern along the angle Ψ and compensate the antenna array along the angle q, for which sampling the output signals of adjacent elements of the antenna array is obtained with a time shift, the value of which is selected based on the angle Ψ for elements having the same coordinates in the elevation plane, and based on the angle q for elements having the same coordinates in the azimuthal plane, form a vector Y real discrete signals, each component of which is obtained by weighting the samples obtained from the output signals of those elements of the antenna array that have the same coordinates in the elevation plane, then produce a set of rays in the elevation plane, and then carry out the selection of the samples of the complex envelope of the signals received for each of the set of rays.  2. The method according to claim 1, characterized in that the formation of each of the set of rays in the elevation plane is carried out by summing the components of the vector Y with complex weights.

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