RU2071066C1 - Method for spectral analysis of signals - Google Patents

Abstract

FIELD: radio engineering. SUBSTANCE: device for realization of method comprises input amplifier 1, multitap delay line 2, amplifiers 3, 4, 5, analog adder 6, band-pass filter 7, mixer 8, linear frequency modulation heretodyne 9, dispersion delay line 10, compensating amplifier 11. EFFECT: higher efficiency. 4 dwg

Description

 The invention relates to radio engineering, can be used in devices for the detection of quasi-monochromatic radio signals and in spectrum analyzers.

 The aim of the invention is to increase noise immunity.

 In FIG. 1 shows a block diagram of a device that implements the proposed method; figure 2 diagram of the conversion of signal volumes after the commission of certain operations of the method; figure 3 time diagrams of the processes of expansion of the radio signal in duration and its band-pass filtering; figure 4 - diagrams of the formation of broadband, in particular, the LFM equivalent and compression of the latter in the spectral-time filter (DLZ).

The device (Fig. 1) includes series-connected input amplifier 1, delay line 2, for example, with N taps, channel amplifiers 4, 5, compensating signal loss in line 2, multi-input analog adder 6, band-pass filter 7, connected to each of the taps of the line , a mixer 8 with a LFM local oscillator 9, the synchronization of which (start) is carried out by an external clock pulse, a dispersion delay line (DLZ) 10 and its compensating amplifier 11, which compensates for signal loss in the DLZ. We will assume that the hollow transmission coefficient of the circuit from the input of the multi-tap delay line 2 to the output of the bandpass filter 7 is equal to unity in the passband of the bandpass filter 7 and rapidly decreases outside this band. We will also assume that the total transmission coefficient of the broadband path circuit from the input of the mixer 8 to the output of the compensating amplifier 11 is also equal to unity in the operating frequency band ΔF _lz . The transmission coefficient of the input amplifier 1 is assumed to be K> 1 in the band of the input signal spectrum Δf _Rin ≥ Δf _c ..

 The method is as follows.

The input quasi-monochromatic signal, for example, a rectangular radio pulse of duration τ _c with a carrier frequency f _c , shown in Fig. 2 as a rectangle 12, is amplified in the mixture with noise in the input amplifier 1, after which it is extended several times in duration in a comb filter from multi-tap delay line 2, channel amplifiers 3, 4 and a multi-input analog adder 6, as a result of which a packet of N mutually coherent radio pulses is formed, which follow inside a packet with a low duty cycle g. The coherence factor of the carrier oscillations of each of these radio pulses of the packet is provided from the condition:
f _c • Δτ = M is an integer, (1)
where Δτ is the delay between adjacent tap numbers.

Let the input radio signal (figure 2) is a radio pulse of duration t _c with a carrier frequency f _c . Its image on the time-frequency diagram is represented by a hatched rectangle 12 of unit volume (base), since the product of the signal duration by the width of its spectrum Δf _c = 1 / τ _c is equal to unity. As a result of an N-fold extension of the signal duration with preservation of its coherence and bandpass filtering of a packet of N mutually coherent radio pulses following with a low duty cycle g≥1, a radio pulse of a single volume (base) with duration τ _and = qτ _c N and spectrum Δf _and = 1 / Nqτ _c . The frequency of the carrier oscillations in the extended pulse width may remain the same or somehow change to a known difference (the latter is shown in Fig. 2). The duration τ _and setting a lesser or commensurate with the duration τ _LZ impulse response of a matched filter used in the spectro-temporal filter compression (DLA), which defines a particular value of the expansion coefficient of length N. Then, the received signal is formed corresponding to the broadband, such as chirped, equivalent, the working duration of which is equal to the duration of τ _{and the} generated radio pulse (in Fig. 2 it is shown by a bold oblique line inside the rectangle characterizing the base of the DLZ). This equivalent is compressed in duration in the corresponding DLZ 10, forming a short correlation maximum at its output. An image of a complex broadband signal (e.g., LFM) equivalent in FIG. 2 is shown by a rectangle 13 inside a rectangle 14 depicting the volume (base) of a spectro-temporal compression filter (DLZ). The increase in the size of the rectangle 14 compared with the rectangle 13, which is displayed by the inequality τ _lz ≥ τ, _and usually takes place due to the uncertainty of the moment when the received radio pulse appears at the input of the matched filter (the position of the rectangle 12 on the time axis, figure 2). If the moment of arrival of the signal is strictly known, then it is advisable to use the mode of full coordination of times τ _and = τ _lz . Uncertainty area in time appears on the input of the matched filter signal Δτ _Σ = τ _LZ -τ _and determines the reliability margin proper operation of the matched filter that is implemented by using the latter in a location, because in this case the value of the time zone of uncertainty associated with an uncertainty of the position of located object of range according to the equality ΔT _Σ = 2 (D _max -D _min ) / C, where D _min and D _max are the minimum and maximum boundaries of the ranges measured by the locator, respectively. When applied to a fixed communication system such timing uncertainty can not exist, and in this case the duration of the extended pulse _and τ is selected optimally high, equal to the duration τ of the impulse response DLA _LZ.

Полосовой фильтр 7 представляет собой инерционную систему, которая устранит "пустые" промежутки между смежными радиоимпульсами в пачке, поэтому на выходе полосового фильтра будет выделяться радиоимпульс длительностью τ_и со структурой моноимпульса (как указано на фиг.3), но с несколько уменьшенной средней (усредненной) амплитудой по сравнению с амплитудой радиоимпульсов в пачке при g>1, а именно с амплитудой

где А_c амплитуда сигнала на выходе входного усилителя 1 (при указанном выше условии единичности коэффициента передачи тракта от входа многоотводной линии задержки 2 до выхода полосового фильтра 7), величина которой связана с входной мощностью принимаемого сигнала P_вх и входным сопротивлением входного усилителя 1 R_вх и его усилением К соотношением

Полагая шум на входе системы "белым" (равномерным гауссовским), находим, что дисперсия шума σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{ш}}$ на выходе полосового фильтра 7 будет равна
σ²=k²NGR_вх/2τ_и, (5)
где G приведенная к нагрузке 1 Ом спектральная плотность мощности шума на входе системы (Вт/Гц).To transmit an extended-duration radio pulse (packs of N mutually coherent radio pulses) without losing its energy, you can have a band system (band-pass filter 7) with a passband comparable to the reciprocal of the specified duration τ _and . In other words, the bandpass filter 7 is selected with a passband, for example, equal to

The band-pass filter 7 is an inertial system that eliminates the “empty” gaps between adjacent radio pulses in the packet, therefore, at the output of the band-pass filter, a radio pulse of duration τ _and with a mono-pulse structure (as indicated in FIG. 3), but with a slightly reduced average (average ) amplitude compared with the amplitude of the radio pulses in the packet for g> 1, namely with the amplitude

where A _{c is the} amplitude of the signal at the output of the input amplifier 1 (under the condition of unity of the transmission coefficient of the path from the input of the multi-tap delay line 2 to the output of the bandpass filter 7), the value of which is related to the input power of the received signal P _in and the input resistance of the input amplifier 1 R _in and its amplification K by the ratio

Assuming the noise at the input of the system to be “white” (uniform Gaussian), we find that the noise variance σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{w}}$ at the output of the bandpass filter 7 will be equal to
σ ² = k ² NGR _in / 2τ _and , (5)
where G is the spectral density of the noise power at the input of the system (W / Hz) reduced to a load of 1 Ohm.

Таким образом, расширенный и отфильтрованный радиоимпульс, изображаемый на фиг. 2 прямоугольником 15, является аналогом-представителем принятого радиоимпульса, изображенного прямоугольником 12 на фиг.2, но в отличие от последнего, имеет значительно более узкий спектр, допускающий применение к нему узкополосной фильтрации, основной смысл введения которой заключается в ограничении шума узкой полосой Δf_ф, что позволяет существенно уменьшить дисперсию шума σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{ш}}$ на входе смесителя 8, которым начинается широкополосный тракт согласованного фильтра заявляемого типа.Therefore, at the output of the bandpass filter 7, the signal-to-noise ratio will be equal to

Thus, the expanded and filtered radio pulse depicted in FIG. 2 by rectangle 15, is an analogue representative of the received radio pulse, shown by rectangle 12 in figure 2, but unlike the latter, it has a much narrower spectrum that allows the use of narrow-band filtering, the main purpose of which is to limit the noise to a narrow band Δf _f , which can significantly reduce the noise variance σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{w}}$ at the input of the mixer 8, which begins the broadband path of the matched filter of the claimed type.

где ψ=τ_лз/τ₄=1 и B=ΔF_лзτ_лз база применяемой ДЛЗ 10, то величиной вклада (σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{ш}}$ )² в общий шум на выходе системы в выражении (8) можно пренебречь и считать, что дисперсия шума на выходе системы не изменилась и равна величине, определяемой из выражения (5).With such a limitation of the noise band by the band-pass filter 7 and in the presence of an amplification K in the input amplifier, choosing the value of K can negate the intrinsic noise of a broadband path having a noise power spectral density G ^* and a working band ΔF _lz determined by the passband of the DLZ 10. So, with the additive addition of noise from the output of the band-pass filter 7 and the broadband path itself (σ $\genfrac{}{}{0ex}{}{\text{*}}{\text{w}}$ ) ² = G ^* R $\genfrac{}{}{0ex}{}{\text{*}}{\text{in}}$ ΔF _lz / 2 the resulting noise at the system output (for a single transmission coefficient in the circuit from the input of the mixer 8 to the output of the compensating amplifier 11 in the passband of the DLZ) will be equal to:
σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{wΣ}}$ = σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{w}}$ + (σ $\genfrac{}{}{0ex}{}{\text{*}}{\text{w}}$ ) ² (7)
If a condition of the form

where ψ = τ _lz / τ ₄ = 1 and B = ΔF _lz τ _{lz the} base of the applied DLP 10, then the contribution value (σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{w}}$ ) ² into the total noise at the system output in expression (8) can be neglected and it can be assumed that the noise variance at the system output has not changed and is equal to the value determined from expression (5).

We note that expression (6), taking into account expressions (5) and (2), proves that there is an important property of the invariance of the signal-to-noise ratio at the output of the band-pass filter 7, with respect to the change in the expansion factor N of the duration of the received radio signal. The multiplier N in the numerator of expression (5) reflects the additivity property of the addition of noise mutually uncorrelated components in the multi-input analog adder 6. The non-correlation of these noise components is justified by the fact that the partial delay Δτ in the multi-tap delay line 2 is chosen greater than the correlation interval for the noise, which is determined in its turn the passband of the input amplifier 1, line 2 and channel amplifiers 3, 4, 5. Broadband of these circuit elements can significantly reduce the interval yatsii noise, i.e. to reduce the possible delay value Dt or, equivalently, to reduce the input signal duration t _c ≅ Δτ. Since for uncorrelated noises the variance of the sum of the noise components arising at the outputs of channel amplifiers 3, 4, 5 is equal to the sum of the variances of these components, and the variances of these components are assumed to be the same, it is clear why the factor N is used in expression (5), but since the bandpass filter 7 has a passband inversely proportional to the number N, then the invariance property of the signal-to-noise ratio μ _f = inv / N is justified.

 Assuming mixer 8 to be a linear frequency-converting element, while also assuming that its transmission coefficient is unity for the component of the converted frequency of the LFM equivalent, the amplitude of the LFM equivalent and the rms noise voltage at the output of the mixer 8 will be the same as follows from expressions (4) and (5) subject to the fulfillment of condition (8). That is, the signal-to-noise ratio at the input of the DLZ 10 will be determined by expression (6).

The formation using a mixer 8 of a broadband, for example, LFM, equivalent is achieved by heterodyning the signal coming from the output of the bandpass filter 7 with the LFM-local oscillator 9 signal, which generates a pulse of duration τ _lz and with a frequency deviation in it ΔF _lz from the moment of the arrival of the external LFM sync pulse linear law as a function of time df / dt = ΔF _lz / τ _lz . Thus, the volume (base) of the signal generated by the LFM local oscillator 9 is depicted by box 14 in FIG. 2 and actually represents the base of the DLZ 10 equal to B = ΔF _lz τ _lz . Since the moment of arrival of the input clock chirped local oscillator 9 is not exactly equal to the time of exposure to the input of the matched filter of the received signal, as can be seen from Figure 2, these moments are separated by time domain a timing uncertainty ΔT _Σ = τ _LZ (1-1 / ψ), then the position of the bold line in FIG. 2, which is simply connected to rectangle 13, characterizes the time-frequency arrangement of the LFM equivalent at the output of mixer 8 inside rectangle 14 in FIG. 2. If the time of arrival of the signal at the input of the matched filter changes, then the bold line of the LFM equivalent experiences a shift along the oblique straight line (the beam coinciding with the bold line), moving the rectangle 13 inside the rectangle 14 along the indicated oblique line, which adequately reflects the time-frequency tuning of the LFM -heterodyne 9 under the action of a sync pulse.

The operation of expanding the duration of the received radio signal is conventionally shown in Fig.3. Plots 3a, 3b and 3c express the voltage at the output of the channel amplifiers 3, 4, 5, the shaded areas being radio pulses, and the unshaded components of the noise voltage (conditionally). In these diagrams, noise is not shown in the time intervals of the action of radio pulses (the duration of which is τ _c and the partial delay in the line is 2 Δτ> τ _c ). On diagram 3g, the type of voltage at the output of the multi-input analog adder 6 is given. It can be seen that the amplitude of the pulses and the rms noise level at the output of the adder 6 are preserved. This is true because the comb filter based on the connection of the multi-tap delay line 2, channel amplifiers 3, 4, 5 and the multi-input analog adder 6 is a narrow-band system, the passband of which is determined by the reciprocal of the total delay τ _and in line 2. In this case, the band-pass filter 7 is an inertial-smoothing link that allows you to eliminate (reduce the effect) temporary gaps between adjacent radio pulses inside the formed packet. This is shown in the 3D plot. It is seen that in the process of such smoothing, the amplitude of the output signal from the band-pass filter 7 decreases slightly, and the rms noise voltage remains the same.

In Fig. 4, the process of generating LFM equivalent in the mixer 8 and its compression in DLZ 10 (in Fig. 4a), the result of which is indicated in Fig. 4b, is explained. The time-compressed DLZ correlation response occurs when the spatial distribution of the antinodes of the ultrasonic wave excited in the DLZ 10 sound duct from the LFM equivalent coincides with the same distribution of the interdigital transducers deposited on the piezoceramic DLZ substrate and representing a non-spatial spatial structure. To increase the zone of temporary uncertainty ΔT _Σ, one should use DLZ with an increased value of t _{lz the} duration of their impulse characteristics.

,
где -ΔT_Σ/2≅ Δt≅ ΔT_Σ/2, f_o центральная частота ДЛЗ 10.It should be noted that for Φ> 1, the compressed radio pulse at the output of the DLZ 10 has a carrier frequency, which is a function of the time shift Δt of the moment of arrival of the input pulse to the filter input of duration τ _c and frequency f _c . This frequency is equal to

,
where -ΔT _Σ / 2≅ Δt≅ ΔT _Σ / 2, f _{o the} central frequency of the DLS 10.

The outputs of the N DLA of the analyzer are connected to a multi-input analog type 6 adder, after which the resulting packet of radio pulses is filtered in a band-pass filter of type 7, and then an LFM equivalent is formed from it, which is compressed in a DLZ of type 10. In this case, the partial delay in the multi-tap delay line of the analyzer is subordinated condition (1) for the frequency f _{about an} - the central frequency of the DLZ analyzer. In this case, in the input path, such signal losses do not occur that are characteristic of losses in the multi-tap line 2 of the circuit of Fig. 1 and are compensated by channel amplifiers 3, 4, 5 with a sufficiently large gain and low level of intrinsic noise.

Claims (1)

 A method of spectral analysis of signals, including signal amplification, the formation of a broadband linear frequency-modulated equivalent, and spectral-time compression of the latter, characterized in that, in order to increase the noise immunity, the amplified signal is expanded in duration with the preservation of coherence to a value of commensurate with the duration of the impulse response of the spectral-time compression filter, and filtered with a passband commensurate with the inverse coherently extended radio signal duration.

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