RU2067787C1 - Double-stage image-suppression frequency changer - Google Patents

Abstract

FIELD: radio engineering; superheterodyne receivers oif various applications used to afford image rejection. SUBSTANCE: frequency changer has band filter 1, frequency changer proper 2, intermediate-frequency filters 3,9 ± 90 deg. phase shifter 4, adder 5, heterodyne 6, 90 deg. phase shifter 7, mixer 8, and wave trap 10. EFFECT: improved signal-to-noise ratio at output. 2 cl, 1 dwg

Description

 The proposed device relates to radio engineering and can be used in radio receivers of the superheterodyne type for various purposes to attenuate interference received at the frequency of the mirror channel (mirror noise).

 Known devices for suppressing interference of the mirror channel (see US patent N 3 070 747, CL 325-437, 12.25.62; Japan patent N 58-26700, CL H 04 B 1/26, H 04 B 1/10), in which an additional (parallel) receiving channel is provided, comprising an additional mixer. In these devices, due to the branching of the input signal, using phase shifters, it is possible to ensure the antiphase of the mirror noise and the in phase of the useful signal at the outputs of the main and additional receiving channels. As a result, provided that the primary and secondary channels are identical, compensation for specular interference is achieved. The disadvantage of these devices is the deterioration of the noise properties of the Converter due to the influence of the noise of the additional channel.

Of the known devices, the closest in technical essence is the device according to US patent N 2 964 622, class. 325-437, 13.12.60, containing mixers, a local oscillator, filters, phase shifters 90 ^o , as well as the adder. The useful signal and mirror noise at the output of the band-pass filter branch out and enter the inputs of the first and second mixer. The output of one mixer through the first PLL is connected to one input of the adder. The output of another mixer through a second intermediate-frequency filter (PLL) and a first phase shifter connected in series with another input of the adder. The local oscillator output is directly connected to the first mixer, and to the second mixer through the second phase shifter.

 Components of mirror noise at the inputs of the adder are out of phase, thereby fulfilling the phase condition for compensation. If the transmission coefficients of the primary and secondary reception channels from the input of the device to the inputs of the adder are the same, then the levels of mirror noise at the inputs of the adder are also the same, i.e. the amplitude compensation condition is satisfied. As a result of the simultaneous fulfillment of both conditions, the mirror noise is suppressed (compensated).

 However, in the known two-channel device, noise properties are degraded in comparison with the classic single-channel frequency converter, since the same useful output signal power is achieved with twice as much output noise power. As a result, the noise figure of the known device is two times worse (more) than in a single-channel, respectively, the signal-to-noise ratio at the output deteriorates (decreases) twice.

 The invention solves the problem of reducing the noise figure and increasing the signal-to-noise ratio at the output.

This result is achieved due to the fact that in the known device containing the input filter, the first and second mixers, the local oscillator, the first and second phase-phase converters, the first and second phase shifters at 90 ^o and the adder, an additional filter plug at the intermediate frequency is introduced and some relations between elements. As a result, a new device emerged: a two-stage frequency converter with mirror suppression, containing a series-connected bandpass filter, a frequency converter, a first intermediate frequency filter, a ± 90 ^o phase shifter and an adder, a local oscillator, a 90 ^o phase shifter, a mixer and a second intermediate frequency filter whose output is connected to another input of the adder. In this case, the other (heterodyne) input of the frequency converter is connected to the output of the local oscillator.

 The claimed device differs from the prototype in that a filter plug is inserted between the output of the frequency converter and the other (signal) input of the mixer, and the frequency converter is made in the form of a balanced frequency converter with a gain greater than unity. The conversion coefficient of the mixer is equal to the ratio of conversion and gain coefficients of the balanced frequency converter with a gain greater than unity.

 The elimination of signal branching at the input of the frequency converter allows, with a sufficiently large value of the gain of the latter, to reduce the noise figure of the device as a whole. This effect is achieved in cascade, i.e. successive switching on of the mixer with respect to the frequency converter, and, as you know, in a multistage circuit, the noise intensity is determined mainly by the noise of the first stage. In the known device, the first and second mixers are connected in parallel, therefore their noise equally affects the resulting noise figure. In addition, in the inventive device, the frequency converter not only converts the frequency of the input signal, but also amplifies the input signal at the same frequency, which is equivalent to increasing the transmission coefficient of the device as a whole. As a result of the combined action of both factors, the noise figure of the claimed device is less than that of the known device. The signal-to-noise ratio at the output is correspondingly higher.

,
где K_2У коэффициент усиления преобразователя. Следовательно, заявленное устройство имеет преимущество перед известным устройством по коэффициенту шума при K_2У > 1/2, когда G > 1. Например, при K_2У 1 G ≈ 1,3, при K_2У 5 G ≈ 1,8.The gain in the value of the noise figure compared with the known device is

,
where K _2U converter gain. Therefore, the claimed device has an advantage over the known device in terms of noise figure at K _2У > 1/2, when G> 1. For example, at K _2У 1 G ≈ 1.3, at K _2У 5 G ≈ 1.8.

 With an increase in the gain of the first mixer, the gain increases, but no more than twice. Accordingly, the signal-to-noise ratio at the output increases, but no more than four times.

 The drawing shows a functional electrical diagram of a two-stage frequency converter with suppression of the mirror channel.

A two-stage frequency converter with suppression of the mirror channel contains a band-pass filter 1, a frequency converter 2, a first intermediate frequency filter (FPF) 3, a phase shifter ± 90 ^o 4, an adder 5, a local oscillator 6, a phase shifter 90 ^o 7, a mixer 8, a second phase-shifting filter 9 filter plug 10.

,
где K_2п и K_2у коэффициенты преобразования и усиления преобразователя частоты 2 соответственно, т.е. коэффициент преобразования смесителя 8 должен быть равен отношению коэффициентов преобразования и усиления преобразователя частоты 2. Второй ФПЧ 9 выделяет составляющие с промежуточной частотой, по характеристикам идентичен первому ФПЧ 3. Фильтр-пробка 10 не пропускает напряжение с промежуточной частотой на вход смесителя 8 (наличие этого напряжения, не прошедшего через фазовращатель на ± 90^o 4, на входе сумматора 5 может нарушить фазовое условие компенсации.The band-pass filter 1 is a standard element of the radio receiver and, in a particular case, may be absent. The frequency converter 2 is performed by the broadband output so that the components not only at the intermediate frequency, but also at the frequencies of the input useful signal and specular interference are allocated at its output. To obtain maximum gain, the gain must be greater than unity, which can be realized only on transistors. In addition, the frequency Converter 2 is balanced to prevent the appearance of voltage at its output with a local oscillator frequency. (The need to eliminate the local oscillator voltage is dictated by the fact that a voltage of a larger magnitude, which has not passed through the phase shifter by 90 ^o 7, can change the phase condition for compensation). The first HPF 3 passes the voltage at an intermediate frequency. Phase shifter ± 90 ^o 4 provides a phase shift of the components at an intermediate frequency of 90 ^o with the upper tuning of the local oscillator and -90 ^o with the lower. The adder 5 summarizes the components at an intermediate frequency formed in two different circuits, the output of the adder is the output of the device. The local oscillator 6 is a standard element of the receiver and, as in the known device, is a common element of two converter circuits. Phase shifter +90 ^o 7 rotates the phase by 90 ^o . The mixer 8 converts the useful signal and specular interference amplified in the frequency converter 2 to an intermediate frequency. In this case, both the main and the mirror channels of mixer 8 are useful. The conversion coefficient of the mixer 8 K _8p should be connected in a certain way with the parameters of the frequency converter 2 in order to ensure equal levels of the components of the mirror noise at the inputs of the adder. Namely:

,
where K _2p and K _{2y are} the conversion and gain coefficients of the frequency converter 2, respectively, i.e. the conversion coefficient of the mixer 8 should be equal to the ratio of the conversion and amplification factors of the frequency converter 2. The second phase converter 9 selects components with an intermediate frequency, is identical in characteristics to the first phase converter 3. The filter plug 10 does not pass voltage with an intermediate frequency to the input of the mixer 8 (the presence of this voltage , not passed through the phase shifter ± 90 ^o 4 at the input of the adder 5 may violate the phase condition for compensation.

 The device operates as follows.

образуется составляющая полезного сигнала на промежуточной частоте ω $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{пч}}$ = ω_г-ω_c с фазой

и составляющая зеркальной помехи на промежуточной частоте ω $\genfrac{}{}{0ex}{}{\text{(з)}}{\text{пч}}$ = ω_з-ω_г с фазой

. Эти составляющие через второй ФПЧ 9 поступают на один вход сумматора 5. Колебания на промежуточной частоте с преобразователя частоты 2 через первый ФПЧ 3 поступают на вход фазовращателя на ± 90^o 4, где получают дополнительный сдвиг по фазе на 90^o. В результате на другом входе сумматора 5 присутствует составляющая полезного сигнала на частоте ω $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{пч}}$ = ω_г-ω_c с фазой

и составляющая зеркальной помехи на частоте ω $\genfrac{}{}{0ex}{}{\text{(з)}}{\text{пч}}$ = ω_з-ω_г с фазой

. Следовательно, на входах сумматора 5 составляющие полезного сигнала синфазны, а составляющие зеркальной помехи противофазны, т.е. выполняется фазовое условие компенсации зеркальной помехи. (Проделанные рассуждения справедливы для верхней настройки гетеродина, когда ω_г>ω_e. При нижней настройки гетеродина, когда ω_г<ω_e, все рассуждения остаются справедливыми, если сдвиг фазы в фазовращателе на ± 90^o 4 составляет -90^o). Для компенсации зеркальной помехи необходимо, чтобы уровни составляющих зеркальной помехи на входах сумматора 5 были одинаковы, т.е. необходимо выполнить амплитудное условие компенсации. Сформулируем это условие. Пусть уровни полезного сигнала и зеркальной помехи на входе преобразователя частоты 2 равны P_c и P_з соответственно. На выходе преобразователя частоты 2 полезный сигнал на частоте ω_e имеет уровень P_2c K_2уP_c, где K_2у коэффициент усиления преобразователя частоты 2 (в режиме усиления без преобразования частоты). Составляющая полезного сигнала на промежуточной частоте P $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{2пч}}$ = K_2пP_с, составляющая зеркальной помехи на промежуточной частоте P $\genfrac{}{}{0ex}{}{\text{(з)}}{\text{2пч}}$ = K_2пP_з, где K_2п коэффициент преобразования преобразователя частоты 2. На выходе первого ФПЧ 3 уровни сигнала и помехи: P_Зс=K_зP $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{2пч}}$ = K_зK_2пP_с, P_Зз=K_зP $\genfrac{}{}{0ex}{}{\text{(з)}}{\text{2пч}}$ = P_зK_зK_{2п =} где K_з коэффициент передачи ФПЧ 3. На выходе фазовращателя на ± 90^o 4 P_4c P_3c K₃K_2пP_c, P_4з P_3з K₃K_2пP_з, так как фазовращатель на ± 90^o 4 не изменяет уровней. На выход смесителя 8 проходят только полезный сигнал и зеркальная помеха на своих частотах без изменения уровня, т.е. P_10c P_2c K_2уP_c, P_10з P_2з K_2уP_з. На выходе смесителя 8 уровни сигнала и зеркальной помехи на промежуточной частоте равны: P_8c K_8пP_10c K_8пP_c и P_8з K_8пP_10з K_8пK_2уP_з, где K_8п коэффициент преобразования смесителя 8. На выходе второго ФПЧ 9 P_9c K₉P_8c K₉K_8пK_2уP_c и P_9з K₉P_8з K₉K_8пK_2уP_з, где К₉ коэффициент передачи ФПЧ 9. На выходе сумматора 5 уровень полезного сигнала с учетом синфазности составляющих P_вых.с P_4c + P_9c (K₃K_2п + K₉K_8пK_2у) P_c. При идентичных ФПЧ 3 и 9 K₃ K₉ K_ф и Р_вых.с K_ф (K_2п + K_2уK_8п) P_c. Коэффициент передачи заявленного устройства по мощности

. Уровень зеркальной помехи на выходе сумматора 5 с учетом противофазности составляющих P_вых.з P_9з P_4з K_ф (K_8пK_2у K_2п) P_з. Компенсация зеркальной помехи, т.е. P_вых.з 0, имеет место если выражение в скобках обращается в нуль. Отсюда оптимальный коэффициент преобразования смесителя 8

,
где K_2п и K_2у коэффициент преобразования и коэффициент усиления преобразователя частоты 2 соответственно. Так как K_2п < K_2у, то

,
например,

. (Следовательно, смеситель 8 может выполняться как на транзисторах, так и на диодах). С учетом (1) коэффициент передачи по мощности устройства в целом
K_Σ= 2K_2пK_ф (2),
что вдвое больше, чем у известного устройства.A useful signal with a frequency ω _e and phase Φ _e and (or) a mirror noise with a frequency ω _s and phase Φ _s are fed through a band-pass filter 1 to the input of a frequency converter 2, where as a result of interaction with a local oscillator 6 with a frequency ω _g and phase Φ _{g the} component of the useful signal is formed at the intermediate frequency ω $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{be}}$ = ω _g -ω _c (the case of the upper tuning of the local oscillator) with the phase Φ _g -Φ _c and the component of the mirror noise at the intermediate frequency ω $\genfrac{}{}{0ex}{}{\text{(H)}}{\text{be}}$ = ω _s -ω _g with phase Φ _s -Φ _g . In addition, in the frequency converter 2 there is an amplification of the useful signal and specular interference as in a conventional amplifier without frequency conversion. The broadband output circuit of the frequency converter 2 provides the selection of all these components. The useful signal and the mirror noise at their frequencies freely pass through the filter plug 10 to the input of the mixer 8, where as a result of interaction with a local oscillation with a frequency ω _g and phase

a component of the useful signal is formed at an intermediate frequency ω $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{be}}$ = ω _g -ω _c with phase

and the component of specular interference at an intermediate frequency ω $\genfrac{}{}{0ex}{}{\text{(h)}}{\text{be}}$ = ω _s -ω _g with phase

. These components through the second HPF 9 are fed to one input of the adder 5. Oscillations at an intermediate frequency from the frequency converter 2 through the first HPF 3 are fed to the input of the phase shifter ± 90 ^o 4, where they receive an additional phase shift of 90 ^o . As a result, on the other input of the adder 5 there is a component of the useful signal at a frequency ω $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{be}}$ = ω _g -ω _c with phase

and component of specular interference at frequency ω $\genfrac{}{}{0ex}{}{\text{(h)}}{\text{be}}$ = ω _s -ω _g with phase

. Therefore, at the inputs of the adder 5, the components of the useful signal are in phase, and the components of the mirror noise are out of phase, i.e. the phase condition for compensating for mirror noise is satisfied. (The above reasoning is valid for the upper tuning of the local oscillator when ω _g > ω _e . For the lower tuning of the local oscillator, when ω _g <ω _e , all arguments remain valid if the phase shift in the phase shifter by ± 90 ^o 4 is -90 ^o ). To compensate for mirror noise, it is necessary that the levels of the components of mirror noise at the inputs of the adder 5 be the same, i.e. it is necessary to fulfill the amplitude compensation condition. We state this condition. Let the levels of the useful signal and the mirror noise at the input of the frequency converter 2 be equal to P _c and P _s, respectively. At the output of the frequency converter 2, the useful signal at a frequency ω _e has a level of P _2c K _2y P _c , where K _{2y is} the gain of the frequency converter 2 (in amplification mode without frequency conversion). The component of the useful signal at the intermediate frequency P $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{2pc}}$ = K _2п P _с , component of specular interference at the intermediate frequency P $\genfrac{}{}{0ex}{}{\text{(h)}}{\text{2pc}}$ = K _2p P _s , where K _{2p is} the conversion coefficient of the frequency converter 2. At the output of the first PLL 3, the signal and interference levels are: P _Ss = K _s P $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{2pc}}$ = K _s K _2p P _s , P _Sz = K _s P $\genfrac{}{}{0ex}{}{\text{(h)}}{\text{2pc}}$ = P _s K _s K _{2p =} where K _{s is} the transfer coefficient of the PLL 3. At the output of the phase shifter, ± 90 ^o 4 P _4c P _3c K ₃ K _2p P _c , P _4s P _3z K ₃ K _2p P _s , since the phase shifter ± 90 ^o 4 does not change levels. The output of the mixer 8 passes only the useful signal and the mirror noise at their frequencies without changing the level, i.e. P _10c P _2c K _2y P _c , P _10c P _2s K _2y P _c . At the levels of the signal output of the mixer 8 and mirror noise on the intermediate frequency equal to: P _{8n 8c} K K P _10c P _{c 8n} and _8p P K P _{8z 10h 8n} K K P _{2y s,} where K _8n coefficient conversion mixer 8. The output of the second FPH _9c K 9 P P _{9 9 8c} K K K _{8n 2y} P _c and P _9z K ₉ P _{8z 9} K K K _{8n 2y} P _s, where K is _9, the transmission ratio FPH 9. The output of the adder 5, the desired signal level with the in-phase component P _out.with P _4c + P _9c (K ₃ K _2n + K ₉ K _8n K _2y ) P _c . With identical phase-to-phase converters 3 and 9 K ₃ K ₉ K _f and P out. _S K _f (K _2p + K _2y K _8p ) P _c . The transmission coefficient of the claimed device by power

. The level of interference at the mirror output of the adder 5 with the antiphase components _vyh.z P P P _{9z 4h f} K (K _{8n 2y} K K _2n) P _s. Compensation for specular interference, i.e. P _{out of} 0, occurs if the expression in parentheses vanishes. Hence the optimal conversion coefficient of the mixer 8

,
where K _2p and K _2y conversion coefficient and gain of the frequency converter 2, respectively. Since K _2n <K _2y , then

,
eg,

. (Therefore, the mixer 8 can be performed both on transistors and on diodes). Given (1) the transmission coefficient for the power of the device as a whole
K _Σ = 2K _2n K _f (2),
which is twice as much as the known device.

(3).We estimate the noise properties of the claimed device using the standard method for determining the noise figure of a multistage circuit. Total output noise intensity:
h $\genfrac{}{}{0ex}{}{\text{*}}{\text{out}}$ = H ₂ K _2n K ₃ + H ₂ K _2y K _8n K ₉ + H ₈ K _8n K ₉ ,
where H ₂ , H ₈ the noise intensity of the frequency Converter 2 and mixer 8, respectively. The first term is the noise of the frequency converter 2, passed through the first phase-transfer filter 3. The second term is the noise of the frequency converter 2, passed through the mixer 8 and the high-pass filter 9. The third term is the noise of the mixer 8, taking into account the passage through the high-pass filter 9. Taking into account (1) and that K ₉ K ₃ K _f , the total intensity is written in the form:

(3).

For a known device, a similar value
h _out 2H ₂ K _2p K _f (4).

On the other hand, for the claimed device h $\genfrac{}{}{0ex}{}{\text{*}}{\text{out}}$ = H ^* K, where H ^* is the noise intensity of the input source, which creates the same noise intensity at the output under the assumption that the mixers are noise-free, or, taking into account (2)
h $\genfrac{}{}{0ex}{}{\text{*}}{\text{out}}$ = H ^* K _2n K _f (5),
and for a known device
h _out HK _2n K _f (6).

(7).Based on (4) and (5), the noise intensity of a two-stage frequency converter with suppression of the mirror channel has the form:

(7).

It follows that the resulting noise depends on the gain of the frequency converter 2 and decreases with its increase. From (4) and (6) a similar value for the known device
H 2H ₂ (8),
those. independent of gear ratios.

(9),
где F₂ и F₈ коэффициент шума преобразователя частоты 2 и смесителя 8 соответственно. Для известного устройства коэффициент шума
F 2F₂ (10).The noise figure of the claimed device is written in the form:

(nine),
where F ₂ and F _{8 are} the noise figure of the frequency converter 2 and mixer 8, respectively. For a known device, the noise figure
F 2F ₂ (10).

, тогда, полагая для простоты F₂ F₈ 1, имеем

(11).We define the gain by noise figure as

, then, assuming for simplicity F ₂ F ₈ 1, we have

(eleven).

, становится заметным при K_2у ≥ 1 и увеличивается с ростом K_2у, но не более, чем в два раза. Соответственно возрастает отношение сигнал/шум на выходе, но не более, чем в четыре раза.It follows that the gain in noise figure appears when

, becomes noticeable at K _2у ≥ 1 and increases with K _2y , but no more than twice. Accordingly, the signal-to-noise ratio at the output increases, but no more than four times.

 The gain is due to the fact that the claimed device implements a two-stage frequency conversion circuit (instead of the two-channel in the known device), in which the total noise figure depends on the gain of the first stage, and with a sufficiently large value of the latter, the noise of the second stage is weakly manifested.

 Thus, the claimed device as effectively suppresses the mirror channel as the known device, and at the same time has a noticeable gain in the value of the noise figure and the signal-to-noise ratio.

Claims (1)

Two-stage frequency converter with mirror channel suppression, comprising a series-pass bandpass filter, a frequency converter, a first intermediate frequency filter, a ± 90 ^o phase shifter and an adder, a local oscillator, a 90 ^o phase shifter, a mixer and a second intermediate frequency filter, the output of which is connected to another input of the adder, while the other input of the frequency converter is connected to the output of the local oscillator, characterized in that between the output of the frequency converter and the other input a filter plug is introduced into the mixer house, and the frequency converter is made in the form of a balanced frequency converter with a gain greater than unity, and the mixer conversion coefficient is equal to the ratio of the conversion and gain of a balanced frequency converter with a gain greater than unity.