NZ275269A - Audio signal scrambling/descrambling, scrambled audio mixed with two fixed frequency signals - Google Patents

Abstract

Audio signals are descrambled by double sideband modulating the scrambled audio signal with a modulation carrier having a carrier frequency slightly above the highest audio signal present in the scrambled audio. This produces a double sideband signal that is passed through a low pass filter which in turn is modulated by a second carrier frequency lower than the first carrier signal by equal to the offset spectrum of the original scrambled signal. The first low pass filter nulls out any residual carrier from the first modulator that results form the intermodulation of the two modulation frequencies that would be audible at its descrambler output. The modulators used are low noise switch type modulators that improve the signal to noise ratio in the descrambled signal over the previously used linear modulators. The use of switch type modulators provides a lower cost device with improved performance. A companion scrambling device uses similar techniques to provide improved performance at a lower cost.

Description

New Zealand No. 275269 International No. PCT/US94/11891 TO BE ENTERED AFTER ACCEPTANCE AND PUBLICATION Priority dates: 14.12.1993; Complete Specification Filed: 18.10.1994 Classification:^) H04K1/04 Publication date: 26 August 1998 Journal No.: 1431 new zealand patents act 1953 COMPLETE SPECIFICATION Title of Invention: Method and apparatus for scrambling and descrambling of Signals Name, address and nationality of applicant(s) as in international application form: MACROVISION CORPORATION, 700 El Camino Real East, Mountain View, CA 94040, United States of America . \ 0\ : EPA Mt'F.\CHE\ U 1 • '• 9-11-95 : 20: CM : + 08740861 0— -49 89 2C-ZV+46Z ■ a :j 11 METHOD AND APPARATUS FOR LOW COST AUDIO SCRAMBLING AND DESCRAMBLING by Ronald Quan BACKGROUND This invention relates to techniques for 'Sow cost scrambling and descrambling of audio information signals. More particularly, this invention relates to a lower cost Hi Fi dsscrambler with an improved performance over the prior art.

The prior art in the art of audio scrambling and descrampling utilized various 10 frequency shifting techniques. The prior arts in audio descrambling suffer from hiss in the form of "white noise", and more importantly In band carrier "whistle caused by inter-modulation of the two carrier frequencies. The prior arts also use expensive circuitry such, as band pass filters for mixer circuits, wide band 0 degr©9 and 90 degree all pass networks and 0 degree and 90 degree circuits for varying the carrier 15 frequencies with constant amplitude and the need for adjustments to balance gain of quaarature mixers for sideband elimination. In addition, since the mixers used in the prior art are generally not stable in time, their drift results in an audibie whistle as the result of carrier leak through.

The prior art requires mixers that require a pure sine wave modulation, 20 therefore a truly analog multiplier is needed. Truly analog multipliers tend to have noise problems because of their circuit configuration that C2usa white thermal or shot noiss components that degrade the signal to noise (SNR) af the audio scrambling system.

Prior art systems having one or more of the identified problems include US 25 patents # 4,636,353 ('353), DYNAMIC AUDIO SCRAMBLING SYSTEM, by Forbes issued on Jan. 13, 1987, #5,058,159, METHOD AND SYSTEM FOR SCRAMBLING AND DESCRAMBLING AUDIO INFORMATION SIGNALS by Quan issued on October 15, 1991 and # 5,159,631, AUDIO SCRAMBLING SYSTEM USING IN BAND CARRIER, by Quan st al. issued on October 27, 1992 ('153). 30 A review of the prior art for a full understanding of the present invention will be helpful. Turning now to •-.the drawings, Fig 1 is a block diagram of the key ems of the Forbes '553 prior art. The Forbes '853 descrambler 10 has a scram L.sd audio input 34 which is connected to an all pass phase shifter 20 16 JAN "327 CEt^ una®® 27526 containing a 0 dag. output 3B and a 90 deg. output 39. The scrambled audio signal has an offset frequency 35 t'1- f2 as shown in Fig. 2a. This shows the scrambled audio offset by an offset frequency determined by the scrambling process. The phase shifted outputs are connected to a first input of linear modulators 21 and 27.

A frequency generator 22 generates a square wave frequency (fi) which is fed to band pass filter 24 to remove any harmonics, thus producing a pure sine wave. This f-. sine wave is connected to a 0 deg. and SO deg. phase shifter 25. The outputs of phase shifter 25 are in turn connected to second inputs of linear modulators 21 and 27 respectively. The outputs of the first and second linear modulators are added in summer 28 to produce signal 37. This output signal 37 is connected to a first input of a second mixer 30 via high pass filter 29 which passes only fi and the upper sideband as shown in Fig. 2b.

A second square wave frequency generator 23 generates a signal f2 as shown Figs, 1 and 2b. This square wave is filtered by band pass filter 26 to remove any harmonics to produce a pure sine wave signal. This pure sine wave signal is connected to a second input of third mixer 30. The output of the third mixer 30 is connected to a low pass filter 31 to produce a descrambled output signal 35.

The second spectral diagram in Fig. 2b shows the input to the 3rd mixer 30. The frequency fi here represents the residua! carrier feed through from mixers 21 and 27. Fig. 2c shows a shows the relationship of a carrier f2 to fi in Fig. 2b and the scrambled audio-signal shown in Fig. 2a. Fig 2d, shows the relationship of the spectra] characteristics of the descrambled signal 35 and the residual difference frequency (f<-f2) component to the spectral characteristics of the signals in Figs. 2a-2c.

The Forbes encoder uses sine wave type modulators for their carriers. Switch type modulators as disclosed in the instant invent produce a lower white noise component and do not require bandpass filters for filtering the carriers. See <*4 and 62 in Figure 1 of Forbes.

Forbes', decoder assumes that their mixers or multipliers have no noise or carrier feed-through. All practical mixers or multipliers have residual white random .Ogisa and carrier leakage from parasitic capacitance and small mismatches of circuit o\ 2 " ■--? "I "" / \ 0'.: EPA \ILE\CHE\ 0 1 i: - 'do ■ Oo 403 t 30 1 0— 83 — 399'i-l-6o • u elements within(i.e. transistor offset voltages).Forbes thus does not take into account of the residual carrier feed through particularly from the first mixer output :c cause in-cand whistle tones caused by the intermodulation of the two carrier -requencies and combinations of harmonics of the two carrier frequencies used in 5 decoding tne scrambled audio signal. The instant firstly takes into account of residual carrier feed through as a problem to high quality (signal to noise ratio) decoding. It identifies primarily that the carrier feed through from the first mixer or multiplier must follow with a specific filter. This filter after the first mixing step must filter out substantially the carrier feed through and harmonics of the carrier 10 feed through in order to have a whistle free decoded output. Forbes does not idemify this problem.

Secondly, there was is a need to reduce random noise via switch type mixers versus the sine wave mixers in Forbes. Switch type mixers will out perform the anaiog types by a substantial amount(i.e.>10 db). Forbes does not identify the 15 random whne noise characteristics of their multipliers or mixers.

Fig. 4 shows the scrambled audio input of the Quan prior art descrambler 11. This shows the scrambled audio 40 offset by an offset frequency determined by the orig,nal scrambling process. The scrambled audio input signal 40 is connected to an ail pass shifter 41 which provides 0 deg. and 90 deg. phase shiftBd outputs 42 and 20 43 to first inputs of first and second mixers 44 and 45.

Carrier frequency generator .46 generates a sine wave signal fc 47 with a frequency of 1 Khz or 2-3 khz. The carrier frequency 47 is filtered by a low pass filter 48 to remove any harmonics to produce a pure sine wave 49. This pure sine wave ssgnal 49 is connectsd io an ali pass phase shifter 50 to produce 0 deg. and 25 90 deg signals 51 and 52 which in turn are connected to second inputs of mixers 44 and 45. The outputs of mixers 44 and 45, signals 53 and 54 are connected to summer 55 to produce descrambled output 56.

Fig. 4b shows the relationship of the in band descrambiing carrier fc to the scrambled audio signal. Figure 4c shows the descramblsd audio spectrum with the 30 residual carrier fc that is typically -60 db below the descrambled audio program, but .still audible during silent passages of the audio program.

% It is therefore an object of at least a preferred embodiment of this invention to rovide a higher performance descrambler and/or lower cost frequency shifted scrambled audio signals. The method and apparatus described set out to achieve the following aims or at least provide the public with a useful choice: 1) eliminates the use of 0 degree and 90 degree phase shift circuits, 2) eliminates the use of quadrature mixer circuits, 3) eliminates the need for band pass filters or low pass filters for the modulating carrier, 4) reduces white noise and cost using switching type mixer circuits instead of linear mixers, 5) eliminates in-band audible whistle via filtering out the residual first carrier whistle. 6) eliminates the need to adjust mixers for minimum in-band carrier whistle and 7) since the SNR has been improved the need for noise reduction circuits has 10 been eliminated.

BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a block diagram of the key elements of the Forbes prior art; * I Fig. 2 is a spectral diagram of the system in the Forbes prior art; Fig. 3 is a block diagram of the key elements of the Quan et si.' prior art; 15 Fig. 4 is a spectral diagram of the Quan et al. prior art; Fig. 5 is a block diagram of the preferred embodiment; Fig. F:sa spectral diagram of the preferred embodiment of the descrambler described on Fig. 5; Fig, 7 is block diagram of a switch type low noise modulator; 00 Fig. 8 is a block diagram of a first implementation a descrambler using the concepts of the invention; Fig. 5 is a block diagram of a second implementation of a descrambler using the concepts of the invention; ^ Fig. 10 is a block diagram of a third implementation of a descrambler using ^5 the concepts of the invention; and Fig. 11 is a block diagram of a preferred embodiment of scrambler using the concepts of the invention; Fig. 12 is a spectral diagram of the scrambler described in Fig. 11; and Fig. "^showa implementations of the 1st and 2nd low pass filters of the 30 invention. r SUMMARY intellectual property office of n.z. -< IX S23 RECEIVPO _.r-\ \fv : cpA Wl E\CHE\ 01 J. 9-13-95 : 20 = 07 : +0874-38610- -4-9 89 239944So:* > 275269 The present invention is directed to a method and system for descrambling frequency shifted scrambied audio signals that satisfies the nesds>described above. Th9 invention comprises a method and sysiem for descrambling frequency shifted scrambles audio signals.

The descrambling system described descrambles a scrambled frequency translated audio information signal by generating a modulation carrier signal at a frequency lying outside the original frequency spectral range of an scrambied audio Signal of about 50 Hz to about 15 Khz by first generating a first modulation carrier signal having a frequency greater than the highest frequency in the original audio 10 signal. Tnis first modulation carrier is used for double sideband modulating the scrambled audio signal into a first modulation frequency, a first upper sideband signal and a first lower sideband signal. This set of signals is filtered by s filter to filter out the first modulation frequency, al! its harmonics, and the upper sideband signal and its harmonics from the double sideband signal and passing the first 15 lower sideband signal.

A second modulation carrier frequency having a frequency less than the first modulation frequency is generated. This second modulation frequency is connected to a second modulating means for double sideband modulating the first lower sideband signal with the second modulation carrier frequency to produce a second 20 modulating frequency, a second upper sideband signal and a second lower sideband signal.'- A second filter passes the second lower sideband signal to produce a descrambled audio signal.

The modulators used are low noise switch type modulators that improve the 25 signal to noise ratio in the dascrambled signal over the previously used linear modulators. Tne use of switch type modulators provides a lower cost device with improved performance.

A companion scrambling device uses similar techniques to provide improved performance.at a lower cost. The method of scrambling of an original audio signal of 30 about 50 Hz to about 15 Khzcomprises: generating a fir3t modulation carrier signal "—"having a frequency greater than the highest frequency in the original audio signal; quadjf&ture modulating said original audio signal into a first lower sideband signal; t - *, 5 EPA MLENCHEN <-'] ; 3-11-95 : 20:07 : 4-08743CC10- ~49 89 23994-4C3: 27 5281; filtering out the first modulation frequency and all its harmonics, at least pari of the upper sidaband signal and all the harmonics from the modulated signal and passing said first lower sideband signal; generating 2 second modulation carrier frequency having a frequency higher than the first modulation frequency; double siasoand modulating the first lower sideband signai with the second modulation carrier frequency to produce a second modulating frequency, a second upper sideband signal and a second .lower sideband signal; filtering the second modulating frequency, part of the second upper sideband signal and the second lower sideband signal to pass the second lower sideband signal to produce a scrambled audio signal.

From a method standpoint, the invention broadly comprises frequency translating the original spectrum of audio information signals to produce scrambled audio :nformation signals by generating a modulation carrier signal having a frequency lying outside the frequency spectral range of the audio information signals, and first single side band modulating followed by double side band modulating the original information signals with the modulation carrier signal to translate the frequency of the original audio information signal in a given direction. Preferably, ihe frequency of the modulation carrier signal(s) are varied during generation in a pseudo random fashion, particularly by sweeping the frequency of 0 the modulation carrier signal between predetermined limits. The step of varying the frequency of themodulation carrier signal preferably includes the steps of initiating a frequency varying operation in response to a first control signal at a rate determined by a second control signal.

For a fuller understand of the nature and advantages of the invention, reference should be made to the ensuing detailed description taken in conjunction with the accompanying drawings.

DESCRIPTION Fig. 5 shows a block diagram, and Fig. 6 shows a spectral diagram of the preferred embodiment of the instant disclosure. Fig. 6a shows the spectral 0 ^..-cb^rac:eristic of the scrambled audio input of the preferred embodiment. Tnrs shows t^ scrambled audio offset by an offset frequency determined by the scrambling process^Eig. 6b shows the relationship of the first mixer's carrier and the output of 6 '7i 6 w ''■> .. J t )t w ' IfcVtt®*3 :C'. VO\ : EPA M1.E\CHE\ 01 : 9-11-95 : 20 ■ 08 : 40874-138010- -r-49 B9 23 994-4-G5:* 9 2 the firs; mixer. Both the upper and low9r sidebands and the residual carrier fA plus the harmonics of all of tnese are at the first mixer's output. Fig. 6c shows the filter characteristics of the first LPF following the first mixer's output. It is crucial that this nrst LP" filter out the residual carrier and its upper sideband harmonics. Fig. 5a the 5 output of the spectral characteristic of the output of the first LPF following the first mixer's output.

Fig. 5e shows the relationship of the second carrier to the output of the first LPF to form the last descrambling step. Fig. 6f shows the relationship of the descrambled audio that has passed through a 2nd LPF with a 12 khz cut-off to filter 10 out fg and its upper sideband above fg with the absence of whistle frequency component (fa-fb). The (fa-fb) whistle frequency component is typically equal or less than -85 db in the descrambled audio.

In this preferred embodiment fA is about 19 Khz and fe is about 15.4 Khz. These choices are for economy, since with these frequencies the first LPF can be 15 designed inexpensively. If increased performance at a greater cost is desired, the carrier frequencies can be higher in order to minimize leakage of components from the scrambled audio input so as to not interfere with the lower sideband output of the first mixer. Note that in Figs. 6a and 6b there is an overlap between the spectra of the lower sideband frequencies and the scrambied audio frequencies. If the first 20 mixer feeds through enough of the scrambled audio, distortion products will occur at the descrambled-output. By setting the carrier frequencies to for example fA = 39 Khz and fg =36.4 Khz scrambled input leak through will not cause distortion products at the descrambled output since it will not overlap with the lower sideband of the first mixer i.e. 36.4 Khz. to 24 Khz. versus 2.6 Khz to 14.6 Khz of the 25 scrambled input. However raising fA and fB two fold causes the steepness of- the first LPF to increase to about two fold. This would require higher order filters such as a 10 pole elliptical low pass filter.

Minimal carrier leakage and scrambled audio leakage with lower shot noise is achieved byusrng a double throw single pole analog switch such as the 74 HCT 30 4053 cr its equivalent i.e. MC1496 switch type mixer with a carrier input equal to or ^-^-•^---.Tiore than 350 mv p-o. ir"c' V ' > 7 J/i;i;;~7 r"; " j f 0-^ \0\-EPA M.i.'E\CHc\ 01 : 9-] 1 -95 : 20:08 : 4087438610- -4-U 89 2 3994-465: tt 1 0 275269 It was found, for instance, with a CD 4053 analog switch the "on" resistance resulted in a measured noise of 2.5 nv/ /Hz which translates into a noise resistance ;/4kT3r = Vm = 2.5 nv//Hz, 3 = 1 Hz, T = 298° Kelvin, k = Bolzmar.'s constant and R - noise resistance) of 400 chms. The "on" resistance of the CD 4053 was 5 measured to be 440 ohms. Thus it was found experimentally that the "on" resistance cf the analog switch (i.e. 4053) produces the same amount of noise as a resistor component of the same resistance. Thus an "on" resistance of 440 ohms in a CD4053 has essentially the same noise as a 440 ohm resistor.

Linear modulators such as the AD 534 produces 0.6 mv RMS over a 10 Khz 10 bandwidth or a noise density of 0.6 mv / /10 Khz = SO nv / /Hz. Therefore the AD 534 linear modulator produces approximately 60/2.5 more noise than the CD 4053 switch. This is eauivalent to a 27 db improvement when using a CD4053 over a iinaar modulator.

Gilbert modulators such as the 1496 or 1495 will produce low noisa, i.e. < 5 15 nv/ /Hz when the carrier input of these devices switch the differential pairs on and off. This is achieved by either overriding the carrier input with a square wave carrier input with a square wave of > =/- 200 mv or a large sine wave of > 1 v pp. When sinusoidal modulators such as the 1495 does not have the carrier inputs over driven to produce linear modulation, the noise is substantially higher versus a switch mode 20 1496 modulator. This is because the 2 differential pair transistors start amplifying their own noise. The internal base resistance of each transistor is usually about 50 - 200 ohms, if one assumes a 100 ohm series internal base resistors on the 2 pairs cf differential pair transistors in series in a 1495 and 1 kohm load for one output a further assume that each of these transistors has a quiescent bias of 1 ma collector 25 current the output noise is then equal to 1/2 * 1000 (gm) V nr = Vp noise, gm = 38 maA/ "'or an lc = 1 ma. Therefore V nr = MOO ohm * 4kT = 2.5 nv/Hz. V0 noise = 19 * 2.5 nv / /Hz = 47.5 nv / /Hz from a 1495 modulator. This is 19 times or 25 db more noise than the CD 4053 with an "on" resistance of 440 ohms. It should be noted that :he output noise decreases in the 1495 or 1496 modulator as the carrier input is ■ncraased.

^Tt /v^%The key to the preferred embodiment is ths use of a Low Pass Filter (LPF) after first mixer which is to reject out a residual carrier from ihe first mixer and - \\ „v ; vtr-j --\ . V0\: EPA. MIE\CHE\ 01 : 9-11-95 : 20:09 : 4-0874-38610- -4 9 8 3 23994-4C5: * 11 27 remove ali sidebands related to harmonics of the carrier and the harmonics of the carrier. If this is not done, harmonics cf the whistle freouency (3fg-3fb)), (5fa-5fb)) and s:c. will appear at the descrambling output in an audible manner. This first LPF .5 generally a 7 pole or mors elliptical filter with at least one zero tune to notch out 5 the first mixer's carrier frequency, fA- In practice a 9 "pole active filter with general impedance convenors is the best choice for a stable and accurate filter. In the rreierred embodiment the 3 db cut off of the first low pass filter is about 17 Khz with at least 40 db attenuation at 19 Khz, A detailed description of the preferred embodiment is described below with 10 reference to Fig. 5. The descrambling apparatus 12 has a scrambled audio signal nput 60 and contains the descrambling process of the preferred embodiment. The scrambled audio 50 is inputted into a first input of a first mixer 63. The second input of this first mixer is a first carrier signal fA generated by frequency generator A, 61 which is approximately 19 Khz. The output first mixer 63 contains carrier feed 15 through of fA, all its sideband components and the harmonics. The output of mixer 53 is fed to a low pass filter 65 that filters out the first carrier, the upper sideband ana all of the harmonics from signal 60. The output of low pass filter 65, signal 66 is fed into a first input of a second mixer 66. The second input of this second mixer is a second carrier signal fs generated by frequency generator B, 62 can be 16.4 Khz 20 of 15.4 Khz +/- 100 Hz shifted pseudo randomly for security reasons. See U.S. Patent 5,095,279 for a further explanation of this security process. Tne output of second mixer 70 contains the baseband descrambled audio, residual second carrier 3nd upper sideband components above fg's frequency. The second low pass filter 71 with a cut-off frequency of approximately 12 Khz removes everything above 12 25 Khz, but passes the descrambled audio to the output line 23.

In the above preferred embodiment the mixers utilize a switch type low shot or thermal noise modulator as described in Fig. 7. The operation cf this .mixer will be described relative to the first mixer. The second mixer operate on the same orinciple. Scrambled audio 60 is fed into the + input of unity gain amplifier 73. Tne 30 output of amplifier 73 is fed on line Vjn 74 to one input of a double pole single throw V A ' §h$To,g switcher 32. The output of 73 is also red to the input of unity gain inversion fz'A*' ci\ . . ".mplifte^consisting of R2a, R2b, and amplifier 65.. The output of amplifier 55 is -V'm 9 -v VO\: EH A Ml. ENC'HEN 01 ■ 0-11-S3 : 20:09 ' 4-087438C10— -49 B9 2'J\.-j944^.- • 27526 75 which is fed to a second input of the switcher 32. First carrier frequency fA is fed into the switching control input of the double pois, single throw switcher 32. The couola pels, single throw switcher used is 1/3 of an 74HCT4053 or its equivalent and is fed to amplifier A220. A220 is the mixer output. For minima! carrier leakage of 5 tne output of mixer 55, the DC zero signal voltage of the two inputs of switch 32 V'm anc -Vjn must ba exactly the same, i.e. Ov. In addition th9 inversion amplifier 73 must be a -1 unity gain to have minimum scrambled audio in ( Vin) feed through. Thus F*2a = R2b within 1% or better is required for a wids band op amp 65 (i.e. NE5532.

Fig. 13 a shows a conventional RLC low pass filter with zeros for the descrambier's first low pass filter. Because inductors L*) through L3 are rather large 2 milli-henri9s through 20 milli-henries to achieve a low cost. These lower cost inductors suffer from a just adequate Q at audio frequencies. Much more expensive inductors with higher Q's will provide better low pass filtering, but will be beyond the ' 5 budget of a low cost descrambling system.

Fig. 13 b shows an active 9 pole elliptic low pass filter that is not as sensitive to parts tolerance as many other active filters. This is important since fa, the first carrier frequency must be filtered out by at ieast -40 db attanuation. Fig. 13 b is a General Impedance Converter (GIC) active low pass filter that was found to provide 20 very high parformance in filtering at low cost. The capacuors can be inexpensive 5% mylar film capacitors. The resistors are inexpensive 1% resistors and the operational amplifiers can be common type such as TL082, NE5532 etc. rig. 13 c shows an example of the 2nd filter as an active 7 pole low pass filter. Amplifiers A1000, A2000 and A3000 can be simple voltage followers of 25 common operational amplifiers or single transistor emittar followers. The 2nd filter in the descrambler can any low pass filter, passive or active with sufficient stop band attenuation to provide a descrambled audio signal without measurable artifacts such 2nd carrier lone its upper sidebands and / or audible artifacts.

Figs, .8 - 11 show various implementations using the concepts of the 30 invention.

T £ In addition to a descrambling system as described above many of the same elements can De used in a scrambler to achieve many cf the same advantages ?D \ 10 .1 /1' // // V. EP i. MLE.\CHi£\ 01 9-ii-SJrj : -{> 10 ; 4-08 <4U861 U- -•! 9 89 _';i994-40C. • » 1: 2752 achieved in tha descrambler described above, i.e. lower shot noise output and less filter requirements than ths prior art such as Forbes ('S53). Fig. 11 is a block diagram and Fig. 12 is a series of spectral diagrams of a preferred embodiment of s scrambler.

An audio signal with a spectra! response of about 30 Hz, to 15 Khz. 91 is fed :,n:.c a iow pass filter 92 to eliminate any unwanted signals beyond 15 Khz. The output of iow pass filter 82, 93, is connected to 0 deg. and 90 deg. all pass phase shiftars 94 and 95. The outputs of phase shifters 94 and 95 are in turn connected to first inputs of switch type low noise modulators 96 and 97. *0 Signal generator 98 generates a square wave signal at approximately 16.4 Khz. with 0° and 90° outputs which are connected to second inputs of modulators 95 and 37. The outputs of modulators SS and 57 are summed to produce signal 103, a quadrarture modulated signal resulting in a residual 15.4 Khz. carrier with a lower sideband. Fig. 12 shows the relationship of the quadrature modulated audio 15 components to the original audio signal 91.

This quadrature modulated signal is fed through low pass filter 104 as signal 105 and is essentially the same filter as the first filter of the descrambler described above. This signal is connected to a first input of a third modulator 106. Modulator 106 is a switch type low thermal or shot noisa modulator as described above and in 20 shown in Fig. 7. A second carrier frequency is generated by a square wave osciliator 99 generating a frequency of approximately 19 Khz. as shown in Fig. 12 e.

The output of modulator 106 contains a 19 khz carrier and upper and lower sidecands. This signal is filtered by low pass filter 107 to produce a scrambled audio signal with an offset cf approximately 2.6 Khz.

Theoretically, to decrease the dynamic artifacts caused by fast step frequency changes of the 16.4 Khz carrier in both the scrambler and descrambler, the low pass filters following the first quadrature mixer the first mixer- of both the scrambler and descrambler respectively should be very nearly identical in group delay responses (transient responses). If the transient response characteristics of the low pass filters in the scrambler are different from the transient characteristics of ? h ' the,descrambler, the step changes of the 16.4 Khz carrier has to be slowed down to // v o i/* achieveyminimal descrambling artifacts. It is preferred to have faster step changes ML"£\CHE\ 01 9-11-95 : 20:n : 40874-38610- ■4-4-9 69 23994-465:# 27526 in the securad carrier (16 Khz +/- 100 Hz) and have the first low pass filter in the descrambler have the same characteristics as filter 104 in the scrambler of Fig. 11. In addition, the second low pass filter in the descrambler should have the same characteristics of filter 107 of the scrambler of Fig. 11. This permits the step shifting specirum of the scrambler to be tracked quickly in the dsscrambler without artifacts caused by time delay skews between scrambler and descrambler tracking the 16 Khz stepped deviations. It should be noted that all carriers for all mixers in this invention for descramblers and scramblers should be preferably square wave signals for minimum artifacts.

While the above provides a full and complete description of the preferred embodiment of the invention, various modifications, alternate constructions and equivalents will, occur to these skilled in the art. Therefore , the above descriptions and illustrations should not be construed as limiting the scope of the invention, which is defined by the appended claims. °£fv e® 12 U\:E?A ML'ENCHEN 01 : 20 : 11 : 4U874-38C10- +49 80 £3994-4 Go : 27

Claims (2)

1. CLAIMS What is claimed is: A system of descrambling a scrambled frequency translated audio signal capable of reproducing a spectral range of an original audio signal of about 50 Hz to 5 scout 15 Khz comprising: a scrambled audio input signal, said signal having a frequency range pf 50 Hertz to substantially 15 Kilohertz, means for generating a first modulation carrier signal having a frequency graater than the highest frequency in the original audio signal, 10 first modulating means for double sideband modulating said scrambled audio signal into said first modulation frequency, a first upper sideband signal and a first lower sideband signal, first filtering means for filtering out said first modulation frequency, all its harmonics, said upper sideband signal and its harmonics from said double 15 sideoand signal and passing said first lower sideband signal, means for generating a second modulation carrier frequency having a frequency less than said first modulation frequency, a second modulating means for double sideband modulating said first lower sideband signal with said second modulation carrier frequency to produce a second 20 modulating frequency, a second upper sideband signal and a second lower sideband signal,'and means for filtering to pass said second lower sideband signal to produce a descrambled audio signal: characterized in that use and selection of said firs' and second modulation 25 carrier signal, said first and second modulation means and second first and second filter means is to produce a descrambled signal containing substantially no_ audible whistle component.
2. 2. A system as in claim 1 wherein said means for generating said first and second modulation carriers comprise square wave generators. 30 3. A system as In claim 1 wherein said first modulating means and second modulating means comprise sv/itch type low noise modulators. 13 <CV. V0\: EPA Ml'ENCHEN. 01 : 9-11-S5 : 20:1 1 : 4-08/4-38010- +49 89 239944-65:^10 2752 4. A system as in claim 1 wherein said first filtering means comprises an elliptical filter containing at least seven poles and a zero to notch out said first carrier frequency. 5 A system as in claim 1 wherein said first filtering means comprises an 5 active filter containing at least seven poles with general impedance convenors. 5. A system as in claim 1 wherein said second filtering means comprises a filter with seven or more poles. I. A system as in claim 1 wherein said first modulation carrier generates a frequency of at least 19 Khz. 10 8. A system as in claim 1 wherein said second modulation carrier generates a frequency at least 500 Hz below the frequency of said first modulation frequency. 9. A system as in claim 8 wherein said second modulation carrier is pseudo randomly varied about +/-100 Hz. 15 10. A system as in claim 3 wherein said first and said second switch type low noise modulators comprise MC 1496 modulators. II. A system as in ciaim 3 wherein said first switch type low noise modulators comprises an analog switch coupled to inverse polarities of said scrambied audio input signal. 20 12. A system as in claim 3 wherein said second switch type iow noise modulators comprises an analog switch coupled to a positive polarity and a negative polarity of said first lower sideband signal. "3. A method of descrambling scrambled frequency spectrum translated single sideband audio information signals capable of reproducing an original audio 25 signal having a frequency range of about 50 Hz to about 15 Khz said method ' comprising the steps of; inputting a scrambled frequency sideband signal audio input signal. generating a first modulation carrier signal having a frequency greater than the highest frequency in the audio input signal, 30 double sideband modulating said scrambled audio signal into said first modulation frequency, a first upper sideband signal and a first lower sideband _ signal, y T E fvf . , rx * r>N 14 V ■% ' <* ta r. t II 18 jr-n rr i! 0\ : EPA MlENCHEX 01 : ,9-11-95 20:12 : 4 087-4-38610- +49 89 239y4-4-65: # 27 filtering out said first modulation frequency, all its harmonics, said upper sideband signal, all its harmonics and al! the harmonics of said lower sideband signal from said double sideband signal and passing said lower sideband signal, generating a second modulation carrier frequency having a frequency less 5 than said first modulation frequency, double sideband modulating said first lower sideband signal with said second modulation carrier frequency to produce a second modulating frequency, a second joper sideband signal and a second lower sideband signal, and filtering out said second modulating frequency, a second upper sideband 10 signal to pass said second lower sideband signal to produce a descrambled audio signal: characterized in that using and selecting said first and second modulation signals, said first and second modulation means and said first and second filter means produces said descrambled audio signal containing substantially no audio 15 whistle component. 14. A method as in claim 13 wherein said means for generating said first and second modulation carriers comprise square wave generators. 15. A method as in claim 13 wherein said double sideband modulators comprise switch type low noiss modulators. 20 15. A method as in claim 13 wherein said filtering out said first modulation frequency ana all its harmonics, said upper sideband signal and all the harmonics from said double sideband signal and passing said icrwer sideband signal uses an elliptical filter containing at least seven poles and a zero to notch out said first carrier fraquancy. 25 17. A method as In claim 13 wherein said filtering out of said first modulation frequency and all its harmonics, said upper sideband signal and ail the harmonics from said double sideband signal and passing said lower sideband signal uses an active filter containing at least seven poles with general impedance converters. 30 18. A method as in claim 13 wherein said second filtering of second modulating frequency, a second upper sideband signal and a second lower .^sidAband signal comprises filter of seven or more poies. ^ r r>\. 15 0 \\ n, <^0^ 1 li 27 52 19. A method as in claim 13 wherein said first modulation carrier generates a frequency of at least 19 Khz. 20. A method as in claim 13 wherein said second modulation carrier generates a frequency less than said first modulation carrier by at ieast 500 Hz. 21. A method es in claim 13 wherein said second modulation carrier ganerates a frequency about 2.5 Khz less than said first modulation carrier. 22. A method as in claim 13 wherein said second modulation carrier generates a frequency that is pseudo randomly varied. 23. A method as in claim 15 wherein said first and said second switch type :ow shot noise modulators comprise switched Gilbert multipliers 24. A method as in claim 15 wherein said first switch type low noise .modulators comprises an analog switch coupled to a positive polarity and a .negative polarity and negative polarity of said scrambled audio input signal. 25. A method as in claim 15 wherein said second switch type low shot noise modulators comprise a analog switch coupled to a positive polarity and a negative polarity of said first lower sideband signal. 26. A method as in claim 15 wherein said first switch type low noise modulators comprise chopper switch modulators. 27. A method as in claim 16 wherein said second switch type low shot noise modulators comprise chopper switch modulators. 28. A' system of scrambling of original audio signal of about 50 Hz to about 15 Khz comprising; an audio input signal with a frequency range of 50 Hz. to 15 Khz., means for generating "a first modulation carrier signal having a frequency greater than the highest frequency in the original audio signal, first modulating means for quadrature sideband modulating said, original audio signal into a first lower sideband signal,. first filtering means for filtering out said first modulation frequency and at least a majority of its harmonics, said upper sideband signal and its harmonics from said quadrature signal and passing said first lower sideband signai, means for generating a second modulation carrier frequency having a frequency higher than said first modulation frequency, 16 S 0\ • EHA MLE.NCHEN 0 1 9-11-95 : 20:13 40874 33010- *"4?J 89 —39944^0:^19 275 second modulating means for double sideband modulating said first lower siosband signal with said second modulation carrier frequency to produce a second modulating frequency, a second upper sideband signal and a second lower sideband signal, and 5 means for filtering to pass said second lower'sideband signal to produce a scrambled audio signal: characterized in that use and selection of first and second modulation carrier signals, said first and second modulation means and said first and second filter means is to produce a scrambled audio signal with a lower noise level and fewer 10 components. 29. A system as in claim 28 wherein said means for generating said first and second modulation carriers comprises a square wave generator. 30. A system as in claim 2B wherein said second modulating means comprise switch type low noise modulators. 15 31, A system as in claim 2B wherein said first filtering means comprises an elliptical filter containing at least seven poles. 32. A system as in 28 wherein said first filtering means comprises a filter without having to tune out said first frequency 33. A system as in claim 28 wherein said first filtering means comprises an 20 active filter containing at least seven poles with general impedance convertors. 34. A-system as in claim 2B wherein said second filtering means comorises a filter of seven or more poles. 35. A system as in claim 28 wherein said first modulation carrier ganerates a frequency of at least 16.4 Khz. 25 36. A system as in claim 28 wherein said second modulation carrier generates a frequency at least 500 Hz above said first modulation carrier. 37. A system as in claim 28 wherein said second modulation carrier generates a frequency that is pseurio randomly varied by about +/-100 Hz. 38. . A system as in claim 28 wherein said switch type low noise modulator 30 comprises a differential pair-balanced multiplier type modulator. 17 (A©®® ML'C\'CHE\ 01 11 -95 20: It +007438010- +49 89 23H94-+C5 : 27 39 A system as in claim 28 wherein said second switch type low noise modulator comprises an analog switch coupled to inverse polarities of said first upper sideband signal. 40. A method of scrambling of an original audio signal of about 50 Hz to about 15 Khz said method comprising the steps cf, inputting an audio signal with a frequency range of 50 Hz. to 15 Khz., generating a first modulation carrier signal having a frequency greater than the highest frequency in the original audio signal, quadrature modulating said original audio signal into a first lower sideband signal, filtering out said first modulation frequency and all its harmonics, said upper sideband signal and all the harmonics from said modulated signal and passing said first lower sideband signal, generating a second modulation carrier frequency having a frequency higher than said first modulation frequency, double sideband modulating said first lower sideband signal with said second moaulation carrier frequency to produce a second modulating frequency, a second upper sideband signal and a second lower sideband signal, and filtering second modulating frequency, said second upper sideband signal and said second lower sideband signal to pass said second lower sideband signal tc produce a scrambled audio signal: characterized in that use and selection of said first and second modulation signals, said first and second modulation means and said first and second filter means is to produce a descrambled audio signal with a lower noise level and fewer components. 41. A method as in claim 40 wherein said first and second modulation earners generators comprise square wave generators. 42. A method as in claim 40 wherein said double sideband modulator comprises a switch type low noise modulator. 43. A method as in claim 40 wherein said filtering out said first modulation frequency and all its harmonics, said upper sideband signal and all the harmonics from said double sideband signal and passing said lower sideband signal uses an : / " 18 I ' < *1^° aa^^' ^ftllipiical filter containing at least seven poles and zero tune notch tuned to notch out ^ said first carrier frequency. At. A method as in claim 40 wherein said filtering out of said first modulation frequency and all its harmonics, said upper sideband signal and all the harmonics from said double sideband signal and passing said lower sideband signal uses ar active filter containing at least seven poles with genera) impedance converters. 45. A method as in claim 40 wherein said second filtering of second moculating frequency, a second upper sideband signal and a second lower 10 sideband signal comprises a filter of seven or more poles. 46. A method as in claim 40 wherein said first modulation carrier generates a frequency of-at least 16,4 Khz. 47. A method as in claim 40 wherein said second modulation carrier generates a frequency at least 50 Hz greater than said first modulation carrier 15 frecuency. 48. A method as in claim 40 wherein said second modulation carrier generates a frequency that is pseudo randomly varied about a frequency substantially equal to 19 Khz. 49. A method as in claim 40 wherein said switch type low noise modulator k20 comprises a differential pair, balanced multiplier type modulator. 50 A method as .in claim 40 wherein said switch type low noise modulator comprises a analog switcii coupled to a positive polarity and a negative polarity of said upper sideband signal. 51. A system of descrambling a scrambled frequency translated audio signal substantially as herein described with reference to figures 5 to 13. 52. A method of descrambling scrambled frequency spectrum translated single sideband audio information signals when performed substantially as herein described with reference to figures 5 to 13. 53. A system of scrambling of original audio signal of about 50 Hz to about 15 Khz substantially as herein described with reference to figures 5 to 13. 54. A method of scrambling of an original audio signal of about 50 Hz to about 15 Khz when performed substantially as herein described with reference to figures 5 to [ INTELLECTUAL OFFICE END w: I -7 MAY W98 41" tfe authorised agents nrpc|\/F nA J fARK & ^ON , receivedpmj