NO891358L - LIGHT INTENSITY CONTROL CLAMP FOR INCLUSIVE LAMPS AND CLUTCH NETWORKS WITH A PROTECTION AND LIMITING CLUTCH FOR AA PROVIDE AN ELECTRONIC SECURITY. - Google Patents
LIGHT INTENSITY CONTROL CLAMP FOR INCLUSIVE LAMPS AND CLUTCH NETWORKS WITH A PROTECTION AND LIMITING CLUTCH FOR AA PROVIDE AN ELECTRONIC SECURITY.Info
- Publication number
- NO891358L NO891358L NO89891358A NO891358A NO891358L NO 891358 L NO891358 L NO 891358L NO 89891358 A NO89891358 A NO 89891358A NO 891358 A NO891358 A NO 891358A NO 891358 L NO891358 L NO 891358L
- Authority
- NO
- Norway
- Prior art keywords
- voltage
- transistor
- light intensity
- resistor
- mos
- Prior art date
Links
- 239000003990 capacitor Substances 0.000 claims description 10
- 230000008878 coupling Effects 0.000 claims description 9
- 238000010168 coupling process Methods 0.000 claims description 9
- 238000005859 coupling reaction Methods 0.000 claims description 9
- 230000009467 reduction Effects 0.000 claims description 7
- 230000005669 field effect Effects 0.000 claims description 4
- 230000007423 decrease Effects 0.000 claims description 2
- 230000009977 dual effect Effects 0.000 claims 1
- 230000007704 transition Effects 0.000 claims 1
- 230000000694 effects Effects 0.000 description 5
- 230000008859 change Effects 0.000 description 4
- 230000001052 transient effect Effects 0.000 description 4
- 230000001965 increasing effect Effects 0.000 description 3
- 230000006978 adaptation Effects 0.000 description 1
- 230000008901 benefit Effects 0.000 description 1
- 230000033228 biological regulation Effects 0.000 description 1
- 230000015556 catabolic process Effects 0.000 description 1
- 238000006243 chemical reaction Methods 0.000 description 1
- 238000013016 damping Methods 0.000 description 1
- 230000001934 delay Effects 0.000 description 1
- 238000001514 detection method Methods 0.000 description 1
- 230000006872 improvement Effects 0.000 description 1
- 230000001939 inductive effect Effects 0.000 description 1
- 238000000034 method Methods 0.000 description 1
- 230000008569 process Effects 0.000 description 1
- 230000001681 protective effect Effects 0.000 description 1
- 230000001105 regulatory effect Effects 0.000 description 1
- 230000001629 suppression Effects 0.000 description 1
Classifications
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
- H03K17/08—Modifications for protecting switching circuit against overcurrent or overvoltage
- H03K17/082—Modifications for protecting switching circuit against overcurrent or overvoltage by feedback from the output to the control circuit
- H03K17/0822—Modifications for protecting switching circuit against overcurrent or overvoltage by feedback from the output to the control circuit in field-effect transistor switches
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
- H03K17/13—Modifications for switching at zero crossing
- H03K17/133—Modifications for switching at zero crossing in field-effect transistor switches
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
- H03K17/16—Modifications for eliminating interference voltages or currents
- H03K17/161—Modifications for eliminating interference voltages or currents in field-effect transistor switches
- H03K17/165—Modifications for eliminating interference voltages or currents in field-effect transistor switches by feedback from the output circuit to the control circuit
- H03K17/166—Soft switching
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
- H03K17/28—Modifications for introducing a time delay before switching
- H03K17/284—Modifications for introducing a time delay before switching in field effect transistor switches
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B39/00—Circuit arrangements or apparatus for operating incandescent light sources
- H05B39/04—Controlling
- H05B39/041—Controlling the light-intensity of the source
- H05B39/044—Controlling the light-intensity of the source continuously
- H05B39/048—Controlling the light-intensity of the source continuously with reverse phase control
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K2217/00—Indexing scheme related to electronic switching or gating, i.e. not by contact-making or -breaking covered by H03K17/00
- H03K2217/0036—Means reducing energy consumption
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps
Landscapes
- Circuit Arrangement For Electric Light Sources In General (AREA)
- Lighting Device Outwards From Vehicle And Optical Signal (AREA)
- Mechanical Operated Clutches (AREA)
Description
I overenstemmelse med svensk patentsøknad nr. 8804209-8 er det foreslått en lysintensitetsstyrekobling i henhold til hvilken man kan unngå de ved vanlige fasestyringer i forbindelse med triacer anvendte midler for gniststøyfjerninger og til tross for dette oppnå tilstrekkelig undertrykkelse av gniststøyspen-ningen med en samtidig bevirket støyfrihet hos styrekoblingen., In accordance with Swedish patent application no. 8804209-8, a light intensity control coupling has been proposed according to which one can avoid the means for spark noise removal used in normal phase controls in connection with triacs and despite this achieve sufficient suppression of the spark noise voltage with a simultaneously effected noise-free control coupling.,
Det særskilte i henhold til den nevnte patentsøknad består i at en selvsperrende felteffekttransistor i en brolikeretters diagonaler gjennom en kontrollkrets styres på en slik måte at felteffekttransistoren leder ved vekselspenningens nullgjennomgang og i samsvar med en innstillbar tid i kontrollkoblingen for et ønsket avsnitt av netthalvbølgen blir i ledende tilstand og deretter med passende innstilt flanke frakobler før neste nullgj ennomgang nås. The special feature according to the aforementioned patent application is that a self-locking field effect transistor in the diagonals of a bridge rectifier is controlled through a control circuit in such a way that the field effect transistor conducts at the zero crossing of the alternating voltage and in accordance with an adjustable time in the control circuit for a desired section of the mains half wave becomes conducting state and then with a suitably set edge disconnects before the next zero crossing is reached.
Den foreliggende oppfinnelse vil nå videreutvikle forslag i henhold til nevnte patentsøknad på en slik måte at lysintensitetsstyrekoblingen kompletteres med en elektronisk sikring. Derved bortfaller det i forbindelse med vanlige dempeanordninger nødvendige skifte av den gjennomsmeltede sikring, f.eks. ved kortslutning forårsaket av gjennombrenning av glødespiralen hos en glødelampe. Ettersom tilpasningen av støy- og transient-pulsbegrensningsorganene samt koblingsanordningen i den forbindelse er nødvendig, kreves i forhold til utførelsen av koblingen i henhold til den nevnte patentsøknad ytterligere koblingstiltak. The present invention will now further develop proposals according to the aforementioned patent application in such a way that the light intensity control link is completed with an electronic fuse. Thereby, in connection with normal damping devices, the necessary replacement of the fused fuse, e.g. in the event of a short circuit caused by burning through the filament of an incandescent lamp. As the adaptation of the noise and transient pulse limitation means and the coupling device in that connection is necessary, additional coupling measures are required in relation to the execution of the coupling according to the aforementioned patent application.
Denne hensikt oppnås i henhold til oppfinnelsen ved de i krav 1 angitte trekk. This purpose is achieved according to the invention by the features specified in claim 1.
Koblingsanordningen i henhold til oppfinnelsen frembyr fordelen av en strømbegrensning ved gjennomkoblet MOS-FET-transistor ved en eventuell kortslutning av lasten eller utladning som følge av en innkoblet forhøyet spenning (transientpulser). En videreutvikling av oppfinnelsen fremgår av de uselvstendige krav. I det følgende beskrives et utførelseseksempel for beskyttelses- og begrensningskoblingen i henhold til oppfinnel- The switching device according to the invention offers the advantage of a current limitation in the case of a through-connected MOS-FET transistor in the event of a possible short-circuit of the load or discharge as a result of a switched-on increased voltage (transient pulses). A further development of the invention appears from the independent claims. In the following, an exemplary embodiment of the protective and limiting connection according to the invention is described
sen ved hjelp av tegningen.then using the drawing.
Fig. 1 viser det kjente koblingsskjerna for lysintensitetsstyrekoblingen hos glødelamper og koblingsnettdeler. Fig. 1 shows the known connection core for the light intensity control connection in incandescent lamps and connection network parts.
"Fig. 2 viser beskyttelses- og begrensningskoblingen ved utførelsen i henhold til oppfinnelsen. Fig. 3 viser en grafisk fremstilling av brytespenningsforløp. Fig. 4 viser en automatisk omkobling for tapseffektreduksjon ved iakttagelse av støyspenningen. "Fig. 2 shows the protection and limiting connection in the design according to the invention. Fig. 3 shows a graphical representation of the breaking voltage progression. Fig. 4 shows an automatic switching for loss effect reduction when observing the noise voltage.
Detaljert viser det kjente koblingsskjerna i henhold til fig. 1 en oversikt over lysintensitetsstyrekoblingen slik den er beskrevet i svensk patentsøknad nr. 8804209-8. I henhold til denne inneholder blokk 1 likeretteren GL, varistoren Vsfor transientpulsbegrensning samt fordrosselen Ls med mindre induktans som formotstand for varistoren Vsved hurtige transientpulser. Shows in detail the known coupling core according to fig. 1 an overview of the light intensity control coupling as described in Swedish patent application no. 8804209-8. According to this, block 1 contains the rectifier GL, the varistor Vs for transient pulse limitation and the pre-choke Ls with a smaller inductance as resistance for the varistor Vs for fast transient pulses.
Blokk 2 omslutter MOS-FET-transistoren Tl og den ikke tegnede beskyttelses- og begrensningskoblingen, samt koblingen for dannelse av bryteflanken. Block 2 encloses the MOS-FET transistor Tl and the protection and limiting circuit, not shown, as well as the circuit for forming the breaking edge.
Blokk 3 består av den allerede kjente matespenningsforsyningen til den monostabile vippe, av deteksjonen av netthalvbølgen og dermed triggingen av den monostabile vippe med netthalvbølgens nullgjennomgang samt selve den monostabile vippe hvis periode kan innstilles ved hjelp av et potensiometer. Block 3 consists of the already known feed voltage supply to the monostable flip-flop, of the detection of the mains half-wave and thus the triggering of the monostable flip-flop with the mains half-wave zero crossing as well as the monostable flip-flop itself whose period can be set using a potentiometer.
Et utførelseseksempel på koblingsanordningen i henhold til oppfinnelsen forklares nedenfor ved hjelp av fig. 2, idet elementene for flankesteilhet og den nødvendige støyfjerning er vist i detalj. I enkelte trekk bevirker dette gjennom elementene RI, Cl; R2, C2; samt ved MOS-FET-transistorens Tl Miller-kapasitans og i liten grad ved R3. An embodiment of the coupling device according to the invention is explained below with the help of fig. 2, as the elements for flank steepness and the necessary noise removal are shown in detail. In certain features, this works through the elements RI, Cl; R 2 , C 2 ; as well as by the MOS-FET transistor's Tl Miller capacitance and to a small extent by R3.
Om den monostabile vippe innkobles ved nullgjennomgangen så forer dette ikke til noen støyspenning. Ved bryting innen halvbølgen-utgang A spenningssprang fra H til L - skulle det uten de tidligere nevnte elementer inntreffe en høy gniststøy-spenning. If the monostable flip-flop is switched on at the zero crossing, this does not lead to any noise voltage. When breaking within the half-wave output A voltage jump from H to L - without the previously mentioned elements, a high spark noise voltage would occur.
Ved glødelamper skjer kollektorspenningsøkningen ved strømfor-andringen i M0S-FET-T1 etter proporsjonal økning av den store Miller-kapasitansen og den største stigning av denne spen-ningsøkning kommer til å skje kort før maksimalspenningen oppnås. Den tilkoblede kapasitans hos C2 forsinker reduksjonen av styrespenningen gjennom en motkobling som frembringes gjennom kollektorspenningen In the case of incandescent lamps, the collector voltage increase due to the current change in M0S-FET-T1 occurs after a proportional increase of the large Miller capacitance and the largest rise of this voltage increase will occur shortly before the maximum voltage is reached. The connected capacitance at C2 delays the reduction of the control voltage through a feedback produced by the collector voltage
dt date
Det dreier seg altså om en Miller-kobling med RI i serie med R2 som formotstand og C2 som Miller-kondensator. Ettersom man på grunn av tapseffekten til M0S-FET-T1 ikke kan gjøre bryteflanken vilkårlig flat, kreves et ytterligere koblingstiltak. Den indre Miller-kapasitans til M0S-FET-T1 bestemmer flanken ved begynnelsen av kollektorspenningens økning ved at Miller-kapasitansen til C2 sammen med RI og R2 bestemmer flankens hoveddel. Derved dimensjoneres C2 på en slik måte at det tas hensyn til den maksimalt tillatelige tapseffekt for Tl. Kondensatoren Cl påvirker i forbindelse med RI og sin utlad-ningskonstant ved T=3t styrespenningen ved de lave verdiene, dvs. kort før M0S-FET-T1 helt sperrer. Gis det avkall på kollektor-emitterspenningen, så fås det avhengig av koblingen et brytespenningsforløp som tilsvarer den grafiske fremstilling på fig. 3. It is therefore a Miller connection with RI in series with R2 as resistor and C2 as Miller capacitor. As, due to the loss effect of the M0S-FET-T1, it is not possible to make the breaking edge arbitrarily flat, an additional switching measure is required. The internal Miller capacitance of M0S-FET-T1 determines the edge at the beginning of the collector voltage rise, with the Miller capacitance of C2 together with RI and R2 determining the main part of the edge. Thereby, C2 is dimensioned in such a way that the maximum permissible loss effect for Tl is taken into account. The capacitor Cl, in connection with RI and its discharge constant at T=3t, affects the control voltage at the low values, i.e. shortly before M0S-FET-T1 completely blocks. If the collector-emitter voltage is waived, then depending on the connection, a breakdown voltage progression is obtained which corresponds to the graphic representation in fig. 3.
I detalj er de ulike forløpene av brytespenningen på fig. 3 vist for de ulike betingelsene. I den forbindelse viser kurven "a" brytespenningsforløpet uten et koblingstiltak, kurven "b" forløpet med kobling av elementene C2, RI og R2, mens "c" gjelder for elementene Cl, C2 samt RI og R2. In detail, the various progressions of the breaking voltage in fig. 3 shown for the various conditions. In this connection, curve "a" shows the breaking voltage sequence without a switching measure, curve "b" the sequence with switching of the elements C2, RI and R2, while "c" applies to the elements Cl, C2 as well as RI and R2.
I henhold til den S-kurve som dannes i henhold til fremstillin- gen "c", reduseres overtoner i stor utstrekning slik at det oppnås en forbedring på 49 dB. According to the S-curve formed according to the preparation "c", harmonics are greatly reduced so that an improvement of 49 dB is achieved.
Med minsking av strømmen ved brytingen blir ved glødelamper straks også spenningen og ballasten mindre, hvilket er 'ekvivalent med en økning av kollektor-emitterspenningen. Således skjer en tilbakekobling via Miller-kondensatorenC2 med en styrke som øker med økningen av kollektor-emitterspenningen. Det forhold som innstiller seg mellom kollektor-emitterspen-ningsforandring og styre-emitterspenningsreduksjon medfører en bestemt kollektorstrøm som etterhvert minker. Ved en koblings-nettdel forekommer store filterkapasitanser som bestemmer spenningen. Ved bryting av styrespenningen forandres kollektor-emitterspenningen bare langsomt og når langt fra sitt maksimum, mens kollektorstrømmen allerede helt er frakoblet. With the reduction of the current at the switching, in the case of incandescent lamps, the voltage and the ballast immediately become smaller, which is equivalent to an increase in the collector-emitter voltage. Thus, a feedback via the Miller capacitor C2 occurs with a strength that increases with the rise of the collector-emitter voltage. The relationship between collector-emitter voltage change and control-emitter voltage reduction results in a specific collector current which gradually decreases. In the case of a switching network part, large filter capacitances occur which determine the voltage. When the control voltage is broken, the collector-emitter voltage changes only slowly and reaches far from its maximum, while the collector current is already completely disconnected.
Den ubetydelige forandring av kollektor-emitterspenningen virker som om det ikke skulle forekomme noen Miller-kondensator i koblingen. Strømreduksjonen skjer vesentlig hurtigere enn ved glødelampelast. Om flanken ved koblingsnettdeldrift altså er valgt i samsvar med støyspenningsbetingelsene, så blir den med samme glødelampelast flatere og dermed reduseres således støyspenningen, men tapet i MOS-FET-Tl øker. The negligible change in the collector-emitter voltage appears as if there were no Miller capacitor in the junction. The current reduction takes place significantly faster than with an incandescent lamp load. If the flank in switching network partial operation is therefore chosen in accordance with the noise voltage conditions, then with the same incandescent lamp load it becomes flatter and thus the noise voltage is thus reduced, but the loss in the MOS-FET-Tl increases.
For å forhindre dette kan man manuelt redusere RI eller Cl for glødelampedrift, slik at støyspenningen fremdeles ligger innenfor tillatte grenser og tapseffekten i overenstemmelse med den ved drift med koblingsnettdeler. På fig. 4 er det vist en automatisk omkobling for tapsreduksjon under iakttagelse av støyspenningen. To prevent this, one can manually reduce RI or Cl for incandescent lamp operation, so that the noise voltage is still within permissible limits and the power loss in accordance with that for operation with switchgear. In fig. 4 shows an automatic switching for loss reduction while observing the noise voltage.
Når kollektorstrømmen ved koblingsnettdrift er blitt 0, så ligger den ved høyeste punkt av netthalvbølgen økende kollektor-emitterspenningen i brytetilfelle ved <_ 150 V. Om også transistoren T4 gjennomkobles ved >. 150 V ved sin basis-styring, så reduseres dermed tidskonstanten RI x Cl til When the collector current in switching mains operation has become 0, then at the highest point of the mains half-wave the increasing collector-emitter voltage in the breaking case is at <_ 150 V. If also the transistor T4 is switched through at >. 150 V at its base control, then the time constant RI x Cl is thus reduced to
RI x R13 xCi>dvs. at ved bryting i glødelampeområdet RI + R 13 akselereres strøm-spenningsforandringen når kollektor-emitterspenningen begynner å overstige spenningen 150 V. Dermed minskes tapseffekten. RI x R13 xCi>ie. that when breaking in the incandescent lamp area RI + R 13, the current-voltage change is accelerated when the collector-emitter voltage begins to exceed the voltage 150 V. Thus, the loss effect is reduced.
Ved brytingen i koblingsnettdelen forandres ingenting ved 'brytef orholdene så lenge strøm går igjennom MOS-FET-T1. When switching in the switching network part, nothing changes in the switching conditions as long as current flows through MOS-FET-T1.
Spenningens flanke under strømgjennomgangen samt strømmens karakteristikk er nøyaktig den samme som uten koblingsautoma-tikk. The edge of the voltage during the current flow as well as the characteristic of the current is exactly the same as without switching automation.
Ved omkoblingsanordningen i henhold til fig. 2 tjener den kjente R6-C3-anordningen til å dempe tilsvarende spenning-stopper, f.eks. ved induktiv last. For strømbegrensning ved gjennomkoblet MOS-FET-T1 i tilfelle kortslutning av lasten eller en utladning som følge av transienter, tjener elementene R4, R5, T2, T3, R7 samt dessuten den monostabile vippens reaksjoner. In the switching device according to fig. 2, the known R6-C3 device serves to dampen corresponding voltage stops, e.g. with inductive load. For current limitation at through-connected MOS-FET-T1 in the event of a short-circuit of the load or a discharge due to transients, the elements R4, R5, T2, T3, R7 as well as the monostable flip-flop's reactions serve.
På kjent måte innsettes en emitter-motstand (R4+R5) i den av transistorstrømmen gjennomløpte strekning for å muliggjøre utnyttelse av den gjennom strømmen på den samme frembragte spenningen som kriterium for et reguleringsforløp. I henhold til oppfinnelsen tilveiebringes denne motstand av R4 og R5, idet R4 er større enn R5. Ettersom hver motstand leverer styrespenningen for en transistors T2 og T3 basis-emitter-strekning oppnås ved økning av strømmen at den monostabile vippe tidsmessig tilbakestilles først via T3 - ved punkt "A" fra "H" til "L" - og først deretter inntreffer strømbegrens-ningen direkte på MOS-FET-T1 ved reduksjon av styrespenningen gjennom T2. Ettersom den over U-styreelektroden innregulerte strøm er høyere enn normal driftsstrøm og denne strøm dessuten direkte fås av høyden av styrespenningen til U-styreelektroden, så kommer Rdsonfor MOS-FET-T1 til å øke meget raskt og til tross for konstantholding av strømmen, kommer tapseffekten ved Tl til å stige. Ved motstanden R4 og dens i forhold til R5 høyere verdi sikres at i hvert tilfelle kobles den monostabile vippes U-styreelektrode etter kort tid mot 0, hvorved en tapsøkning forhindres. In a known manner, an emitter resistor (R4+R5) is inserted in the section traversed by the transistor current to enable utilization of the through current on the same produced voltage as a criterion for a regulation process. According to the invention, this resistance is provided by R4 and R5, R4 being greater than R5. As each resistor supplies the control voltage for a transistor's T2 and T3 base-emitter section, by increasing the current it is achieved that the monostable flip-flop is temporally reset first via T3 - at point "A" from "H" to "L" - and only then does current limiting occur -ning directly on MOS-FET-T1 by reducing the control voltage through T2. As the regulated current over the U control electrode is higher than normal operating current and this current is also directly obtained from the height of the control voltage to the U control electrode, the Rdson for MOS-FET-T1 will increase very quickly and despite keeping the current constant, the loss effect at Tl to rise. With the resistance R4 and its higher value compared to R5, it is ensured that in each case the U control electrode of the monostable flip-flop is connected to 0 after a short time, whereby an increase in loss is prevented.
Videre kreves ved sperret M0S-FET-T1 en spenningsbegrensning. For dette formål benyttes på kjent måte varistoren Vsi blokke 1 på fig. 1 og RC-organet R6 samt C3 i henhold til fig. 2. For hurtige pulser ved transienter med betydelig energiinnhold er 'disse tiltak imidlertid ikke tilstrekkelige. Kondensatoren benyttes i den forbindelse til å koble inn Tl. Zenerdioden Dl sorger for at U-styreelektroden ikke kan oke over zenerspen-ningen. Furthermore, when M0S-FET-T1 is blocked, a voltage limitation is required. For this purpose, the varistor Vsi block 1 in fig. is used in a known manner. 1 and the RC element R6 and C3 according to fig. 2. However, these measures are not sufficient for fast pulses at transients with significant energy content. The capacitor is used in this connection to connect Tl. The zener diode Dl ensures that the U control electrode cannot exceed the zener voltage.
Det i og for seg kjente koblingstiltak å eliminere overspen-ningen ved hjelp av strøminnkobleren på Tl tilveiebringes hva oppfinnelsen angår med C2 som forøvrig som tidligere nevnt, også allerede er nødvendig for å danne flankesteilheten. The per se known switching measure to eliminate the overvoltage by means of the current switch on T1 is provided as far as the invention is concerned with C2 which, incidentally, as previously mentioned, is also already necessary to form the flank steepness.
Claims (3)
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| DE3810743 | 1988-03-30 | ||
| DE3839373A DE3839373C2 (en) | 1988-03-30 | 1988-11-22 | Brightness control circuit for incandescent lamps and switching power supplies with a protection and limiting circuit to maintain an electronic fuse |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| NO891358D0 NO891358D0 (en) | 1989-03-30 |
| NO891358L true NO891358L (en) | 1989-10-02 |
Family
ID=25866531
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| NO89891358A NO891358L (en) | 1988-03-30 | 1989-03-30 | LIGHT INTENSITY CONTROL CLAMP FOR INCLUSIVE LAMPS AND CLUTCH NETWORKS WITH A PROTECTION AND LIMITING CLUTCH FOR AA PROVIDE AN ELECTRONIC SECURITY. |
Country Status (8)
| Country | Link |
|---|---|
| BE (1) | BE1002887A3 (en) |
| DK (1) | DK13489A (en) |
| FI (1) | FI891382A (en) |
| FR (1) | FR2629663A2 (en) |
| GB (1) | GB2217123A (en) |
| IT (1) | IT1228666B (en) |
| NL (1) | NL8900231A (en) |
| NO (1) | NO891358L (en) |
Families Citing this family (15)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4949020A (en) * | 1988-03-14 | 1990-08-14 | Warren Rufus W | Lighting control system |
| ATE109319T1 (en) * | 1990-04-17 | 1994-08-15 | Siemens Ag | DEVICE FOR THE CONTINUOUS CONTROL OF ELECTRICAL CONSUMERS ACCORDING TO THE PHASE CONTROL PRINCIPLE, IN PARTICULAR BRIGHTNESS REGULATORS AND USE OF SUCH A DEVICE. |
| EP0452716B1 (en) * | 1990-04-17 | 1994-07-27 | Siemens Aktiengesellschaft | Apparatus for continuous control of electric devices according to the phase chopping principle, especially light dimmer, and the use of such apparatus |
| US5239255A (en) * | 1991-02-20 | 1993-08-24 | Bayview Technology Group | Phase-controlled power modulation system |
| DE4201744C2 (en) * | 1992-01-23 | 1997-12-11 | Insta Elektro Gmbh & Co Kg | Additional circuit in a switching power supply for low-voltage halogen lamps |
| GB2278746A (en) * | 1993-06-03 | 1994-12-07 | Peter Levesley | A power controller for motors |
| AU757994B2 (en) * | 1999-03-25 | 2003-03-13 | H.P.M. Industries Pty Limited | Control circuit |
| AUPP944799A0 (en) * | 1999-03-25 | 1999-04-22 | H.P.M. Industries Pty Limited | Control circuit |
| US8564919B2 (en) | 2007-09-19 | 2013-10-22 | Clipsal Australia Pty Ltd | Dimmer circuit with overcurrent detection |
| FR2927479B1 (en) * | 2008-02-08 | 2012-05-04 | Schneider Electric Ind Sas | SYSTEM FOR CONTROLLING AND PROTECTING A NEGATIVE LOGIC OUTPUT OF AN AUTOMATION EQUIPMENT. |
| GB2493562B (en) * | 2011-08-12 | 2018-10-17 | E2V Tech Uk Limited | Drive circuit and method for a gated semiconductor switching device |
| GB2511571A (en) * | 2013-03-08 | 2014-09-10 | Zano Controls Ltd | Dimmer switches suitable for LED lamps |
| JP6653452B2 (en) * | 2016-09-20 | 2020-02-26 | パナソニックＩｐマネジメント株式会社 | Protection circuit for dimmer and dimmer |
| EP3570439A1 (en) * | 2018-05-16 | 2019-11-20 | Siemens Aktiengesellschaft | Limiting an electrical current in a power semiconductor switch |
| CN113847973B (en) * | 2021-10-22 | 2023-12-26 | 重庆智慧水务有限公司 | Water meter working current testing tool and testing method |
Family Cites Families (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4540893A (en) * | 1983-05-31 | 1985-09-10 | General Electric Company | Controlled switching of non-regenerative power semiconductors |
| US4528494A (en) * | 1983-09-06 | 1985-07-09 | General Electric Company | Reverse-phase-control power switching circuit and method |
| DE3445340A1 (en) * | 1984-12-12 | 1986-06-19 | Staiber, Heinrich, 8201 Bad Feilnbach | MOSFET two-way switch with current limiting |
| NL8601761A (en) * | 1986-07-07 | 1988-02-01 | Tjeerd Venema | ELECTRONICALLY CONTROLLED ELECTRIC CURRENT FUSE. |
-
1989
- 1989-01-12 DK DK013489A patent/DK13489A/en not_active Application Discontinuation
- 1989-01-31 NL NL8900231A patent/NL8900231A/en not_active Application Discontinuation
- 1989-03-06 BE BE8900239A patent/BE1002887A3/en not_active IP Right Cessation
- 1989-03-13 FR FR8903247A patent/FR2629663A2/en active Pending
- 1989-03-20 IT IT8919810A patent/IT1228666B/en active
- 1989-03-22 FI FI891382A patent/FI891382A/en not_active Application Discontinuation
- 1989-03-30 GB GB8907177A patent/GB2217123A/en not_active Withdrawn
- 1989-03-30 NO NO89891358A patent/NO891358L/en unknown
Also Published As
| Publication number | Publication date |
|---|---|
| BE1002887A3 (en) | 1991-07-16 |
| DK13489D0 (en) | 1989-01-12 |
| FR2629663A2 (en) | 1989-10-06 |
| FI891382A (en) | 1989-10-01 |
| NO891358D0 (en) | 1989-03-30 |
| FI891382A0 (en) | 1989-03-22 |
| GB8907177D0 (en) | 1989-05-10 |
| IT8919810A0 (en) | 1989-03-20 |
| NL8900231A (en) | 1989-10-16 |
| IT1228666B (en) | 1991-07-03 |
| GB2217123A (en) | 1989-10-18 |
| DK13489A (en) | 1989-10-01 |
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