NL2014897B1 - Method for determining or estimating an impedance of at least one electrode. - Google Patents
Method for determining or estimating an impedance of at least one electrode. Download PDFInfo
- Publication number
- NL2014897B1 NL2014897B1 NL2014897A NL2014897A NL2014897B1 NL 2014897 B1 NL2014897 B1 NL 2014897B1 NL 2014897 A NL2014897 A NL 2014897A NL 2014897 A NL2014897 A NL 2014897A NL 2014897 B1 NL2014897 B1 NL 2014897B1
- Authority
- NL
- Netherlands
- Prior art keywords
- signal
- input
- amplifier
- signals
- impedance
- Prior art date
Links
Classifications
-
- A—HUMAN NECESSITIES
- A61—MEDICAL OR VETERINARY SCIENCE; HYGIENE
- A61B—DIAGNOSIS; SURGERY; IDENTIFICATION
- A61B5/00—Measuring for diagnostic purposes; Identification of persons
- A61B5/72—Signal processing specially adapted for physiological signals or for diagnostic purposes
- A61B5/7225—Details of analog processing, e.g. isolation amplifier, gain or sensitivity adjustment, filtering, baseline or drift compensation
-
- A—HUMAN NECESSITIES
- A61—MEDICAL OR VETERINARY SCIENCE; HYGIENE
- A61B—DIAGNOSIS; SURGERY; IDENTIFICATION
- A61B5/00—Measuring for diagnostic purposes; Identification of persons
- A61B5/24—Detecting, measuring or recording bioelectric or biomagnetic signals of the body or parts thereof
- A61B5/316—Modalities, i.e. specific diagnostic methods
- A61B5/369—Electroencephalography [EEG]
Landscapes
- Health & Medical Sciences (AREA)
- Life Sciences & Earth Sciences (AREA)
- Engineering & Computer Science (AREA)
- Medical Informatics (AREA)
- Surgery (AREA)
- Biophysics (AREA)
- Pathology (AREA)
- Signal Processing (AREA)
- Biomedical Technology (AREA)
- Heart & Thoracic Surgery (AREA)
- Psychiatry (AREA)
- Molecular Biology (AREA)
- Physics & Mathematics (AREA)
- Animal Behavior & Ethology (AREA)
- General Health & Medical Sciences (AREA)
- Public Health (AREA)
- Veterinary Medicine (AREA)
- Power Engineering (AREA)
- Psychology (AREA)
- Artificial Intelligence (AREA)
- Computer Vision & Pattern Recognition (AREA)
- Physiology (AREA)
- Measurement And Recording Of Electrical Phenomena And Electrical Characteristics Of The Living Body (AREA)
Abstract
Method for determining or estimating an impedance of at least one electrode, the method being characterized by utilization of: a reference amplifier apparatus for processing physiological measuring signals, in particular EEG-signals (electroencephalogram signals), wherein the apparatus is provided with a plurality of channels (1) with signal inputs for receiving input signals, which input signals each comprise a specific signal component and a signal component common to all input signals, wherein each channel is provided with an impedance transforming input amplifier (3), wherein the apparatus is configured for supplying to a first input of each input amplifier a respective signal and to a second input an analogue reference signal which is equal for all channels, wherein the apparatus is provided with a digital signal processor (210) and one or more analogue digital converters (5) for supplying the signals provided by the input amplifiers (3) to the digital signal processor (210), wherein the signal processor (210) is designed for converting signals received from the one or more analogue digital converters (5) at least into one or more output signals, wherein the signal processor (210) is designed for processing the signals received from the one or more analogue-digital converters (5), for providing a digital reference signal, wherein the apparatus is provided with a digital-analogue converter (111) for converting the digital reference signal into said analogue reference signal. The invention also provides a system including a number of electrodes and a reference amplifier apparatus.
Description
Title: Method for determining or estimating an impedance of at least one electrode
The invention relates to a method for determining or estimating an impedance of at least one electrode. Also, the invention relates to a corresponding system.
Applicant’s European patent application No. 08793828.8 discloses a reference amplifier apparatus. The apparatus is configured for processing physiological measuring signals, in particular EEG-signals (electroencephalogram signals). To that aim the wherein the apparatus is provided with a plurality of channels with signal inputs for receiving input signals from electrodes during use. Such input signals usually comprise a specific signal component and a signal component common to all input signals. Also, each channel is provided with an impedance transforming input amplifier, wherein the apparatus is configured for supplying to a first input of each input amplifier a respective signal and to a second input an analogue reference signal which is equal for all channels.
The known apparatus is provided with a digital signal processor and one or more analogue digital converters for supplying the signals provided by the input amplifiers to the digital signal processor. Advantageously, the signal processor is designed for converting signals received from the one or more analogue digital converters at least into one or more output signals, and also for processing the signals received from the one or more analogue-digital converters, for providing a digital reference signal. The apparatus is provided with a digital-analogue converter for converting the digital reference signal into said analogue reference signal.
As follows from EP08793828.8, in this way a DC reference amplifier can be obtained, with a relatively high dynamic range. Further, in this manner, use of an analogue high pass filters can be omitted. As a result, undesired effects of artefacts in input signals, for instance saturation of an amplifier part of the apparatus for a particular period of time, can be suitably avoided.
For a lot of electro-physiological measurements the measurement of the impedance of the measurement electrode is prescribed. In the past this was technically necessary, because the input impedance of earlier amplifier types was not very high. In such cases the amplitude of the measured signal was dependent on the impedance of the input amplifier and the impedance of the electrode. Presently, the amplifiers do have an input impedance that is that high that the measurement of the impedance is not so important anymore. This holds for example for the known reference amplifier apparatus.
Still, presently, national guidelines may prescribe that the impedance of the measurement electrode is determined, especially for the measurement of EEG.
In the known method, to measure the impedance of electrodes, an electronic circuit is placed before the input amplifier which is connected to the several electrodes using electronic switches. The electronic circuit consists of a sine wave (or any other wave) oscillator that introduces a current through the electrodes. Because the current is known, the impedance can be calculated. During measurement of EEG this circuit normally is switched of, although there are impedance solutions using a higher frequency that is filtered out of the EEG signal. This gives the possibility to measure the impedance continuously.
The disadvantage of the electronic circuit in front of the input amplifier is that the input impedance of the measurement circuit has been decreased in a significant extend. And this is an important decrease of the properties of the input of the measurement system. Also, using the electronic circuit is rather inefficient and cumbersome.
The present invention aims to solve or alleviate the above-mentioned problems. Particularly, the invention aims to provide an improved, more efficient method to determine or at least estimate the impedance of at least one electrode.
To this aim, the invention provides a method that is characterized by the features of claim 1.
According to an aspect of the invention there is provided a method for determining or estimating an impedance of at least one electrode, the method being characterized by utilization of: a reference amplifier apparatus for processing physiological measuring signals, in particular EEG-signals (electroencephalogram signals), wherein the apparatus is provided with a plurality of channels with signal inputs for receiving input signals, which input signals each comprise a specific signal component and a signal component common to all input signals, wherein each channel is provided with an impedance transforming input amplifier, wherein the apparatus is configured for supplying to a first input of each input amplifier a respective signal and to a second input an analogue reference signal which is equal for all channels, wherein the apparatus is provided with a digital signal processor and one or more analogue digital converters for supplying the signals provided by the input amplifiers to the digital signal processor, wherein the signal processor is designed for converting signals received from the one or more analogue digital converters at least into one or more output signals, wherein the signal processor is designed for processing the signals received from the one or more analogue-digital converters, for providing a digital reference signal, wherein the apparatus is provided with a digital-analogue converter for converting the digital reference signal into said analogue reference signal, wherein a number of electrodes are electrically connected to respective first inputs of said input amplifiers, wherein a leakage current is supplied via the electrodes to the reference amplifier, wherein a resulting signal is being processed by said signal processor for determining or estimating the impedance of at least one of the number of electrodes.
In this way said impedance can be determined or estimated in an efficient manner, and particularly without having to connect a dedicated measuring circuit in front of the amplifier. A basic idea behind the present invention is to use the reference amplifier itself as an electrode impedance determiner. It has been found that the signals that reach the signal processor of the reference amplifier provide sufficient information to obtain at least a good estimate of electrodes that are connected to the inputs of the amplifier. Particularly, the reference amplifier apparatus can estimate, determine or calculate impedances of a relatively large number of electrodes connected there-to, making use of leakage currents running via the electrodes to the amplifier.
According to a further advantageous embodiment, the signal processor can be configured to determine a common mode signal, and to use the determined common mode signal to determine or estimate the impedance of at least one of the number of electrodes.
In addition, according to a further embodiment, the signal processor calculate an amplitude of a leakage current signal present in the channel of the electrode of which the impedance is to be estimated or determined.
Also, it is preferred that the signal processor uses at least one predetermined input impedance of the reference amplifier apparatus to calculate or estimate the impedance of at least one of the number of electrodes, said at least one predetermined input impedance for example being stored in a memory of the signal processor.
Generally, during operation of the amplifier all the electrode signals are amplified against a mean of all the connected electrodes. And the mean of all the connected electrodes is per definition the common mode of the connected signals. Normally, this common mode is removed in a second stage of the amplifier, but the present reference amplifier can keep the common mode digitized in the data. The reference amplifier can be configured such that the common mode signal is removed from the date during an operational mode of actual measurement of signals electroencephalogram signals, so that those signals can be output (in processed form) by the amplifier.
According to an embodiment of the present invention, during a electrode impedance measuring mode of the reference amplifier, it is not necessary to remove the common mode signal in the amplifier hardware, wherein that signal is used in the determination/estimation of a said impedance. However, according to a preferred embodiment, during a electrode impedance measuring mode of the reference amplifier, the common mode signal is determined and removed in the amplifier hardware, so that that signal can be used in the determination/estimation of a said impedance and additionally accurate measurements of physiological signals can be carried out.
Particularly, in an embodiment, the common mode can consist of the mean of the offsets of the electrodes, the mean of any bio-electrical signals and e.g. a 50 or 60 Hz signal (relating to nearby mains interference) that is introduced by the leakage current though a patient ground electrode.
Normally the leakage current (e.g. a 50 or 60 Hz signal) has an amplitude in the range of millivolts. In very “clean” environments the amplitude of the mains interference signal does not exceed lOOOuV. But in some conditions the amplitude can increase up to 200.000uVolts.
The mains interference signal is not always stable, especially in situations where the measured person is moving. In that situation, as will be explained below, an artificial leakage current can be used, supplied to the patient ground electrode. Preferably such an artificial leakage current has a frequency hat differs from the 50/60 Hz interference, for instance 35 Hz. This leakage current builds an artificial common mode signal.
In normal situations, which means that the input impedance of the amplifier channels are much higher than the impedances of the measurement electrodes, the common mode signal on the inputs of the amplifiers is exactly the same for all channels. In figure 3, an example is given of the conversion of the leakage common mode signal to the input of the amplifier. The following formula applies for the impedance of the electrode Zel:
(1) wherein Vin is the detected signal (Volt) at the input of the reference amplifier to which the electrode is connected, Zinp is a predetermined imput impedance of the reference amplifier, and Vc is the common mode signal (Volt) determined by the reference amplifier (particularly by said signal processor). If the average of the Vout signals as illustrated in fig 1 the result will be the common mode signal.
It follows that to have an indication of the electrode impedance, the amplitude value of the leakage current signal in every channel can be calculated by the reference amplifier (particularly by said signal processor). This gives an indication of the variation of the electrode impedances per channel.
According to a particular embodiment, the common mode signal represents the mean of the electrode impedances. The differences of the leakage current (e.g. 50/60Hz) amplitudes in the several channel signals indicate the variation of the electrode impedance with respect to that mean.
According to a further embodiment, the number of electrodes are placed in electrical contact with a conducting body to receive the leakage current therefrom, for example a body of a living species, for example a patient.
Particularly, at least part of the leakage current can be induced by a leakage current generator, generating an alternating leakage current signal, for example a sinus wave signal.
According to a further elaboration, the leakage current generator can include a signal source that is connected to a first input of a second impedance transforming amplifier, wherein an output of the second impedance transforming amplifier is electrically connected to a first position of the conducting body wherein a second input of the second impedance transforming amplifier is electrically connected to a second position of the conducting body.
Advantageously, the leakage current can be an alternating current, preferably having a constant frequency, for example a frequency lower than 100 Hz, and preferably having an amplitude lower than 10 mV, for example lower than 5 mV, for example about 1 mV or 2 mV. In this way good results can be achieved.
As follows from the above, in particular, a plurality (i.e. at least two) electrodes are connected to two respective inputs of the reference amplifier, wherein leakage current runs through all electrodes into the reference amplifier, leading to a respective common mode signal
As follows from the above, at least part the leakage current can be induced my mains interference. The leakage current may e.g. be entirely induced my mains interference, so that no additional leakage current generator has to be installed.
To obtain precise results, preferably a dedicated leakage current generator, not being a mains interference inducer, is provide to provide the entire leakage current to the electrodes. In that case it is preferred that the alternating signal has a frequency that differs from a frequency of local mains electricity, the frequency of the alternating signal for example being in the range of 1 to 100 Hz.
According to a further embodiment, each channel is provided with a compensation amplifier of which a first input is arranged for receiving the signal of a respective input amplifier, and of which an output is directly or indirectly coupled to a said analogue-digital converter, wherein a second input of the compensation amplifier is coupled via a digital-analogue converter to said signal processor for receiving a control signal therefrom. For example the signal processor can be designed for detecting a signal offset voltage for each channel (at least, a channel-specific offset, for instance a electrode offset), and for adapting the control signal such that the respective compensation amplifier can remove at least a part of the signal offset-voltage from the signal under the influence of this control signal -supplied via a D/A converter to this amplifier. Thus, in particular, physiological measurements can be made with high accuracy by the amplifier.
According to an embodiment, the signal processor can digitally average the signals received from the one or more analogue-digital converters.
Further, an aspect of the invention provides a system that is characterized by the features of claim 13.
The innovative system includes a number of electrodes and a reference amplifier apparatus, the apparatus being configured for processing physiological measuring signals, in particular EEG-signals (electroencephalogram signals), wherein the apparatus is provided with a plurality of channels with signal inputs for receiving input signals from said electrodes, which input signals each comprise a specific signal component and a signal component common to all input signals, wherein each channel is provided with an impedance transforming input amplifier (3), wherein the apparatus is configured for supplying to a first input of each input amplifier a respective signal and to a second input an analogue reference signal which is equal for all channels, wherein the apparatus is provided with a digital signal processor and one or more analogue digital converters for supplying the signals provided by the input amplifiers to the digital signal processor, wherein the signal processor is designed for converting signals received from the one or more analogue digital converters at least into one or more output signals, wherein the signal processor is designed for processing the signals received from the one or more analogue-digital converters, for providing a digital reference signal, wherein the apparatus is provided with a digital-analogue converter for converting the digital reference signal into said analogue reference signal, wherein a said signal processor is also configured for processing signals received from the one or more analogue digital converters for determining or estimating an impedance of at least one of the number of electrodes.
In this way the above-mentioned advantages can be achieved.
For example the system can be configured to operate in a first mode, a electrode impedance testing mode, wherein impedances of the electrodes are determined or estimated. Also, the system can be configured to operate in a second mode, a physiological signal measuring mode, wherein physiological signals are being measured (and e.g. output via amplifier output means). Besides, a said first mode and second mode can partly of fully overlap, wherein electrode impedances are determined simultaneously with the measuring of physiological signals.
The invention will presently be elucidated in further detail on the basis of the Figures, in which:
Fig. 1 shows an embodiment of a reference amplifier;
Fig.2 schematically shows an embodiment of the invention;
Fig. 3 shows an electric diagram relating to the embodiment shown in
Fig. 2; and
Fig. 4 schematically shows a further embodiment of the invention.
In this application, identical or corresponding features are indicated with identical or corresponding reference numerals.
As described hereinabove, Fig. 1 shows an example of a known reference amplifier. In the following, various apparatuses are described, which offer various advantages over the system represented in Fig. 1.
Particularly, the (DC) reference amplifier RA serves for processing signals, in particular physiological measuring signals supplied through sensors (not represented), is provided with several (N) channels with N signal inputs 1 for receiving input signals, which input signals El, E2, E3, ..., En (with n being 1, 2, 3, ..., N) each comprise a specific signal component and a signal component common to all input signals. Furthermore, for a particular period of time, one or more input signals El, E2, E3, ..., En can contain sensor related offset signal parts, as is described hereinabove.
Preferably, the different channels (also called signal paths) can be designed in the same manner. Each channel is preferably provided with an impedance transforming input amplifier 3. These amplifiers 3 are preferably operational amplifiers (Opamps). The present apparatus is configured for supplying to a first input (in the example being the non-inverting input) of each input amplifier 3 a respective input signal El, E2, E3, ..., En and, to the other (in this case inverting) input, an analogue reference signal Vref common to all channels, for providing an associated amplifier output signal via an output of the input amplifier 3.
To this end, the apparatus RA is provided with an averager 6 which is designed for forming the reference signal.
In particular, the second input of each input amplifier is communicatively coupled, by means of a first respective impedance Ril, R21, R31, ..., Rnl to an output of the average 6, for receiving the reference signal Vref. Furthermore, by means of a second respective impedance R12, R22, R32, ..., Rn2, the second input of each input amplifier 3 is brought into signal connection with the output of this amplifier. In particular, the first impedances Ril, R21, R31, ..., Rnl, are resistances wherein, according to a relatively simple design, all resistances Ril, R21, R31, ..., Rn are preferably identical. The same holds for the second impedances R12, R22, R32, ..., Rn2. As will be clear to the skilled person, the gain supplied by each input amplifier 3 is set by the ratio between the respective first and second resistance.
In an advantageous manner, the apparatus RA is provided with a digital signal processor 210, and several analogue-digital converters (A/D converters) 5, i.e. one per channel, for supplying the amplified signals provided by the input amplifiers 3, to the digital signal processor 210.
The signal processor 210 is designed for converting signals received from the one or more analogue-digital converters 5 into at least one or more output signals, and outputting this/these, for instance, via a signal output 219. The signal output 219 can be designed in different manners, and for instance comprise a digital output, in particular a serial output, and can for instance be suitable for supplying output signals relating to a large number of channels (for instance 128 channels, or another number). An output signal supplied by signal processor 210 can for instance comprise a digital signal stream, which is provided with the digital signals coming from the different channels, for instance in a predetermined order.
Alternatively, the signal processor 210 may be designed for parallel outputting of the digital signals coming from the different channels via a parallel signal output (not represented).
The exemplary embodiment RA according to Fig. 1 is further provided with low pass filters 7. In particular, each channel is provided with such a low pass filter 7, which allows for instance passage of signal parts LD with frequencies from 0 Hz up to a particular cut-off frequency, and substantially prevents passage of signal parts with frequencies higher than the cut-off frequency. Such cut-off frequency can in particular be such that the generally known Nyquist criterion is met with respect to the frequencies of the signals to be detected, which will be clear to the skilled person.
An input of each filter 7 is for instance directly or indirectly coupled to the output of a respective input amplifier 3, of the same channel, for receiving the signal coming from this amplifier. The output of each filter 7 is each time in particular upstream with respect to a respective analogue-digital converter 5. The different channels of the apparatus themselves are not provided with high pass filters, this contrary to the apparatus represented in Fig. 1. In this manner, the adverse effects of a sustained saturation resulting from the time constant of such a high pass filter are avoided.
With the exemplary embodiment of Fig. 1, the averager 6 is communicatively connected to outputs of the input amplifiers 3, via signal connections which are located signal-upstream with respect to the low pass filters 7. The analogue averager 6 is provided with an impedance transforming amplifier (preferably an Opamp), of which an output is coupled in particular to the second inputs of the input amplifiers 3, via said first impedances Rll, ..., Rnl for supplying said reference signal Vref to the input amplifiers 3.
In one embodiment (not shown), the averager may be is provided in a simple manner with a series of parallel arranged third impedances, in particular third resistances rl, r2, r3, ..., m which are preferably identical to each other, and which couple the outputs of the input amplifiers 3 to the non-inverting input of the averaging amplifier 6a. Iln that case t is preferred that the inverting input and output of the amplifier of the averager are directly coupled, via a signal connection. In other words: the amplifier can be is provided in an input-follower configuration, and thus provides for a gain lx with a high input impedance and low output impedance, which configuration is generally known per se.
Preferably, each of the A/D converts 5 has a high dynamic range. To this end, use can for instance be made of A/D converters of a resolution of 20 bits or higher, for instance a resolution of 21 or 22 bits, or a different resolution. Example: suppose that an analogue-digital range is approximately ± 3 Volts and the LSB (least significant bit) resolution is 1 microvolt, the number of bits of the A/D converter 5 should then be at least 22 bits (21 bits is sufficient for 3 Volt; therefore, for +3 and -3 Volt, the resolution should be 22 bits, the ‘sign bit’ included), which will be clear to the skilled person. If the input stage of the apparatus (comprising the input amplifiers 3) is for instance configured for supplying a gain of 20x, an A/D converter resolution of 0.07 microvolts LSB follows.
The signal processor 210 can comprise, for instance, a digital signal processor (DSP), and can be provided by hardware, software or a combination thereof. The signal processor 210 itself can be designed for processing digital signals received from the A/D converters 5 such that certain sensor related offsets in those signals are substantially removed from the signals. To this end, the signal processor 210 can for instance be designed for detecting such offsets. Here, processing of the data is preferably such that a detection and modification of a digital signal part coming from one of the channels mentioned has no influence on the digital signal parts coming from the other channels.
Optionally, the signal processor 210 can be designed for carrying out a high pass filtering of each of the digital signals received from the A/D converters while utilizing, for instance, one or more suitable high pass filters designed in software.
In an advantageous embodiment, the signal processor 210 is only designed to receive the digital data streams received from the A/D converts 5 and convert these into one digital output signal stream, wherein the signal processor 210 utilizes no digital signal offset compensation and carries out high pass filtering. The output signal stream can for instance be processed further by a data processor (for instance computer, not represented) and/or can be saved. Such a data processor can for instance comprise suitable hardware and/or software, which is designed for detecting signal offsets in the output signal steam, per channel, and for removing the detected offsets for each channel. The data processor (not represented) can for instance be designed for carrying out a high pass filtering of the digital channel parts received from the signal processor 210.
In the present exemplary embodiment, it is further advantageous when a data processor is designed for digitally detecting the signal component common to all input signals. Then, the data processor can process the received signals for removing a detected common signal component from the signals. Thus, with additional advantage, the data processor can take over the function of analogue signal compensation means, per channel.
The advantage of the apparatus represented in Fig. 1 is that amplification of input signals needs only be supplied by one amplifier stage (comprising the reference input amplifiers 3). Furthermore, the complete amplifier stage is DC-coupled. Subsequently filtering and compensating the digitized signals can simply be carried out by suitable digital hardware and/or software (preferably by a data processor, not represented).
Furthermore, as is depicted in Fig. 1, preferably the signal processor 210 is designed for supplying a digital reference signal, which digital reference signal is converted into said analogue reference signal Vref by suitable converter means 111, 6. In that case the apparatus according to Fig. 3 not provided with an analogue averaging system, but with a digital signal averager which forms part of the signal processor 210.
In Fig. 1, in particular, the digital signal processor 210 is designed to process, in particular digitally average, the digital signals received from the A/D converters 5, for providing the digital reference signal. Said averaging is schematically represented in Fig. 3 and comprises, for instance, a summation step Σ wherein the digital signals, coming from the N different channels, are added up, and a division step 1/N wherein the sum of the digital signals (i.e. the result of said summation step) is divided by the number N of those signals (i.e. the number of channels). The digital result of this operation comprises a digital reference signal, which is supplied via a signal output to the digital-analogue (D/A) converter (DAC) 111 (for instance a 20 bit D/A converter 111). This D/A converter 111 converts the digital reference signal into an analogue signal, which analogue signal is processed via an impedance transforming amplifier 6 for providing the analogue reference signal Vref. To this end, the output of the D/A converter 111 is coupled via a suitable impedance (in particular comprising a resistance m) to the non-inverting input of this amplifier 6, while the inverting amplifier input is electrically short-circuited to the amplifier output. In this case, the non-inverting amplifier input is preferably earthed via a condenser.
Preferably, the signal processing is integrally provided with digital high pass filters, for instance designed in software, for filtering signal components below a particular cut-off frequency from the signals. It will be clear to the skilled person how such a digital high pass filter, as such, can be designed. Preferably, the digital high pass filtering is not carried out before the summation step Σ is carried out, so that, also, an average offset signal becomes part of the reference signal, and the amplification of the input opamps 3 can be based on the variance of the offset signals. Here, a cut-off frequency of each digital high pass filter can for instance be 1Hz or less, in particular 0.1Hz or less, more particularly 0.01 Hz or less, in particular for substantially removing DC components from the digital signals. A great advantage of the apparatus represented in Fig. 1 is that it comprises a digital DC reference amplifier, provided with digital components for calculating the reference signal. The calculation (comprising, for instance, the summation step Σ and division step 1/N) of the digital reference voltage can for instance be carried out by suitable software of the signal processor 210. A further advantage is that this calculation is carried out digitally and therefore no longer depends on analogue components (such as the aforementioned various resistances rl, r2, ..., m). In a further elaboration, the signal processor 210, such as in the exemplary embodiment represented in Fig. 2, can process the signals such that specific sensor related offsets are removed from the signals, per channel.
Furthermore, as shown in Fig. 1 it is preferred that a compensated digital DC reference amplifier configuration is used. In that further elaboration a second amplifier stage is provided, comprising compensation amplifiers (in particular Opamps) 212 which are each arranged in a respective channel between a low pass filter 7 and A/D converter 5. In this example, a first (inverting) input of each compensation amplifier 212 is arranged for receiving the low pass filtered amplified signal from a respective input amplifier 3, via a filter 7. In the present exemplary embodiment, an output of each compensation amplifier is directly coupled to an A/D converter 5 for providing a signal compensated by the amplifier 212 to the A/D converter 5, wherein the A/D converter supplies the digitized signal to a respective input (In) of the signal processor 210. Alternatively, a low pass filter 7 is arranged in a channel between the compensation amplifier 212 and respective A/D converter 5 (in that case, the input of this filter 7 is still located signal-downstream with respect to the respective input amplifier 3).
In an advantageous manner, the second (non-inverting) input of each compensation amplifier 212 is coupled via a suitable digital-analogue converter 213 to a respective output (Out) of the signal processor 210 for receiving a control signal therefrom. In this case, the signal processor 210 comprises, for instance, a microcontroller, with suitable software for carrying out various signal processing functions of the signal processor 210. These signal processing functions comprise, in particular: calculating a digital reference signal, producing suitable control signals to be supplied to the D/A converters 213 and providing output signals, via the (preferably serial) output 219.
The signal processor 210 can further be provided with, for instance, one or more other inputs, for instance a serial input 221 for communication with other modules or components. According to a further elaboration, the input 221 can communicate with, for instance, other modules, so that improved modularity and synchronization with respect to other signal systems, and the flexibility with regard to extension of the number and types of channels can be obtained.
According to a further elaboration, the signal processor 210 is designed for providing each control signal under the influence of and/or while utilizing one or more digital signals obtained from the analogue digital converters 5.
Preferably, the signal processor 210 is designed for detecting a signal offset voltage for each channel (at least, a channel-specific offset, for instance a electrode offset), and for adapting the control signal such that the respective compensation amplifier 212 can remove at least a part of the signal offset-voltage from the signal under the influence of this control signal - supplied via the D/A converter 213 to this amplifier 212.
In particular, from each channel an average offset potential and amplified offset variance is then compensated so that the respective compensation amplifier 212 only amplifies the physiological component and a part of the common component, this being the common part not belonging to the low frequent offset. Even the use of different electrode materials is thus an option.
If the signal processor 21 detects, for instance, that in the second channel (with respect to the second input signal E2), during a particular measuring period, a particular low frequent offset voltage variation is present, the signal processor 210 can control the D/A converter via the respective digital output (out(2)’) for supplying the same voltage variation to the inverting input of the respective compensation amplifier 212. Thus, per channel, the offset variance can be rapidly and accurately compensated.
During use, in case of measuring physiological signals, the average offset and the offset variance are preferably completely compensated. In this manner, the average of the offset still present 1 x amplified in the signal, and the offset variance can be completely compensated. The signal processor 210 needs not distinguish the offset variance. This very low frequent and fairly high amplitude of the offset signal is for instance digitally known, in processor 210, and can be removed from the signal by means of compensation amplifier 212. As a result, a substantially physiological signal remains. This signal is additionally amplified and digitized. In the signal processor 210, the compensated offset and the digitized physiological component can be joined into one digital signal, if this is required. Owing to the additional amplification, for instance A/D converters 5 with relatively low resolutions can be used.
According to a further elaboration, the signal processor 210 can be arranged to take the specific design of the compensation amplification stage 212 into account for producing an accurate signal offset compensation per channel. The signal processor 210 may be provided with calibration data with respect to linearity and gain of the compensation amplifiers 212, which calibration data are for instance used by the signal processor 210 when determining the control signals. Such calibration data can for instance be saved in a memory (not specifically represented) of the signal processor 210, and for instance be pre-entered or be determined by the signal processor 210 via a suitable calibration step. With such a calibration step, known input signals may be supplied to the inputs 1 of the apparatus. An optional compensation stage calibration can also comprise a different calibration method, which will be clear to the skilled person.
In the exemplary embodiment RA of Fig. 1 too, it is advantageous when the signal processor 210 is designed to carry out a high pass filtering of each of the digital signals received from the A/D converters 5.
The elaboration according to Fig. 1 has several advantages. For instance, a relatively high gain can be used, for instance 500x (for instance lOx in the input stage 3 and 50 x in the compensation stage 212, or a different, suitable proportion), so that A/D converters 5 with a relatively low resolution, low disturbance, high sample frequency and virtually no delay can be used. In this manner (for instance physiological) measuring signals can be distinguished from other signal parts particularly well and be detected. In particular, offset per channel can be compensated well, so that the amplification per channel no longer depends on the offset variances, but only on the electrophysiological signal characteristics (i.e. the specific signal components mentioned). Furthermore, the capacity to remove common signal components from the signals can thus be improved considerably, while power consumption can decrease.
According to a further embodiment, described as such in WO2009/017413 which is incorporated by reference in the present application in its entirety, it is advantageous when the apparatus is provided with a multiplexer and only one analogue-digital converter, wherein the arrangement is set up such, at least coupled to each other such that the multiplexer supplies the amplified signals coming from the various channels in a predetermined order to the analogue digital converter . The A/D converter supplies the successively digitized signals to the signal processor. In this case, the compensation amplifiers are therefore indirectly coupled to the A/D converter.
The reference amplifier can offer various advantages, as has already been described hereinabove. Different signals can be processed via different channels, for amplifying specific signal components, for the purpose of detection, and, in particular, for distinguishing the specific components of common signal components (separation of specific signal components and common components can be carried out by the apparatus itself, or, for instance, by digital signal processing means arranged downstream). Here, in particular, a relatively high gain can be utilized on the input signals by an amplifier stage 3 located signal-upstream, for instance a gain of 50x or more, in particular lOOx or more, or another value.
Here, in particular a digital circuit, at least a digital signal processor 210, is integrated into the apparatus. The apparatus is preferably designed such that DC components typically present in the input signals (for instance DC offset parts) can reach the digital signal processor via respective channels, wherein the channels are preferably not provided with DC filtering means. A common signal component can for instance be digitally calculated from digital information. Preferably, a feedback loop 111, 6 is available for feeding a digitally calculated common signal component as reference signal back to the reference amplifier part 3 of the apparatus.
More preferably, furthermore, a specific offset per channel is calculated by the digital signal processor (for instance by suitable software of the digital signal processor) while utilizing the signals digitized by the A/D converter(s) 5, 305, such as, for instance, in the elaboration according to Figs. 4 or Fig. 5. The thus calculated specific offset can then simply be compensated, per channel, via said compensation amplifier stage 212.
In any case, removing a signal component (in particular a mains component, for instance a 50Hz or 60Hz AC component) common to all input signals from the analogue input signals is preferably completely controlled by/via the digital signal processor 210, while utilizing the signals digitized by the A/D converter(s) 5. This is, however, not essential, see the elaboration according to Fig. 2.
In the above, the reference amplifier RA has been explained during a first operational mode, wherein the amplifier can detect and measure physiological measuring signals.
Figure 2 schematically depicts a setup of the amplifier RA during operation. It shows several electrodes EL (two in this embodiment) having respective impedances Zell, Zel2, that are connected to respective inputs 1 of the amplifier RA. The electrodes EL are in electrical contact with a conducting body P, for example of a patient, to receive leakage current there-from.
As follows from the drawing, the conducing body P can be grounded or connected to a null-line, in this case via a grounding electrode EG having a respecting impedance Zpgn, the grounding electrode EG being connected to null-line IS. A mains power line MPL is shown, as well as a ground line GND, and also a mains induced leakage current Iieak. Input impedances of amplifier inputs 1 are schematically indicated by Zinp. Two virtual capacitances Cpowl, Cpow2 are shown, regarding electromagnetic coupling between the mains power line MPL and the body P and input lines of the amplifier RA. Also, a virtual capacitance Cbody is indicated regarding electromagnetic coupling between the body P and ground line GND. A virtual capacitance Ciso concerns electromagnetic coupling between the nulldine IS and the ground line GND. The skilled person appreciates that said leakage current can rung from the mails power line MPL to the ground line GND via paths running through such capacitances (capacitive coupling). During operation, as follows from the above, more than two electrodes EL can be connected to respective inputs 1 of the amplifier RA.
The reference amplifier RA does not have to operate all the time to provide a said digital reference signal, and to convert the digital reference signal into said analogue reference signal.
In one mode of operation the amplifier’s the signal processor 210 is designed for processing the signals received from the one or more analogue-digital converters 5, a the signal processor 210 is configured for processing signals received from the one or more analogue digital converters 5 for determining or estimating an impedance Zell, Zel2 of at least one of the number of electrodes EL.
In the present example, the signal processor 210 is configured to determine a common mode signal Vc, and to use the determined common mode signal to determine or estimate the impedance Zell, Zel2 of at least one of the number of electrodes EL. Also, in this example, the number of electrodes EL placed in electrical contact with the conducting body P to receive the leakage current there-from, for example a body of a living species, for example a patient.
In this example, the mains power line MPL fucntion as a leakage current generator configured to induce or generate said leakage current Iieak.
According to a further preferred embodiment the system (e.g. the amplifier RA) includes a memory for storing at least one predetermined input impedance Zimp of the reference amplifier apparatus RA, wherein the signal processor 210 is configured to use the at least one predetermined input impedance Zimp of the reference amplifier apparatus to calculate or estimate the impedance Zell, Zel2 of at least one of the number of electrodes EL.
Figure 4 shows an alternative example, wherein the leakage current generator includes a signal source 50 that is connected to a first input of a second impedance transforming amplifier, and wherein the system includes first electrical connecting means 52, 53 (for example a conductor 52 and electrode 53) for connecting an output of the second impedance transforming amplifier 51 to a first position of the conducting body P. In this alternative configuration, the system includes second electrical connecting means 54, 55 (e.g. a conductor 54 and respective electrode 55) for connecting a second input of the second impedance transforming amplifier 51 to a second position of the conducting body P. In this example, both positions are relatively close to each other, e.g. on the same body part (a leg). In this embodiment, the signal processor 210 can be configured to calculate an amplitude of a leakage current signal present in the channel 1 of the electrode EL of which the impedance is to be estimated or determined.
Particularly, the signal processor 210 may make use of the above equation (1) or a derived equation for the relation between the signal Vin at an input 1 of the reference amplifier RA, the common mode signal Vc and impedances Zinp, Zel (as indicated in Fig. 3).
During operation of the embodiment of Figures 1-2, a method can be carried out by the reference amplifier wherein at least part of the mains power induced leakage current Iieak (mains interference) is supplied via the electrodes EL to the reference amplifier RA, wherein a resulting signal is being processed by said signal processor 210 for determining or estimating the impedance Zell, Zel2 of at least one of (e.g. each of) the number of electrodes EL.
The signal processor 210 can determine the common mode signal Vc, use the determined common mode signal to determine or estimate the impedance Zell, Zel2 of each of the electrodes EL. As follows from the above, the signal processor 210 can calculate the amplitude of the leakage current signal present in the channel 1 of the electrode EL of which the impedance is to be estimated or determined. The signal processor 210 can use a predetermined input impedance Zimp of the reference amplifier apparatus (e.g. stored in a memory) to calculate or estimate the electrode impedance Zell, Zel2.
Thus, a straight-forward method is presented to swiftly determine, i.e. estimate or calculate, electrode impedances using a reference amplifier RA. A result of the electrode impedance determination can e.g. be output via a digital output 219 of the amplifier RA, for example to be processed further, presented to a user et cetera.
Figure 4 shows an extra advantageous method wherein at least part of the leakage current is induced by a dedicated leakage current generator, generating an alternating leakage current signal, for example a sinus wave signal. In this case, a signal source 50 of the current generator has been connected to a first input of a second impedance transforming amplifier 51 and an output of that second impedance transforming amplifier 51 has been electrically connected to a first position of the conducting body P. A second input of the second impedance transforming amplifier 51 has been is electrically connected to a second position of the conducting body P.
In this way, again, a leakage current can run through the body P and measuring electrodes EL to the reference amplifier RA, effecting respective input signals Vin, Vin’ at amplifier inputs 1, 1’. Again, as in the embodiment of Fig. 2, the signal processor 210 may make use of the above equation (1) or a derived equation for the relation between each signal Vin at an input 1 of the reference amplifier RA, the common mode signal Vc and impedances Zinp, Zel (as indicated in Fig. 3).
Good results can be obtained in case the leakage current is an alternating current, preferably having a constant frequency, for example a frequency lower than 100 Hz, and preferably having an amplitude lower than 10 mV, for example lower than 5 mV, for example about 1 mV or 2 mV. Preferably, as in the embodiment of Figure 4, the alternating signal has a frequency that differs from a frequency of local mains electricity, the frequency of the alternating signal for example being in the range of 1 to 100 Hz.
Besides, as follows from the above, in a method according to the invention each channel is preferably provided with a compensation amplifier 212 of which a first input is arranged for receiving the signal of a respective input amplifier 3, and of which an output is directly or indirectly coupled to a said analogue-digital converter 5, wherein a second input of the compensation amplifier 212 is coupled via a digital-analogue converter 213 to said signal processor 210 for receiving a control signal therefrom. During operation the signal processor 210 may digitally averages the signals received from the one or more analogue-digital converters 5.
Advantageously, the invention described here can use the properties of the Reference amplifier RA to measure the impedance of the electrodes without an electronic circuit in front of the measurement amplifiers 3. This invention can use the common mode signal introduced by the leakage current to measure the impedance of the electrodes. In case there is no leakage current add a leakage current circuit can be includes in a patient ground lead, which does not influence the properties of the measurement electrodes.
Preferably, the measurement is continuously or semi-continuous, which gives the advantage that impedance variation can be measured simultaneously with the bio-electrical signal. The impedance variation can be more or less correlated with body movements, therefore, any DC offset movement artefacts can be rejected (e.g. automatically by the processor 210, or afterwards by a processor processing amplifier output signals, or a user of the system) using the present impedance measurement procedure.
For example, in case variations in the electrode impedances are used to reject the movement artefacts in the measured electrode signal, it is not necessary to have the absolute value of the impedance. In such a case, 50Hz or 60Hz signal (powerline interference) can be extracted by band-pass filtering, demodulated, and used as a reference signal for dynamic motion artifact cancellation by e.g. an adaptive filter. A possible implementation can be provided by an LMS adaptive filter.
As follows from the above, in case the absolute value of the electrode impedances is to be measured, an artificial leakage current can be used, wherein an impedance of the patient ground can be known or almost zero. This can be achieved by using a virtual ground, and in the feedback loop a small common mode signal is added (as in Figure 4).
In case the impedance of signal source 50 is zero, this type of feedback is known as virtual ground. The patient ground impedance is very low. As an example this circuit can keep the common mode signal Vc on the body P at 1 mV, with a frequency of 35 Hz. The common mode signal is known, and the input impedances of the reference amplifier can be predetermined. By this procedure the absolute electrode impedance of all the electrodes can be calculated. The common mode signal Vc can easily be rejected by common mode rejection after the calculation of the impedances. It is e.g. possible to split the output signals (output from output line 219) in a computer; one chain having the raw acquired data for impedance measurement, and one chain for the common mode rejected signals. A continuous impedance measurement is possible this way, without influencing the input of the amplifiers, and without adding parallel amplifier stages. Even in case the electrode impedances are not equal, the common mode rejected signals do not have to contain the leakage current induced signal anymore, the signal being far below noise level.
Examule
Suppose the amplitude of the common mode signal of the signal source 50 is lmV, 35 Hz. The electrode impedances of all the reference amplifier input electrodes EL is 50kOhm. The input impedances of the reference amplifier input amplifiers 3 is lOOOMOhm. In that case the common mode signal on the input of the amplifiers Vcminp will be Vcminp=1000M/(1000M+50k)*lmV=999,50002 qV.
In case the common mode rejection equals lOOdB, the common mode part in the output signal will be 0,009995002 uVolts, far below noise level. Using proper software algorithms the 50kOhm can be calculated, for instance with the use of a lock-in algorithm or a matched filter or a very steep inverse notch filter.
Suppose we now have the situation that five electrodes EL are connected to five amplifier inputs 1, wherein four respective electrode impedances (channels 2-5) are low, IkOhm, and one (channel 1) high, IMOhm. This situation gives the next figures:
Channel 2,3 4 and 5 will contribute to the common mode:
Vcminp2,3,45=1000M/(1000M+lk)*lmV=999,999000 qV.
The contribution of channel 1 to the input common mode is:
Vcminpl=1000M/(1000M+lM)*lmV= 999,000999 qV.
In that case the real common mode will be:
= 999,7993998 μν.
This means that channel 2..5 will have a differential common mode of: Vdifcm=-0,1996002 μν
And channel 1 will have a differential common mode signal of: Vdifcm=0,798408 μΥ
The differences from the common mode in the channels are amplified by the gain, in this case 20. This means that after common mode rejection the 35Hz still will be visible in the signal. But the electrode impedance of 1 M really is relatively high, so in normal conditions no problem with the common mode rejection will occur. And the mean electrode impedance and the impedances of the different channels can easily be calculated.
The circuit that introduces the artificial common mode signal of lmV, 35Hz takes care of the fact that the electrode impedance of the patient ground will be very low. The open loop of an operational amplifier can easily be 1 M, so the real impedance of the patient ground is almost zero. As already described this is called virtual earth, and has the advantage that the 50/60Hz mains interference is reduced to a minimum.
It follows that using a normal leakage current between the amplifier R and the body P, an impression of the electrode impedances and the variation in the impedances can be measured without adding electronic circuitry to the input of the amplifier .The impedance can be measured ondine without interfering with the acquired bio-electrical signals. The impedance measurement can be performed completely in the software of the computer, without decreasing the integrity of the bio-electrical signals. It is not necessary to use any parallel amplifier to perform the impedance measurement. In case no leakage current is available an artificial leakage current is fed to the patient ground electrode using a so called grounding earthing grounding circuit. The absolute value of the electrode impedance can be calculated without decreasing the properties of the input amplifier. Also, the absolute value of the electrode impedances can be calculated on-line and continuously. Preferably, an amplitude of the artificial common mode signal is controllable or adjustable in case the electrode impedances and especially the differences are high.
Although the illustrative embodiments of the present invention have been described in greater detail with reference to the accompanying drawings, it will be understood that the invention is not limited to those embodiments. Various changes or modifications may be effected by one skilled in the art without departing from the scope or the spirit of the invention as defined in the claims.
It is to be understood that in the present application, the term "comprising" does not exclude other elements or steps. Also, each of the terms "a" and "an" does not exclude a plurality. Also, a single processor, converter or other unit may fulfil functions of several means recited in the claims. Any reference sign(s) in the claims shall not be construed as limiting the scope of the claims.
Claims (19)
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| NL2014897A NL2014897B1 (en) | 2015-06-01 | 2015-06-01 | Method for determining or estimating an impedance of at least one electrode. |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| NL2014897A NL2014897B1 (en) | 2015-06-01 | 2015-06-01 | Method for determining or estimating an impedance of at least one electrode. |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| NL2014897A NL2014897A (en) | 2016-12-12 |
| NL2014897B1 true NL2014897B1 (en) | 2017-02-03 |
Family
ID=55085851
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| NL2014897A NL2014897B1 (en) | 2015-06-01 | 2015-06-01 | Method for determining or estimating an impedance of at least one electrode. |
Country Status (1)
| Country | Link |
|---|---|
| NL (1) | NL2014897B1 (en) |
Family Cites Families (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| NL2008441C2 (en) * | 2012-03-09 | 2013-09-10 | Twente Medical Systems Internat B V | Apparatus for processing signals. |
| EP3235426A3 (en) * | 2012-10-15 | 2018-02-21 | Jordan Neuroscience, Inc. | Wireless eeg unit |
-
2015
- 2015-06-01 NL NL2014897A patent/NL2014897B1/en not_active IP Right Cessation
Also Published As
| Publication number | Publication date |
|---|---|
| NL2014897A (en) | 2016-12-12 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| NL1034223C2 (en) | Device for processing signals. | |
| TWI481196B (en) | Bioelectricity signal sensing apparatus and device for canceling baseline drift in bioelectricity signal | |
| JP5474707B2 (en) | Detection circuit and voltage detection device for voltage detection device | |
| US9717431B2 (en) | Circuit arrangement for suppressing common-mode interference signals during the measurement of bioelectric signals | |
| JP5068811B2 (en) | Method and apparatus for amplifying multiple signals using a single multiplexed amplifier channel with software controlled AC response | |
| JP5191791B2 (en) | Elimination of coherent signals in ECG | |
| JP4415215B2 (en) | Bioelectrical impedance measurement system and method in the presence of interference | |
| US8914099B2 (en) | Biomedical acquisition system with motion artifact reduction | |
| JP6616987B2 (en) | Impedance measuring apparatus and impedance measuring method | |
| CN112969924A (en) | Impedance measuring device | |
| NL2014897B1 (en) | Method for determining or estimating an impedance of at least one electrode. | |
| EP2822458B1 (en) | Apparatus and method for processing signals | |
| JP2017534338A (en) | Wearable device for skin conductance measurement | |
| JP2003235823A (en) | External ac noise eliminating system for biological electric signal | |
| CN104897946A (en) | Current detection device and system | |
| CN105380647B (en) | A kind of faint motional impedance detection device and its method based on four electrode half-bridge methods | |
| JP2962244B2 (en) | PCB passive element isolation measurement circuit | |
| US9817035B2 (en) | Impedance measuring circuit | |
| JP6778515B2 (en) | Impedance measuring device and impedance measuring method | |
| US9936893B2 (en) | Device for detecting electric potentials | |
| KR101848752B1 (en) | Biological signal measurement apparatus having multi-channel for phase separation and common mode noise removal | |
| CN105242095B (en) | The device and method for detecting current potential | |
| US11779253B1 (en) | Measurement of electrode impedances in biopotential-physiological-phenomena sensing systems | |
| JP2017203726A (en) | Impedance measurement device and impedance measurement method | |
| JP2015152382A (en) | Measurement instrument |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| MM | Lapsed because of non-payment of the annual fee |
Effective date: 20210701 |