MXPA97008290A - Recovery of carrier phase on a tdm / t receiver - Google Patents

Abstract

A demodulator for a receiver of digital data messages sent in predetermined time slots within time frames of fixed duration includes an adaptive filter operative in each received data packet to determine digital bit values and to adapt the filter coefficients. The filter coefficient values after the filtering of a data packet in a time slot are used as initial values in the adaptive filtering of the next data packet received in the corresponding time slot of the next quad.

Description

RECOVERY OF CARRIER PHASE ON A TD / TDMA RECEIVER DESCRIPTION OF THE INVENTION The present invention relates to a demodulator for a signal receiver of digital data messages sent in predetermined time slots within time frames of fixed duration. Numerous methods of equalizer adaptation (ie, adaptive digital filtering) have been developed and widely applied. The most widely reported are the algorithms known as Medium Minimum Squares (LMS) and Minimum Recursive Squares (RLS) algorithms. The fundamental difference between the two types is the criterion of minimization of error used to adjust the filter coefficients. As its name suggests, LMS minimizes the (average) value of statistical expectation of the error and theoretically only converges to an optimal solution after an infinite number of iterations. In contrast, RLS minimizes the instantaneous error for a given group of operation parameter and has convergence properties directed only by the data that is being given to the procedure. The emergence of these two types of method can be attributed to their relative advantages and disadvantages: LMS is comparatively slow to converge, making moderate tracking poor at fast channel variations, but efficient to implement. RLS converges rapidly, has good tracking properties, but has a high computational cost and susceptibility to instability. Over the years, both "Fast" and compromise variants of the basic RLS algorithms have been developed, in an attempt to reduce their computational requirements, but these are still 5 to 10 times more computationally intense than LMS. A coverage of LMS, RLS and adaptive techniques is generally given in the book "Adaptative Filter Theory" by Simon Haykin, Prentice Hall 1991, 2a. edition. Adaptive filters (equalizers) are used in networks of TDM / TDMA to compensate for multipath interference. The signals are reflected from buildings, mountains and high-side vehicles, and thus can take several trajectories between a transmitter and a receiver. As discussed in Cellular Radio Systems, DM Balston and RCV Macario Editors, Artech House Ind., 1993, page 167 et seq., The equalization is taken by estimating the signal transfer properties of the transmission medium (for example, determining the response of impulse) and then processing the received signal and consequently, compensate. There are several known methods for estimating the transfer function of the transmission path and most of these methods are based on the reception of an expected data sequence. This is a training sequence sent as part of a data packet. The receiver detects the sequence and knowing that bit symbol pattern (1.0, etc.), that is, symbol, pretends to be present, is able to estimate the transfer function most likely that produced the received signal, and the coefficients filter (equalizer) required to compensate for multipath distortion. In known mobile TDM / TDMA networks, ie those with mobile subscribers, the propagation delays can vary from frame to frame to such a degree that a full re-teaching of the equalizer is necessary before the demodulation of each newly received data packet. . Unfortunately, this means either that an RLS algorithm must be used at a high computational cost, or that a large number of training symbols must be incorporated into each data packet in order to be able to retrain, with less than other data that are sent The invention is defined in the claims to which reference should now be made. Preferred aspects are set forth in the sub-claims. The present invention, in its first aspect, preferably comprises a demodulator for a TDM / TDMA receiver unit which includes adaptive filter means on each data packet received in each time slot of a frame to determine the digital bit values and for adapt the filter coefficients, wherein the values of filter coefficients after filtering a data packet in a time slot are used as initial values in the adaptive filtering of the next data packet received in the corresponding time slot of the following table. In the periods between the corresponding time slots, the filter coefficients are preferably stored in a memory to be reused. Preferably, the demodulator includes operative correlation means for performing a complex correlation between the received and expected synchronization data, to determine the carrier phase at a predetermined symbol in the received packet. By "complex correlation" is meant a correlation of data, which include values that have both real and imaginary parts. Said demodulators are particularly applicable in TDM / TDMA networks having base stations and subscriber units, which substantially have fixed sites. Although fade effects due to multipath propagation can also occur, these effects change only slowly, compared to the speed of the transmission frame. The preferred demodulator takes into account the slowly variable nature of multi-path propagation expected by reusing filter coefficients adapted from previous frames. Consequently, the length of training sequences can be greatly reduced by providing a large proportion of available bandwidth for user data. In the preferred TDM / TDMA network, which includes demodulators according to the present invention, the data packets continue to include data sequences suitable for training, although the sequence is short. The preferred demodulator receives this expected sequence in order to determine the carrier phase and packet time, pro not necessarily to capacitate the adaptive filter coefficients. The preferred demodulator according to the present invention advantageously minimizes the amount of training data required, thus maximizing the available bandwidth for user data and to avoid the use of an RLS adaptation algorithm. Consequently, the preferred demodulator can be of construction if simple and has a low power consumption. Also, using the filter coefficients of the corresponding data packet of the previous frame as a starting point, a slow and simple convergence can be used to implement the filter coefficient adaptation method. The present invention also relates to a method for adaptive filtering of each data packet received in each time slot of a frame to determine digital bit values and to adapt filter coefficients, where filter coefficient values are used after of the filtering of a data packet in a time slot, as initial values in the adaptive filtering of the next data packet received in the corresponding time slot of the following frame. The present invention in its second aspect provides a demodulator which preferably includes operational correlation means for performing a complex correlation between received and expected synchronization data to determine the carrier phase at a predetermined symbol in the received packet. This presents the advantage of a computational efficiency and speed of phase acquisition. The present invention also relates to a corresponding method for determining the carrier phase in a demodulator. By way of example, reference will now be made to the accompanying drawings in which: Figure 1 is a schematic diagram illustrating the system including a base station (BTE-Base Termination Team) and subscriber unit ( Termination of NTE-Network); Figure 2 is a diagram illustrating a frame structure and time calculation for a double link; Figure 3 is a schematic diagram showing different types of data packets transmitted from a base station to a subscriber unit (i.e., downlink); Figure 4 is a block diagram representing the symbol processor of the demodulator in a subscriber unit; Figure 5 is a block diagram illustrating the correlator shown in Figure 4; Figure 6 is a block diagram illustrating the rotator and an Automatic Gain Control (AGC) shown in Figure 4; Y Figure 7 illustrates an equalizer output quantization according to p / 4-Differential Square Displacement Key Modulation Scheme.

The Basic System As shown in Figure 1, the preferred system is part of a telephone system, in which the local cabling loop of the exchange to the subscriber has been replaced by a full double radio link between a fixed base station (BTE) and a fixed subscriber unit (NTE). The preferred system includes a double radio link (Air Interface), and transmitters and receivers to implement the necessary protocol. There are similarities between the preferred system and digital cellular mobile telephone systems such as GSM, which are known in the art. This system uses a protocol based on a layered model, in particular the following layers: PHY (Physical), MAC (Medium Access Control), DLC (Data Link Control), NWK (Network). A difference compared to GSM is that, in the preferred system, the subscriber units are in fixed locations and there is no need for loose command arrangements or other aspects in relation to mobility. This means, for example, in the preferred system, that directional antennas and grid electricity can be used. Each base station in the preferred system provides six double radio links to twelve frequencies chosen from the full frequency distribution, in order to minimize interference between nearby base stations. The frame structure and time calculation for a double link is illustrated in Figure 2. Each double radio link comprises an uplink from a subscriber unit to a base station and, at a frequency offset, a downlink from the base station to the subscriber unit. The downlinks are TDM, and the uplinks are TDMA. The modulation for all links is p / 4 - DQPSK, and the basic frame structure for all links is ten slots per 2560-bit frame, that is, 256 bits per slot. The bit rate is at 512kbps. The downlinks are continuously transmitted and incorporated into a broadcasting channel for essential system information. Where there is no user information that is transmitted, downlink transmissions continue to use the basic box and slot structure and contain an adequate fill pattern. For both uplink and downlink transmissions, there are two types of slots: normal slots, which are used after a call placement, and pilot slots used during call placement. Each normal downlink slot comprises 24 bits of synchronization information followed by 24 bits designated as field S, which includes an 8-bit initiator followed by 160 bits designated as field D. This is followed by 24 bits of Front Correction of Errors and an 8-bit queue, followed by 12 bits of the broadcasting channel. The broadcasting channel consists of segments in each of the slots of a frame, which together form the common downlink signaling channel, which is transmitted through the base station, and contains control messages containing information of links such as lists of slots, information of multiple frames and super frames, messages without connection, and other basic information for the operation of the system. During the call placement step, each downlink pilot slot contains frequency correction data and a training sequence to receive the initialization, with only a short S field information and no field D. The uplink slots basically contain two different types of data packets. The first type of packet, called a pilot packet, is used before a connection is established, for example, for an ALOHA call request and to allow an adaptive time alignment. The other type of data packet, called the normal packet, is used when a call has been established and is a larger data packet, due to the use of adaptive time alignment. Each normal uplink packet contains a 244-bit data packet, which is preceded and followed by a ramp with a duration of 4 bits. The remaining ramps and bits left in the 256-bit slot provide a security gap against interference from the surrounding slots due to time calculation errors. Each subscriber unit adjusts the time calculation of its slot transmissions to compensate for the time it takes the signals to reach the base station. Each normal uplink data packet comprises 24 bits of synchronization data followed by an S field and a D field of the same number of bits as in each normal downlink slot.

Each uplink pilot slot contains a pilot data packet, which has a length of 192 bits preceded and followed by 4-bit ramps, defining an extended security gap of 60 bits. This larger security gap is necessary, since there is no available time calculation information and without it, the propagation delays could cause the surrounding slots to interfere. The pilot pack includes 64 bits of sync. followed by 104 bits of field S, which starts with an 8a initiator and ends with a Cyclic Redundancy Check of 16 bits, 2 inverted bits, 14 FEC bits, and 8 tail bits. There is no field D. The S fields in the aforementioned data packages can be used for two types of signaling. The first type is MAC (MS) signaling and is used to indicate between the MAC layers of the base station and the MAC layer of a subscriber unit, so the calculation of time is important. The second type is called associated signaling, which can be slow or fast and is used to signal between the base station and the subscriber units in the DLC or NWK layers. Field D is the largest data field, and in the case of normal telephony contains samples of digitized language, but may also contain data samples without language. The provision is made in the preferred system for subscriber unit authentication using a challenge response protocol. General cryptic coding is provided by combining the language or data with an unpredictable sequence of encrypted bits produced by a key current generator, which is synchronized to the transmitted super-frame number. In addition, the transmitted signal is mixed to remove the components of. The demodulator of the subscriber unit has to do with the physical reception of data transmitted in the base address to the subscriber (downlink). There are currently types of downlink packets, two of which are shown in Figure 3. From the demodulation perspective, the third type of package (Available Package) is equal to the Pilot Pack shown, except that the data field, DOWN-P-DATA (descending P data) is replaced by a fixed fill pattern.

The Demodulator of the Subscriber Unit The following functions are taken by a subsection of the subscriber unit demodulator apparatus known as the Symbol Processor: Sync Correlation (sync detection, slot time calculation recovery, initial carrier phase recovery), Digital AGC, Equalization, Carrier Phase Tracking, and Deviation (symbol decisions). The Symbol Processor operates as a basic coherent (unmatched) receiver, a linear equalizer, or a decision feedback equalizer (DFE). Which is the best for any particular subscriber unit that will be governed by the characteristics of the RF propagation path. The basic receiver is probably to perform the best, where the multipath effects are not significant, the linear equalizer will offer a performance benefit, where the multipath interference is present, but is not severe, and the DFE has the potential to operate through severely dispersive channels.

Symbol Processing The functions performed by the Symbol Processor are shown in Figure 4, which is a flowchart, in which the double-edge arrows denote complex data paths.

The output signal of the radio frequency (RF) section (not shown) of the subscriber receiver is digitized and presented to the symbol processor at the base of the band as a sequence of complex samples. These samples are regulated to allow non-real time processing. The demodulated bit sequence (output), which can be a normal or pilot packet, or a fragment of broadcast data, depending on the mode of operation, is passed to a separate circuit block responsible for the bit-level protocol processing and processing. With the exception of correlator 2, which operates at the input sample rate, all processing is performed iteratively at the symbol rate. The time calculation is arranged in such a way that the sequence of Sync. Slot received from the captured pack falls within a predetermined region of the input slider used by the correlator 2.

Complex Correlation The complex correlation in correlator 2 with a stored representation of the sync. Expected (Sync. Slot or Sync. Box) then produces estimates of instantaneous carrier phase and signal level (gain), which are subsequently used to scale, and phase alignment (ie, rotate), the input data samples . The rotation is taken by the rotator 3, to establish the average path of the carrier phase through the sync sequence. and that has a zero degree reference defined by the sync pattern. stored. The scale is taken through the operation of the Automatic Gain Control (AGC) 1 circuit system. The sync sequences expected (Sinc. Ra nura in slots 1 to 9, Sync. Box in slot 0) each are stored as two sequences of samples N, one sequence being the real components, ReY [n] as shown in Figure 5 , and the other sequence being the imaginary components, I mY [n] as shown in Figure 5. The sequence Y [n] represents the expected constellation points produced by optimally sampling a baseband carrier signal, which has been p / 4 - DQPSK modulated with a Binary Slot Sync or Frame Sync sequence, and filtered through a matching reception fi lter. The sequences Y [n] are stored either as constants with rigid wire or preferably programmed to static registers 16. The correlator 2 processes one sample per symbol from a shift register 18, which maintains the input data X [n ] of the slot controller (not shown), real and imaginary components ReX [n] and I mX [n] being maintained separately. Static recorders 16 maintain the expected values Y [n]. The shift register 18 is updated once per input sample and effectively maintains the decimated sequences of the sync window. (see below), for example, sample 1, 3, 5, 7, in the case of two samples per symbol. As shown in Figure 5, the correlator consists of two main functional blocks. A block 20 takes the sum of product calculations in the real component of the input data ReX [n]. The other block 22 takes the sum of product calculations in imaginary component of the input data I mX [n]. The respective real and imaginary output signals 24, 26 of the sum of product circuits 20, 22 are combined in the respective adders 28, 30, to provide real and imaginary components ReRxy, ImRxy, of a discrete cross correlation function Rxy [ n]. The received Sync sequence is known to occupy a certain region of the slot controller upon receiving a packet. The cross-correlation is made through a limited region of the slot controller (sync window), which is known to continue the Sync input pattern. For each element of the correlation function, the output energy is evaluated by squaring the squares 32, 34 and adding in the adder 36. An energy peak is detected by the peak detector 38, when the expected sequence Y [n ] and the sync sequence Decimal entry are aligned in time. The detector then outputs a peak signal Rxy (peak), which is independent of the input carrier phase. The reciprocal movement of the peak energy value Rxy (peak) is determined and comes out as a scaling factor applied to the AGC 1 circuit system as shown in Figure 6. After the peak is detected, the adders 28, 30 provide real and imaginary peak energy components Re Rxy (peak) and Im Rxy (peak), which are applied as phase correction signals to the rotator 3, as shown in Figure 6. As illustrated in Figure 6, in rotator 3, the real and imaginary components of input data samples ,, ReX [n] and ImX [n] are respectively multiplied by the real and imaginary peak energy values Re Rxy (peak) and Im Rxy (peak). The resulting real and imaginary products were summed to give corrected output signals of phase 42, 44. These output signals 42, 44 are applied to the AGC 1 circuit system to scale by the scale factor before leaving as phase and samples corrected gain ReX [n] 'and ImX [n] \ Demodulation The phase and the corrected gain samples, starting with a very close to half sync. they are applied to the main demodulation loop, which performs: the symbol derivation (absolute phase decoding); Carrier tracking (phase locked loop); multiple path equalization. The equalizer is implemented in four sections: an advance feed filter 2, a feedback filter 4, a quantizer 8, and a filter adaptation mechanism. The two filter sections, each consisting of a complex tapered delay line (ie, a Finite Impulse Response filter) with variable taper weights (ie, coefficients). The feed feed filter 4, which has at least one delay / coefficient element per symbol period, takes the input data from the AGC 1 block, rolls the samples held in its taper delay line with the group of real coefficient and presents its output to the rotator 10 of the phase locked loop (PLL) 21. Likewise, the feedback filter 4, which has only one delay / coefficient element per symbol period, winds up constellation decisions of the quantizer 8 with a group of additional coefficient. The combined output of feed and feedback feed filters 4, 6, constitutes the output of the equalizer and this particular configuration of filter sections is generally referred to as a decision feedback equalizer (DFE). During operation, the equalizer generates an output sample (equalized) per symbol period, which is fed to the equalizer 8. The function of the equalizer 8 is then compared to the output with the group of "ideal" constellation points that are characterized by the modulation scheme and to select the constellation point, which is closer in the Euclidean sense. This procedure is represented by the modulation scheme p / 4 -DQPSK in Figure 7, which shows an equalizer output sample, X being selected as having the closest constellation point Y 'of the possible Y constellation points. The constellation point Y 'selected, forms the decision of equalizer 8 for the real reception symbol t, as such, the next input sample for the feedback filter 4. The decisions of the successive equalizer 8 are also fed to a circuit of symbol decoding, where they are processed to retrieve the transmitted bit streams. The difference between the output X of the equalizer and the selected constellation point Y represents the decision error Z for the real symbol and this is used by the coefficient adaptation mechanism to drive the error to zero in the long term. The equalizer is said to have been converged when the coefficients, in the feed and feedback feed filters 4, 6, have reached values, which adequately mitigate the effects of interference between symbols. Equalizer coefficients are initialized with constants (zero, except for the 'main taper', which is fixed to the unit) before processing the pilot packet (the extended ETS training sequence is used to initially train the equalizer). Then, the final values in a slot are used as the starting values in the corresponding slot in the following table. The outputs of the two filters are combined on the equalizing side of a phase 10 rotator, which is activated by a locked loop of phase directed by decision 12. The deviation produces a phase error term and, subtracting the vector from Rotator output from the closest candidate constellation point, a symbol error vector for updating the equalizer coefficient. The phase error term is passed to the carrier tracking algorithm, which modifies the actual phase estimate in preparation for the next symbol. A sine query table 13 is used to convert the phase estimate to an equivalent Cartesian representation. At the beginning of each package, or more specifically for the first sample that will be processed, which is the average sample in the sync sequence. , the phase reference (a variable state within the carrier tracking algorithm) is set to zero. Then, it is adapted through a dedicated carrier tracking algorithm.

Two representations of the symbol error vector are required: the unprocessed error to update the feedback and an 'unrotated' error vector, which reintroduces the phase shift removed by the phase locked loop, for feed feed updates . The lack of rotation by the non-rotator 14 is necessary to re-establish the correlation relationship between the decision error and the samples in the feed feed filter. The coefficients are adjusted using the so-called Stochastic Gradient LMS algorithm, although any adaptation algorithm can be used directly. The adaptive properties of the carrier tracking loop are selected to ensure that carrier phase variations (including frequency deviation) are removed by the actions of the phase locked loop, leaving the equalizer to exclusively compensate for multiple path channel variations. Upon completion of the slot demodulation, the coefficients of the equalizer are stored for use in the corresponding slot of the following frame. The operation of the present invention will now be related to the steps involved in a normal and pilot packet processing. To process a pilot package, the following steps are involved: 1) Scan and capture the required pilot packet in the slot controller (in the preferred demodulator, sync processing and packet capture are overlapped to minimize the delay group). ) Restore equalizer coefficients to their values at the end of the preceding slot, one frame before. (For the first pilot packet, the coefficients are initialized with constant data). 3) Correlate for the Slot Sync data (that is, Frame Sync in slot 0) on the sync window. Use the peak output of the correlator to scale and rotate all the samples in the sync region. of the slot controller.

This aligns the phase of the input carrier with the equalizer coefficients. 4) Pass the samples Sync input scaled and rotated through the demodulator / equalizer, adapting the equalizer coefficients and the local phase reference based on the sequence (Sync.) Of known symbol. (5) Demodulate the Sync sequence to provide an indication of the integrity of the packet. A sync sequence received in error can be used, for example, to inhibit matching of the equalizer thus avoiding potential corruption. (6) Correlate the ETS extended training sequence on the sync window. delayed Use the peak correlator output to scale and rotate the samples in the ETS and DOWN-P-DATA regions of the slot controller. This aligns the phase of the input carrier with the coefficients of the equalizer. 7) Determine the peak deviation of the sync position. nominal and, if necessary, realign the demodulator box to compensate. 8) Restore the local phase reference (to 0 degrees) and then pass the scaled and rotated ETS samples through the demodulator / equalizer, adapting the coefficients of the equalizer and the phase reference based on the known sequence (ETS). This is the normal training procedure. 9) Pass the samples (scaled and rotated) DOWN-P-DATA through the demodulator / equalizer, adapting the equalizer coefficients and the phase reference based on the constellation decisions. This is typically a decision-driven adaptation. The demodulated DOWN-P-DATA contribution is passed over the bit-level protocol processing.

) Store the equalizer coefficients for the next lotus or normal packet in this carrier (ie in the following table). A switch to normal packet reception occurs once the equalizer has been successfully trained from the pilot packets. The preferred procedure for normal packet reception is then as follows: 1) Scan and capture the normal packet required in the slot controller (in the preferred demodulator), the sync processing and the packet capture are overlapped to minimize the group delay). 2) Restore the equalizer coefficients to their values at the end of the preceding slot, a previous frame. (For the first normal package, the coefficients are established during the pilot training). 3) Correlate for the Slot Sync data (that is, Frame Sync in slot 0) on the sync window. Use the peak output of the correlator to scale and rotate all the samples in the slot controller. This aligns the phase of the input carrier with the coefficients of the equalizer. 4) Pass the samples Sync input scaled and rotated through the demodulator / equalizer, adapting the equalizer coefficients and the local phase reference based on the sequence (Sync.) Of known symbol. Demodulate the Sync sequence to provide an indication of the integrity of the packet. A sync sequence received in error can be used, for example, to inhibit matching of the equalizer thus avoiding potential corruption. 5) Pass the samples (scaled and rotated) DOWN-P-DATA through the demodulator / equalizer, adapting the equalizer coefficients and the phase reference based on the constellation decisions. This is typically referred to as a decision-driven adaptation. The demodulated DOWN-P-DATA is passed over the bit-level protocol processing. Store the equalizer coefficients for the next pilot or normal packet in this carrier (that is, in the following table).

Claims (3)

1 .- A digital demodulator of data messages received as data packets in time slots within time frames of fixed duration, including operational correlation means to perform a complex correlation between the received synchronization data (x [n] ) and expected (and [n]) to determine real and imaginary components (ReRxy, ImRxy) of the correlation in predetermined symbols in a received data packet, and also includes a peak operational energy detector to determine the real and imaginary components (ReRxy (peak), Im Rxy (peak)) of the correlation at which the peak energy occurs, and operational application means to apply the determined values as the real and imaginary components (ReRxy (peak), ImRxy (peak)) of the correlation at which the peak energy occurs in order to effect an adjustment of the received symbol phase in the subsequent demodulation of received data, the application means give a rotator in which the real and imaginary components of the received symbols (x [n]) are multiplied by the real and imaginary components of the correlation (ReRxy (peak), ImRxy (peak)), at which peak energy occurs , and operative adder means to sum the resulting real and imaginary products to give corrected exit symbols. 2 - A digital data message demodulator according to claim 1, wherein the peak energy detector is operative to determine a peak energy value and the application means are further operative to apply the peak energy to correct the magnitude of the exit symbols. 3. A digital data message demodulator according to claim 2, wherein the application means correct the magnitude of output symbols by escalating with the reciprocal movement of the peak energy value. 4. A digital data message demodulator according to any of the preceding claims, wherein the received synchronization data is selected by the reception that has been in a predetermined position or positions within the received data packet. 5. A demodulator of digital data messages according to any of the preceding claims, wherein the received data packet is stored in a memory for processing. 6 - A digital data message demodulator according to any of the preceding claims, wherein the correlation means comprise first means and second means, the first means being operative to determine the sum of product values for real data components of received synchronization multiplied by the corresponding real and imaginary components of the expected synchronization data, the second means being operative to determine the sum of product values for imaginary components of the received synchronization data multiplied by the real and imaginary components of the data of expected synchronization, the respective real and imaginary output signals of the first means being combined by the combining means to provide real and imaginary components of the correlation, the demodulator further comprises means for squaring, operative to provide the peak energy detector values proportional to the real and imaginary components of the squared correlation, the combining means providing the real and imaginary components to which the energy peak occurs. 7. A demodulator of digital data messages according to any of the preceding claims, including adaptive filter means operative on each data packet received in each time slot of a frame to determine the digital bit values and to adapt filter coefficients, where the values of the filter coefficients after filtering a data packet in a time slot, are used as initial values in the adaptive filtering of the following data packet received in the corresponding time slot of the following table. 8. A demodulator for a receiver of digital data messages sent in predetermined time slots within time frames of fixed duration according to claim 7, wherein in the periods between the corresponding time slots, the filter coefficients they are stored in a memory to be used again. 9 - A receiver for a receiver of digital data messages sent in predetermined time slots within time frames of fixed duration according to any of the preceding claims. 10. - A receiver according to claim 9, which is a subscriber unit operable to receive time division multiplexer (TDM) data signals. 1 1 .- A receiver according to claim 10, which is a subscriber unit that has a fixed site. 1

2. A receiver according to claim 9, which is an operational base station for receiving timestamp multiple access (TDMA) data signals. 1

3. A receiver according to any of claims 9 to 12, which operates to receive digital data messages sent by radio. 14 - Means of communication comprising a plurality of subscriber units each operable to receive digital data messages comprising data packets in predetermined time slots within fixed duration time frames of a base station, and the station operating base for receiving digital data messages comprising data packets in predetermined time slots within time frames of fixed duration of the subscriber units, the base station and the subscriber units each comprising a receiver in accordance with Any of claims 9 to 13. 15. A method for demodulating a message of digital data received as data packets in time slots within time frames of fixed duration, including carrying out a complex correlation between synchronization symbols. received (x [n]) and expected (y [n]) to determine the real and imaginary components (ReRx and, ImRxy) of the correlation to predetermined symbols in a received data packet, determining the real and imaginary components (ReRxy (peak), ImRxy (peak)) of the correlation at which the peak energy occurs, and applying the values determined as the real and imaginary components of the correlation at which peak energy occurs, so that the real and imaginary components of received symbols are multiplied by the real and imaginary components of the correlation at which the peak energy occurs and by summing the actual products and resulting images to give the phase-corrected output symbols in order to effect an adjustment of the received symbol phase in a subsequent demodulation of received data. 16. A method for demodulating a digital data message according to claim 15, wherein a peak energy value is determined and applied to correct the magnitude of output symbols. 17 - A method for demodulating a digital data message according to claim 16, wherein the magnitude of the output symbols is scaled by the reciprocal movement of the peak energy value. 18. A method for demodulating a digital data message according to any of claims 15 to 18, wherein the sum of the product values is determined for the actual received synchronization data components multiplied by the real and imaginary components corresponding to the expected synchronization data, the real and imaginary output signals combine to provide real and imaginary components of the correlation, proportional values are provided to the real and imaginary components of the squared correlation, a peak energy is detected and real and imaginary components of the correlation are provided, at which the energy peak occurs. 19. A method for demodulating a digital data message according to any of claims 14 to 16, which includes the adaptive filtering of each of the received data packets to determine the digital bit values and to adapt filter coefficients , wherein the values of filter coefficients after filtering a data packet in a time slot are used as initial values in the adaptive filtering of the following data packet in the corresponding time slot of the following table.