MXPA96004220A - Carrier recovery system for a side band signal vestig - Google Patents

Abstract

The present invention relates to a system for receiving a high definition television (HDTV) signal transmitted as a vestigial sideband (VSB) signal formatted as a one-dimensional data constellation of symbols representing digital image data and subjects to exhibit a carrier phase shift, a carrier recovery network for changing the frequency of the received VSB signal to the baseband, the carrier recovery network that is capable of achieving carrier recovery without relying on a pilot component if present in the received signal, the carrier recovery network comprising: an input network for processing the BSV signal, a filter network which responds to an output signal from the input network and which has a buffer response. band with respect to at least one of the upper and lower band edges of a frequency spectrum of said signal VSB to produce an amp signal double sideband modulated (AM) latitude at an output of the filter network; a phase detecting network for processing the dual sideband AM output signal to produce a control signal representing a carrier phase shift when present; and an output processor that responds to the control signal to produce a demodulated output signal at or near the band of

Description

CARRIER RECOVERY SYSTEM FOR A VESTIGIAL SIDEBAND SIGNAL Field of the Invention This invention relates to a digital signal processing system. In particular, the invention relates to a carrier recovery system for use in a receiver of a vestigial sideband (VSB) signal, such as can be modulated with high definition television (HDTV) information, for example.

Background of the Invention The recovery of data from a vestigial sideband signal or quadrature-modulated amplitude (QAM) in a receiver, requires the implementation of three functions: time recovery for symbol synchronization, carrier recovery (frequency demodulation) and compensation. Time recovery is the process by which the receiver clock is synchronized (time base) with the transmitter clock. This allows the received signal to be sampled at the optimum time to reduce the opportunity for a splice error associated with the processing directed to the decision of the values of the received symbols. Carrier recovery is the process by which a received radiofrequency signal, after changing its frequency to a lower intermediate frequency bandpass, is changed in frequency to the baseband to allow information retrieval of the base band modulation. Compensation is a process that compensates for the effects of the alterations of the transmission channel on the received signal. More specifically, the compensation removes the interference between symbols (ISI) of the baseband, caused by alterations of the transmission channel, including the filtering effect of the low pass of the channel. Interference between symbols causes the value of a given symbol to be distorted by the values of the previous and next symbols. For quadrature-modulated amplitude signals, time recovery is usually the first function implemented in a receiver. The time is recovered, either from the intermediate passband signal or from the signal near the baseband, i.e., a baseband signal with a carrier phase shift that is corrected by a carrier recovery network. In any case, the time can be set before demodulation of the baseband. The carrier recovery demodulation process is usually a two-step process. First, the passband signal is demodulated to near the baseband by a frequency changer that uses a "better ** - assumption" over which frequency offset is between the input passband signal and the signal of desired baseband. This frequency change is usually made by analog circuits; that is, before the analog-to-digital conversion in the receiver. Next, compensation is made on this signal near the baseband. Finally, the carrier recovery is performed, which removes any residual frequency offset from the signal near the baseband, to produce a true baseband output signal. This function is performed by digital circuits of the receiver. The compensator is inserted between a first local oscillator that performs the change close to the base band, and the carrier recovery cycle network. This is because the carrier recovery process is typically a decision-driven process (as it is known) that requires at least a partially open "eye" that is provided by the compensator function. A quadrature modulated amplitude signal is represented by a constellation of two-dimensional data symbols defined by the Real and Imaginary axes. In contrast, a vestigial sideband signal is represented by a constellation of one-dimensional data symbols in which only one axis contains quantized data to be retrieved at a receiver. The -•'*. Synchronized demodulation of a vestigial sideband signal has usually been performed with the aid of a pilot signal. The pilot signal facilitates the demodulation of the vestigial sideband signal to the baseband in a single step, typically without residual phase or frequency errors. The performance of time recovery, demodulation, and compensation functions in the order in which they are performed for quadrature modulated amplitude signals does not work for vestigial sideband signals using conventional techniques. For quadrature modulated amplitude signals, several time recovery methods are known that are independent of the frequency offset between the signal near the baseband and the baseband signal. However, it is generally accepted that recovery of time regardless of frequency for vestigial sideband signals is not feasible. For this reason, in vestigial sideband systems, absolute demodulation has been historically implemented up to the baseband. In a pilot-assisted vestigial sideband system, the pilot component is injected into the baseband signal in the transmitter as a small direct current phase shift. This direct current phase shift generates a carrier "tone", because when the direct current phase shift is multiplied by an alternating "-" signal, such as in the form of a high frequency cosine function, a Carrier tone of similar phase. This carrier tone can be used by a phase locked cycle (PLL) in the receiver demodulator to transfer the vestigial sideband signal to the baseband. Since the pilot represents the direct current component of the vestigial sideband signal, and this DC component is in the middle of the vestigial sideband, the tone appears to be half the modulation "noise" caused by the data themselves. Normally this modulation noise is treated as an unwanted signal for the secured phase cycle of tracking the pilot in the receiver. The pilot can be removed by means of a very narrow bandpass filter before the assured phase cycle. However, this requires a filter with such a narrow bandwidth to achieve a sufficient rejection of data noise, that the phase-locked tracking cycle is not able to track all phase and frequency phase shifts of the signal. input, especially when using consumer grade tuners. The change of the pilot component to the baseband is not easily done in the digital domain. If analog circuits are used, tolerance compensation problems and the effects of temperature will have to be solved. The component.

The pilot also wastes energy, and it is possible that a disturbance of the transmission channel, such as a null channel, cancels the pilot. For these and other reasons, it is now recognized as desirable to provide a system capable of demodulating a vestigial sideband signal up to the baseband without relying on a pilot component of the received signal. An example of a vestigial sideband system that includes a pilot component is the Grand Alliance high definition television transmission system recently proposed for the United States. This system employs a vestigial sideband digital transmission format to transport a packet of data stream, and is being evaluated in the United States by the Federal Communications Commission through its Advisory Committee of Advanced Television Service (ACATS). A description of the Grand Alliance high definition television system, as presented to the ACATS Technical Subgroup, on February 22, 1994 (project document) is in 1994 Proceedings of the National Association of Broadcasters, 84th Annual Broadcast Engineering Conference Proceedings, March 20-24, 1994.

SUMMARY OF THE INVENTION In accordance with the principles of the present invention, a carrier recovery demodulation system suitable for use with vestigial sideband signals, conveniently achieves recovery of the carrier without relying on a pilot signal or a signal f equivalent. This is done using a filter network and a phase detector that responds to the output signals from the filter network. The filter network has a bandwidth response with respect to at least one of the upper and lower band edges of the frequency spectrum of the vestigial sideband signal, to produce a double band modulated amplitude (AM) signal Carrier side suppressed at the outlet of the filter network. In accordance with one feature of the invention, the filter network comprises first and second digital edgeband filters having edgeband responses respectively associated with the upper and lower band edges of the frequency spectrum of the sideband signal. vestigial, to produce signals of modulated amplitude of double lateral band in the respective outputs of the first and second filters. The filter responses are complementary to the frequency spectrum of the vestigial sideband signal received at the edge of the band which are filtering the first and second filters, respectively.

BRIEF DESCRIPTION OF THE DRAWINGS In the drawing: Figure 1 is a block diagram of a portion of an advanced television receiver, such as an HDTV receiver, including a carrier recovery system in accordance with the principles of the invention. Figures 2 to 6 illustrate the amplitude versus frequency responses of the signals associated with the operation of the system of Figure 1.

Detailed Description of the Drawings In Figure 1, an analogue high definition television signal modulated in vestigial sideband broadcasting received by an antenna 10, is processed by an input network 14 including radio frequency tuning circuits, a tuner double conversion to produce a suitable intermediate frequency step band to convert to digital form, and appropriate gain control circuits, for example. The vestigial sideband signal received illustratively is an 8-VSB signal with a symbol rate of approximately 10.76 Msymbols / second, which occupies a conventional 6 MHz NTSC frequency spectrum, in accordance with the Grand HD television specification Alliance In a specific manner, the vestigial sideband signal received in that example is an 8-VSB signal having a data constellation of a dimension defined by the following eight data symbols: -7 -5 -3 -1 1 3 5 7 The Nyquist bandwidth for this system is nominally 5.38 MHz, for example, with an excess bandwidth nominally 0.31 MHz in each band edge. The described system can also be used for 16-VSB signals, for example. The output signal from the input processor 14 is converted from the analog form to the digital form by an analog-to-digital converter 16, which operates at a sample rate of 2 samples / symbol. The received vestigial sideband signal may include a pilot component, and has been demodulated by unit 14, such that the center of the 6 MHz band is nominally located at 5.38 MHz. The frequency spectrum of this signal in the The input of the analog to digital converter 16 occupies a scale of 2.38 MHz to 8.38 MHz. When the time synchronization is established, the unit of analog-to-digital converter 16 samples this signal at 21.52 MHz, which is twice the speed of the analogue to digital converter. symbols. The pilot component, which represents the direct current point of the pulse-modulated amplitude signal (PAM) of the original baseband, is nominally 2.69 MHz (the Nyquist frequency), which is 1/8 fsr. In the following discussion: fc is the carrier frequency of the transmitted signal (nominally 5.38 MHz), fst is the frequency of transmitted symbols (10.76 Msymbols / second, or four times the Nyquist frequency, and fsr is the sampling frequency of the receiver (21.52 MHz) In the time lock, fsr = 2fst It results in a carrier lock when demodulating the baseband, fu = 1/4 fsr The digital signal from the analog-to-digital converter unit 16 is applied to two complex bandpass filters 20 and 22, which are mirror image filters around the Nyquist frequency. It displays real and imaginary functions in such a way that the output signals from these filters contain real and imaginary components In Figure 1, a letter "C" designates the signal lines that carry complex signals with real and imaginary components. Signals carry only real components, filters 20 and 22 produce output signals without image components, that is, the output signals contain positive or negative spectral components, but not both. This has the advantage of not generating noisy components that may be difficult to remove subsequently. In this system, filters 20 and 22 are designed to have complex analytical output signals with negative spectral components, as will be seen in Figure 2. The negative spectrum is arbitrary; The positive spectrum could also have been selected. Figure 2 illustrates the negative frequency spectrum encompassed by the bandpass responses of filters 20 and 22, and by the bandwidth of the received vestigial sideband signal applied to the inputs of filters 20 and 22. Actual input signal has positive and negative spectra. The positive spectrum is canceled using known techniques, leaving the negative spectrum. The filter 20 extracts the upper band edge of the negative spectrum of the vestigial sideband signal, and the filter 22 extracts the lower band edge of the negative spectrum of the vestigial sideband signal. The upper band edge is the one that contains the highest frequency components, regardless of whether they are positive or negative components. The lower band edge is associated with the lower frequency components. The band edge responses of the filters 20, 22 and the vestigial sideband signal are interspersed at the Nyquist points. In Figure 2 and in the following figures, a symbol "?" designates a frequency offset of the carrier, such as may be associated with a signal near the baseband, i.e., a signal not completely changed in frequency to the baseband. This phase shift will be discussed in detail in relation to the carrier recovery network (baseband demodulation). The responses of filters 20 and 22 are complementary to the frequency spectrum of the input signal at the band edge that the filters are extracting, as shown in Figure 2. This has the effect of producing a modulated amplitude signal ( AM) of the suppressed double sideband carrier, when there is no pilot component present in the received vestigial sideband signal (Figure 3), and which produces a modulated amplitude signal of the residual double sideband carrier when present a pilot on that side of the band. The response of the filter 20 to the left of the frequency fj is not critical, and the response of the filter 22 to the right of the frequency f2 is not critical. Before setting the time and carrier insurance, these amplitude-modulated signals contain frequency (and faee) shifts that can be used for time and carrier recovery. In a specific manner, the center of the modulated amplitude signal obtained from the output of the upper band edge filter 20 is set to -fc-l / 4 fsl. If a pilot signal is present (such as in the Grand Alliance high definition television system), it would appear on this frequency. The frequency 1/4 fst is a quarter of the symbol frequency, if the signal is treated as a vestigial sideband signal. In a similar manner, the center of the modulated amplitude signal obtained from the output of the lower band edge filter is set to -fc + 1/4 fst (the Nyquist frequency). Time synchronization is achieved when the frequency of the sampling clock input (CL) of the analog-to-digital converter unit 16 is four times the frequency difference between the carriers of these two band-width modulated signals. of upper and lower suppressed carrier (Figure 3). The time recovery system operates as follows. The output signal from the filter 22 is conjugated by the unit 25 to send the spectrum of the filter output signal 22 from negative to positive. This is illustrated by Figure 4. Conjugation is a well-known process performed in unit 25 by first separating the signal into its real and imaginary components using well-known methods. The imaginary component is inverted by multiplying it by a negative unit factor. The inverted imaginary component and the original real component are recombined. The modulated amplitude signal of the recombined lower band edge is multiplied with the modulated amplitude signal of the upper band edge from the filter 20 in the multiplier 26, A modulated amplitude signal produced in the output of the multiplier 26 has the component frequency of carrier fc removed, as illustrated by Figure 5. The removal of this component fc results because the negative carrier of the component of the upper bandwidth cancels the positive carrier of the amplitude modulated signal conjugate from the lower band edge. The amplitude signal "" * "modulated at a suppressed carrier center frequency of 0 1/2 fst is maintained because both amplitude modulated signals are double sideband signals.In the frequency domain, these signals are represented in the baseband as even functions (ie, as only real components), and its convolution is represented in the baseband as functions valued in par.The bandwidth of the amplitude output signal modulated from the multiplier 26 is has doubled by the process of convolution (the multiplication in time produces the convolution in the frequency.) By the 0 pulse of each third sample to zero, the sampling clock frequency of the CLK receiver can be synchronized with the frequency of symbols of the vestigial input sideband signal independently of any carrier phase shift (?) This is done by a phase detector 28 5 in a time recovery network 30 as follows.

The imaginary component of the double-sideband modulated amplitude output signal from the multiplier 26 (Figure 5) is an indication of the magnitude of the bad weather of the signal. The real component indicates the direction of bad weather (the signals of the suppressed carrier of modulated amplitude have an ambiguity of 180 degrees that must be resolved). If this signal of modulated amplitude is perfectly set in time, the imaginary component is absent. The double-sideband modulated amplitude signal from the multiplier 26 is separated into its constituent real and imaginary components by means of a unit 32 in the phase detector 28 using known separation techniques. Using known techniques, a translation unit 34 determines the sign of the real component (for the address information), and multiplies this sign by the separate samples of the imaginary component. The output of the multiplier 36 represents an error signal that is driven to zero by the action of the time control cycle when the time lock is reached. Since the carrier frequency of the double-sideband signal is nominally 1/2 fsr, in the safe of the imaginary component of the output signal from the multiplier 36 it will be zero. The multiplication of the imaginary component with the sign of the real component gives the phase detector 28 the ability to differentiate between positive and negative frequency phase shifts. The output signal from the phase detector 28 is filtered by a low pass cycle filter 38 which contains both an integral line and a proportional line, as is known, and is set to clock at a frequency of 1/2 fsr. The cycle filter 38 is set to clock to process every third sample of the input signal, since the purpose of the cycle is to drive every third sample of the imaginary component to zero. The output of the filter 38 is a direct current voltage that is applied to a controlled voltage oscillator (VCO) 40. The oscillator 40 provides the sampling clock of the CLK receiver for the analog-to-digital converter unit 16 as a function of the Direct current voltage. Time synchronization is achieved when the sampling clock of the analog-to-digital converter provided by the described time recovery system including the network 30, is four times the frequency difference between the carriers of the two amplitude output signals. modulated from filters 20 and 22 (Figure 3). The proportional and integral control portions of the filter 38 can be adjusted as known, by utilizing gain control climbers Kl and K2, respectively. These climbers are set to large values to facilitate acquisition of the signal in acquisition mode, and can be reduced in value during the tracking mode to increase noise immunity. The time required to achieve time insurance varies as a function of the amount of noise and distortion of multiple lines present in the signal, the bandwidth of the control cycle, and the time constants of the control cycle, for example . The carrier ee can recover using two different methods in the system of Figure 1. One method uses both the modulated amplitude signals of the band edge from the outputs of the filters 20 and 22 in a manner that is similar to that previously described. for the recovery of time. The second method uses only one bandwidth of the received signal. In the second case, typically the band edge that is used is that which may contain the pilot. The extra energy associated with the pilot component improves the performance of the carrier recovery cycle in conditions of low signal to noise. However, it is noted that both methods conveniently do not require the presence of a pilot component. The carrier recovery method using both band edges, multiply the outputs of the filters 20 and 22 together in a multiplier 45 without conjugating the signal, as it did for the recovery of time. This multiplication produces a carrier modulated amplitude signal suppressed at the output of the multiplier 45 with a carrier frequency of -2fc. The symbol velocity component fst has been completely removed from this modulated amplitude signal. If there is a carrier phase shift (?), The carrier frequency is -2fc-2? as illustrated in Figure 6. At this point, the carrier recovery process is independent of the sampling clock frequency of the receiver demodulator, fsr. In digital signal processing applications, it is generally desirable to design controlled voltage oscillators (VCOs) or spectral exchangers that conveniently produce signals that are harmonically related to the clock frequency of the digital signal processor. In this regard, it is noted that the complex double-sideband modulated amplitude signal at the output of the multiplier 45 (Figure 6) is centered on a frequency of -2fc (neglecting any carrier delay), or on -2fc- 2? (including phase shifting, this means that this modulated amplitude signal is being mounted in the symbolic overlay.) More specifically, in practice, the left-sideband portion of the modulated amplitude signal shown in Figure 6 is actually "wrap around" in the positive frequency spectrum No symbolism occurs, because this modulated amplitude signal is a complex signal where the adjacent positive frequency component -'- of the first negative repeat band has been removed To simplify the design of the carrier recovery network, an associated phase detector 54 is essentially the same as the phase detector 28 used in the time recovery network 30. In order to accomplish this, the carrier of the Modulated amplitude input signal up to phase detector 54 should be boosted to 1/4 fsr The carrier of the amplitude-modulated signal input to the detector Phase 54 is nominally changed up to a frequency of 1/4 fsr by using a complex spectral changer that operates at +1/4 fsr. The spectral changer comprises complex multipliers 52 and 64. The multiplier 64 responds to a sampling signal of 1/4 fsr to change the output signal of the controlled voltage oscillator 62 by +1/4 fsr. The response of the controlled voltage oscillator 62 is similar to that of the controlled voltage oscillator 40 in the time control cycle, and responds to a direct current voltage produced by a low pass cycle filter 60 that is similar to the filter 38 in the time recovery cycle. A complex output signal resulting from the multiplier 64, which contains the frequency offset generated by the cycle plus the fixed 1/4 fsr, is applied to an input of the complex multiplier 52. The other input of the multiplier 52 receives the amplitude signal modulated centered at -2fc from the output of the complex multiplier 45. The phase detector 54 operates the same as the phase detector 28 in the time recovery cycle. The faece detector 54 includes a real / imaginary component eeparator 55, a translational network of the sign function 56, and an output multiplier 57. The phase detector 54 operates by multiplying the imaginary component of the output signal from the multiplier 52 with the sign of the real component. This causes the value of every third sample of the imaginary component to be driven to zero. Since the phase detector 54, as the faece detector 28, operates on the same series of samples (ie, nons or pairs, but not both), the cycle 60 filter (such as the filter 38) is required to supply output samples at the symbol speed rather than twice the symbol speed. This reduces in a significant way the complexity of the cycle filter, comparing with what would be required for implementations of two samples per symbol. The output of the multiplier 45, at the input to the carrier recovery network 50, is a modulated carrier-amplitude signal of complex double-sideband centered at a frequency -2fc-2 ?. The output of the controlled voltage oscillator 62 in the carrier cycle is a signal approximately equal to 2 ?. To produce this signal at the output of the controlled voltage oscillator 62, a 1/4 fsr signal is added to the carrier cycle by means of the multiplier 64 to cancel the component 2fc. The complex signal 2? at the output of the controlled voltage oscillator 62, is it moved to an output signal? of the carrier recovery network 50 by means of a frequency divider that divides between 2, 70. The output signal? is a tone (without a frequency spectrum) that represents phase / frequency phase shifting of the carrier. The vestigial sideband output signal from the analog-to-digital converter unit 16 is applied to an input of the multiplier 64 after being delayed by a unit 72, which compensates for the signal delay through the filters 20 and 22. The output of the delay unit 72 is a symmetrical side-band, side-band signal, close to the baseband, complex, as illustrated by the frequency spectrum diagram adjacent to block 72. This signal is changed to be closer to the base band by the multiplier 74, which is clocked at 1/8 fsr to produce a vertical sideband near the bae band at its output, as illustrated by the frequency spectral diagram in the output of the multiplier 74. The vestigial sideband signal near the base band from the output of the multiplier 74 is applied to an input of a complex multiplier 71, and the output signal (representative of the desfaeamiento)? from the output of the carrier recovery network 50 is applied to another input of the multiplier 71. The function of the multiplier 71 is to cancel the phase shift substantially? in the vestigial sideband signal, such that a vestigial band-sideband signal results. A complex demodulated vestigial sideband signal appearing at the output of multiplier 71 must be in baseband, and often it is. However, in practice, this signal may contain residual phase defects that may have to be compensated. This is done by a compensator 75, which can be of a known configuration. The compensator 75 compensates for the alterations of the channel as known, and produces a compensated output signal that is decoded by the unit 76, and is processed by an output processor 78. The decoder 76 may include, for example, the trellis decoder, the deinterleaver, the Reed-Solomon error correction, and the redeeming of audio / video decoders, as they are known. The output processor 78 may include the audio and video processors, and the audio and video reproduction devices. The recovery of the carrier can also be performed using a single band edge of the input signal, as follows. The lower bandwidth filter 22 produces a double-sideband modulated amplitude output signal with a carrier frequency of -fc + 1/4 fsl. This is done by setting the input to the multiplier 45 from the filter 20 to a value of the unit. This can be done by placing a multiplexer in the signal line between the output of the filter 20 and the upper input of the multiplier 45. One input of the multiplexer receives the output signal from the filter 20, and another input receives a signal of unit value. The last signal is conveyed to the input of the multiplier 45 in response to a control signal applied to a control input of the multiplexer. The output signal from the upper bandwidth filter 20 is decoupled from the multiplier 45 when the unit value signal is used. In a seventh that uses two band filters, the time and carrier insurance can be presented at approximately the same time. In a system that uses only a one-band filter, the carrier's insurance can occur after the timeout, depending on a variety of factors, such as noise, cycle gain, and bandwidth. of the cycle. If the time lock has been set by the network 30, then fst = 1/4 fsr, and the fst component of the carrier can be removed by changing the modulated amplitude output signal of the filter 22 with a spectral exchanger of -1 / 8 fsr. This spectral change is made in Figure 1 by changing the illustrated clock input of 1/4 fsr to the multiplier 64, in a clock of -1/8 fsr. After this spectral change, the signal carrier of amplitude modulated at the output of the multiplier 52 will be at a frequency -fc (in contrast to -2fc in the case of the double bandwidth method discussed above). In this mode, the carrier frequency of the modulated amplitude signal fc ee can easily impute to -1/4 fsr using a fae detector similar to the unit 54 used for the example of dual-band carrier recovery, and forcing every third mueetra of the real component to zero haeta. The output of the fae detector is integrated by a low-pass cycle filter that drives a voltage-controlled oscillator in a cycle that acts to drive the carrier frequency of the modulated amplitude signal up to -1/4 fsr of in a manner analogous to that described above in relation to the example of dual edge bank carrier recovery. In the described embodiments, the recovery of time (safe) can be achieved even in the presence of a carrier phase shift, and time protection is not supported by synchronization components in the vestigial sideband signal for this purpose. Even more, the recovery of the carrier is carried out without relying on a pilot component. The choice to operate phase detectors 28 and 54 at the midpoint of the Nyquiet region is a possible implementation. However, phase detectors could also be operated in the baseband with the same result. The frequency of the negative frequency spectrum is arbitrary. The positive spectrum could also have been used with analogous results, by using a different implementation of the circuits described. For example, the conjugate filters 20 and 22 would be used, and the input to the multiplier 64 would be -1/4 fsr.

Claims (10)

1. In a system for receiving a high definition television (HDTV) signal transmitted as a modulated vestigial side band (VSB) signal formatted as a constellation of data of a symbol dimension representing digital image data, and subject to displaying a carrier phase shift, a carrier recovery network (50) to change the received vestigial sideband signal to the baseband, this carrier recovery network being capable of achieving carrier recovery without relying on a pilot component, ei is present in that received signal, the carrier recovery network comprising: an input network (14, 16) for processing the vestigial sideband signal; a filter network (20, 22) that responds to an output signal from the input network, and that has a bandwidth response with respect to at least one of the upper and lower band edges of a spectrum of frequency of the vestigial sideband signal, to produce a double-sideband modulated amplitude (AM) signal at an output of the filter network; a phase detector network (54) for processing the double-sideband modulated amplitude output signal to produce a control signal (?) representing a phase shift of the carrier when present; an output processor (71, 74, 75) that responds to the control signal to produce a demodulated output signal at or near the baseband. A seventh according to claim 1, wherein: the filter network comprises a first bandwidth filter (20) and a second bandwidth filter (22) having bandwidth reepueetas respectively associated with the upper and lower band edges of a frequency spectrum of the veinigial sideband signal, to produce double-sideband modulated amplitude (AM) signals at the respective outputs of the first and second filters. A system according to claim 1, wherein: the filter network is a digital filter network comprising first and second filters with responses that are complementary to the frequency spectrum of the vestigial sideband signal received on the edges band of the first and second filters, respectively. 4. A seventh according to claim 1, wherein: the received vestigial sideband signal exhibits a frequency spectrum with band edge resp. At frequencies fc - 1/4 fst and fc + 1/4 fst, respectively, with respect to a medium band carrier frequency fc +/- ?; and the first filter exhibits a bandwidth response at a Nyquist frequency fc - 1/4 fst, and the second filter exhibits an e-band edge response at a Nyquist frequency fc - 1/4 fst, where: fc is the carrier frequency of the vestigial sideband signal transmitted; fst is the frequency of the transmitted symbols, • and? it is a shifting of the carrier when it is present. A system according to claim 2, and further comprising: a multiplier (45) having first and second inputs for receiving the output signals from the first and second filters, respectively, and an output coupled with an input of Fae detector network; wherein: an output signal of the multiplier is a signal of amplitude modulated of double sideband of suppressed carrier (Figure 6) centered around a frequency 2fc, where fc is the carrier frequency of the transmitted vestigial sideband signal, being the centered frequency subject to displaying a carrier phase shift (?). 6. A system according to claim 1, wherein: the input signal to the network of the phase detector is a complex signal having real and imaginary components; and the phase detector network includes elements (55, 56, 57) to cancel the imaginary component in an output signal from the phase detector network. A system according to claim 1, wherein the output processor comprises: a first translation network (74) that responds to an output signal from the input network to produce a vestigial sideband signal near the baseband; and a second translation network (71) that responds to the vestigial sideband signal near the baseband from the first translation network, and which responds to the control signal from the carrier recovery network, to produce a signal in or near the baseband. 8. A seventh according to claim 7, and further comprising: a compeneador (75) to remove the residual desfaeamientos of an output signal from the second brinjal network, to produce a vee-band sideband signal in the baseband . 9. A system according to claim 7, wherein: the first and second translation networks are signal multipliers; and the first translation network receives an input sideband signal in a double-sideband manner, and responds to a clock signal at a frequency related to fsr, where fsr is a sampling frequency of the receiver, to produce a vestigial sideband signal near the base band, of a single sideband, of exit. A system according to claim 1, wherein: the received vestigial sideband signal is an 8-VSB signal having a data constellation of a dimension defined by the following eight data symbols: -7 -5 - 3 -1 1 3 5 7