KR920002840B1 - Tv sound detection system - Google Patents

Abstract

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Description

TV voice detector

1 is a block diagram of a television receiver having a speech detection device constructed in accordance with the present invention;

2 is a television block diagram with another embodiment of a frequency converted PLL speech detection system constructed in accordance with the principles of the present invention.

3 is a block diagram of another embodiment of a frequency converted PLL speech detection device for the television receiver of FIG. 2 constructed in accordance with the principles of the present invention;

4 is a block diagram of a television receiver circuit, showing another embodiment of the present invention, including means for supplying an impulse noise indication signal.

* Explanation of symbols for main parts of the drawings

34, 36, 40, 42, 138: amplifier 36: second filter network

38: first filter network 40: 2F amplifier

44, 144, 234: Mixer 46: First 2F Amplifier

48, 152, 238: phase detector 50, 154, 252: oscillator

52, 156, 244: low pass filter 136: first filter

140, 148: buffer 142, 150: limiter

146: second filter

The present invention relates to television voice signal processing, in particular frequency modified phase locked loops (PLL) for detecting audio information.

Multichannel audio for stereo and bi-lingual broadcasts includes the use of one or more audio subcarriers, which increase the television audio signal bandwidth from 15KHz to nearly 90KHz or more. As a result, audio buzz that appears in voice signal processing channels tends to be more severe.

Audio buzz, which can be defined as a result of modulation associated with the image handed down to the audio signal, is always present to some extent in the television signal processing circuit, but is kept within acceptable limits by various circuit techniques.

In older televisions, video and audio signals were processed in a separate channel after the tuner circuit. This separation process prevents any significant interaction of the image and audio carriers in the receiver and removes the audio buzz generated in the receiver. Unfortunately, due to drift or auto-tuning (AFT), the frequency variation of the tuner local oscillator is transmitted to the intermediate frequency voice carrier as well as the IF picture carrier and spurious interference (buzz) by an audio frequency modulation (FM) detector. Is detected as. In addition, since the audio channel also has a much narrower pass band than the picture channel pass band, the sound on the receiver should be best tuned, but the picture does not.

Today, television receivers are almost universally using the intercarrier method of speech signal processing. In such a method, (1) picture and voice carriers are processed after the common IF channel tuner includes a special IF band pass response that attenuates the voice carriers by approximately 20 db than the picture carrier and places the picture carrier 6 db lower on the high frequency slope of the IF pass band. do. Thereafter, a large amplitude image broadcast wave is processed in the video channel to detect video information and reproduce audio information, and the two IF carriers are mixed in the form of an inter-carrier speech signal having a frequency corresponding to the difference of the carrier IF frequencies. That is, for an NTSC system, a 45.75 MHz picture carrier is mixed with a 41.25 MHz voice carrier to obtain a 4.5 MHz intercarrier voice signal. The intercarrier detection method uses a common mode FM (e.g., fluctuations of the tuner local oscillator in the accessory of the television with which the receiver itself and the receiver are coupled, such as a cable TV converter) in voice and image carriers to generate an intercarrier speech signal. This is a particular advantage due to the fact that it is suppressed by IF signal mixing. However, the intercarrier method does not eliminate buzz. Although the intercarrier IF passband response recorded above is required for proper detection of the picture carrier, its use reduces the signal-to-noise ratio of the audio signal by attenuating the voice carrier and distorts the voice signal during intercarrier mixing. Compared to discrete picture and voice channel methods that cause dissimilar IF picture carrier sideband attenuation, which delivers granular phase modulation (ICPM) to an image carrier, commonly referred to as “Nyquist ICPM,” there is a tendency to significantly increase audio buzz. have.

Thus, the high degree of modulation and light modulation (commonly caused by localized network station insertion by leveling the image) of the image carrier in an intercarrier system is the image carrier (the signal can be removed and the video line and field ratio (i.e. 15,734 in NTSC devices). Hz and 60 Hz) The inter-carrier buzz problem is a paper published by P. Pockens and C. G. Illers, published in the IEEE pictorial of Conschemo Electronics in August 1981. This is discussed in more detail on pages 381 to 397 of Intercarrier Buzz Analysis and Countermeasures.

Although audio buzz produced by existing intercarrier systems may be allowed in mono audio TV receivers due to the relatively folded pass band of the voice channel in such a receiver. Unfortunately, audio buzz with increased audio bandwidth of multiple channels is also increased and no longer in tolerance.

Voice detection systems have been improved by combining separate and inter-carrier systems and can be described as 'filt-inter carrier'. Such a system is described in a paper previously described by P. Pockens and C. G. Illers. In this system, following the tuner, the picture carrier signal is processed separately from the voice carrier signal to derive the video information. However, IF picture and voice carriers are processed together in a voice channel for detecting audio information according to the intercarrier system described above. In this system, the audio filter buzz due to Nyquist ICPM is almost eliminated because the IF filter response needed to detect residual sideband video signals is not included in the voice channel. However, as the intercarrier method is maintained, the IF picture and the voice carrier are processed jointly in the IF amplification channel. As such, the distortion and video modulation associated with the image of the audio signal generated by the nonlinear effects in the IF amplifier are the cause of two audio buzz unique to an intercarrier system that is not removed. Furthermore, it requires additional circuitry, tuning circuitry, and identifiers, such as intercarrier amplifiers.

The present invention relates to an audio signal detection system that combines the advantages of a separation and intercarrier method that eliminates individual drawbacks and minimizes circuit complexity. According to the present invention, a system for reproducing audio information from a television signal of an intermediate frequency (IF) picture and an audio carrier signal comprises a controlled frequency output signal of a frequency shifted phase locked loop (PLL) comprising a frequency mixer and a phase detector. The branch has a controllable oscillator. The mixer has an oscillator output signal as one input and an IF image carrier signal as the other input. The phase detector has an IF audio signal as one input and an output from the mixer as another input. A low pass filter coupled between the output of the phase detector and the control input of the oscillator completes the PLL by removing the control signal to the oscillator to control the output signal frequency. The detected audio information is supplied at the output of the low pass filter. In the present phase locked loop system, it is convenient to have a device for detecting a television audio signal and a device for indicating the presence of impulse noise in a video signal.

Referring to FIG. 1, the television broadcast signal received by the antenna 8 is applied to a television tuner 10 including a radio frequency (RF) amplifier 12, a mixer 14 and a local oscillator 16. Tuner 10 selectively converts RF picture and audio carrier signals of television channels that are special to intermediate frequency (IF) carriers at 45.75 MHz and 41.25 MHz, respectively, in NTSC devices. The IF carrier is useful at tuner output terminal 918. A 45.75 MHz IF picture carrier is basically an amplitude modulated signal that contains composite video information. In other words, the 41.25 MHz voice carrier is a frequency modulated signal. Conventional color television signal processing circuits, including IF filtering amplification channel 20, video detection 22 and video signal processor 24, are red (R), green (G) at kinescopes (not shown). And respond to an IF picture carrier at terminal 18 to supply a blue (B) color video signal and to produce a color image of the water scene. In addition, the tuning limiter amplifier stage 26 supplies a variation in which the IF image carrier signal is appropriately filtered, amplified, and limited to the video detector 22 for synchronous detection of the composite video signal from the IF image carrier. The IF image carrier output of the limiter / amplifier 26 is automatically untuned via a 90 ° phase mobile network 30 to supply an AFT control signal at the output of the AFT detector 28 supplied to the local oscillator 16. AFT) is applied to the detector 28. In this method, the frequency and phase of the local oscillator signal applied to the mixer 14 follow the frequency and phase of the received RF image carrier. The construction and operation of the video detector 22, the limiter and amplifier 26, the AFT detector 28 and the phase shift network 30 was confirmed by W. April 21, 1981. G. This is described in more detail in US Pat. No. 4,263,611 published by Gibson and his co-inventors.

IF channel 20 has IF picture carrier P positioned as low as 6 db on the higher frequency slope of passband response 32 and IF voice carrier S as low as 25 db on the lower frequency slope of passband response 32. Supply a conventional passband response 32 located. As such, residual sideband video information can be detected from the IF speech carrier without sufficient interference. Unfortunately, the IF passband response 32 causes accidental carrier phase modulation (ICPM), called “niquist”, for the IF picture carrier due to the asymmetric attenuation of the IF picture carrier sideband. In a conventional intercarrier voice detection system, Nyquist induced by ICPM distortion of an IF picture carrier results in distortion of the intercarrier voice signal due to the process of mixing the IF voice and the picture carrier. This distortion of the intercarrier speech signal occurs in the audio buzz. The amount of audio buzz generated by the Nyquist ICPM increases in direct proportion to the increase in the band of the audio signal. As such, for example, the monophonic and stereophonic and second audio program signals are much larger than the bandwidth of the monophonic signal generally processed by the inter-carrier system. In view of the bandwidth, it becomes clear that the intercarrier method of audio detection is likely to be unsatisfactory because the audio buzz generation is almost increased.

According to the present invention, the IF audio carrier is processed in a passage separated from the IF image carrier processing circuit, and the composite audio signal is detected by a frequency shifted phase locked loop (PLL) 53. Thus, in order to detect audio information, IF picture and voice carriers provided at the terminal 18 are applied to the picture and voice band pass filters 36 and 38, respectively, via the buffer amplifier 34 and the terminal A. FIG. do. The picture band pass filter 36 is symmetrical and provides a relatively narrow (1 MHz) pass band response 37 symmetrically focused on the IF picture carrier frequency (e.g. 45.75 MHz in NTSC devices) to select only IF picture carrier signals. ) IF amplifier 40 and limiter 42 moderately amplify and limit the IF picture carrier signal, apply a limited result, thereby hardly modulating the IF picture carrier for one input of frequency mixer 44. .

Voice band pass filter 38 is a symmetric input of phase detector 48, which is moderately amplified by IF amplifier 46, and then selects the IF voice carrier frequency (nearly) to almost select only the applied IF voice carrier signal. For example, it has a relatively narrow (eg 1 MHz) passband 39 centered at 41.25 MHz in an NTSC device. The IF amplifiers 40 and 46 are similarly constructed and each have an integrated circuit IF amplifier, such as TA7607 manufactured by Doco Shibaura Electric Corporation Limited. The limiter 42 simply has the opposite poled Schottky barrier diode connected in parallel. Their output signals are responsive to the output of the IF amplifier 46 (in other words, the output of the amplifier 40) for supplying the AGC control voltage to the IF amplifiers 40 and 46 for gain control to set to a predetermined level.

A varactor tuned voltage controlled oscillator (VCO) 50 having a normal oscillation frequency equal to the frequency difference between the IF picture and audio carriers (e.g., 4.5 MHz in NTSC devices) supplies a second input to the mixer 44. . The mixer 44 has a dual balanced multiplier circuit, such as the commercially available MC1496 manufactured by Motorola Semiconductor Products, Inc., which mixes the IF image carrier signal and the VCO output signal and provides a frequency and phase difference between its input signal. It acts as a switching mode in response to an amplitude limited IF picture carrier (which acts as a switching control signal) to supply its representative signal to its output. The frequency difference between the input signals of mixer 44 is 41.25 MHz. This 41.25 MHz output signal of the mixer 44 is supplied to the phase detector 48 as a second input, which also has an MC1496 integrated circuit. Phase detector 48 supplies an output signal having a varying amplitude with respect to the integration of the phase difference of its input signal and is frequency modulated with respect to the FM modulated IF voice carrier coupled to its first input from amplifier 46. It operates as a (FM) type demodulator and provides a composite baseband audio signal (according to the undesirable signal from the detection process) to its output. For a more detailed explanation of the operation of analog doubling as an FM demodulator, see A. Virotti's Application of the Momolithic Analog Multiplier, pages 373-380, in the IEEE Newsletter of the December 1968 Solid State Circuit. It is described in.

The low pass filter 52 connected to the output of the phase detector 48 filters its output signal to control the frequency of the VCO 50. Mixer 44, phase detector 48, VCO 50 and low pass filter 52 have a frequency shifted phase locked loop 53. The low pass filter 52 has sufficient bandwidth to select a composite baseband audio signal (to reject undesirable signals of significantly higher frequency) and to feed the composite audio signal to, for example, the stereo detector 54. If stereo programming is supplied to the composite audio signal, decoder 54 decodes the composite audio signal and supplies left and right stereo signals to speakers 56 and 58, respectively. If stereo programming is not supplied, decoder 54 supplies only the monophonic signal to the speaker.

In operation, mixer 44 frequency converts the IF picture carrier signal applied to its input and supplies an output signal with a frequency corresponding to the frequency difference (eg 41.25 MHz) of its input signal. The sum of the undesirable output signal and the input signal corresponding to the feedback amount of the input signal is attenuated by the relatively narrow bandwidth of the low pass filter 52. The output voltage amplitude of the phase detector 48 is the phase difference between its input signal. This output voltage is filtered by low pass filter 52 and applied to VCO 50 as a control voltage. The output frequency of the VCO 50 and the 41.25 MHz modified signal from the mixer 44 are directed to change in direct relation with the amplitude of the control voltage and to reduce the phase difference between the signals at the input of the phase detector 48. When the phase of the input signal of the detector 48 is 90 degrees (right angle), the minimum amplitude control voltage is applied to the VCO 50. Therefore, because of the feedback component of the loop 53, when it is closed, the control voltage at the output of the filter 52 corresponds to the audio information, and the frequency of the modified difference signal at the output of the mixer 44 is IF It is equal to the average frequency of the voice carrier and its phase is perpendicular to the phase of the IF voice carrier. As such, the phase detector 48 acts as an FM detector for frequencies that demodulates IF speech carriers. Thus, the common mode FM delivered to both the image and audio carriers by a tuner local oscillator 16 or a television accessory, such as a cable television converter, prior to tuner 10, is canceled in phase detector 48 by passing it on a modified difference signal. do.

The present voice detection system is advantageous because it uses a predetermined frequency interval between the IF picture and the voice carrier to allow for simplified receiver tuning, and any common mode FM of the IF carrier is canceled out. Moreover, the Nyquist ICPM of the image carrier due to the top slope of the IF pass band does not generate audio buzz in the present detection device. More particularly, the image and audio processing channels are immediately separated after the tuner 10. In the speech processing channel, the image band pass filter 36 symmetrically attenuates the sidebands of the IF image carrier, thereby removing the “niquist” form of the audio buzz. Moreover, the band pass filter response 37 is a function of the video signal present at frequencies corresponding to the second and third harmonics of 4.5 MHz (e.g., 2.25 MHz and 1.5 MHz, respectively, from a 45.75 MHz picture carrier in an NTSC system). It is a narrow band that almost attenuates the component. Since the frequency of the output signal of the VCO 50 is also at 4.5 MHz, audio frequency zero beats are prevented from occurring at the mixer 44 output. As a result, other sources of audio buzz associated with the picture inherent in the present intercarrier detector system are actually removed.

Thus, compared to an intercarrier device in which the IF channel passband is attenuated by approximately 25 db to the IF voice carrier, the signal-to-noise in the audio channel has the opposite effect in the system of the present invention, and the acoustic bandpass filter 38 Provides the IF voice carrier to the IF amplifier 46 which is relatively attenuated. As such, the amplitude level of the voice carrier is nearly 25db larger than in an intercarrier system and thus the signal-to-noise level of the detected audio signal is improved for a given RF signal level.

It should also be noted that since no intercarrier speech signal is generated, no intercarrier amplifier, intercarrier discriminator or tuning circuit associated therewith is required.

It should be noted that the amplitude levels of the input signals to mixer 44 and phase detector 48 are their input devices first responding to the frequency and phase of the input signal. In this way, a unique AM wave of the IF image carrier modulation is obtained. Moreover, even though the sum frequency signal (i.e. 82.5 MHz) is supplied at the phase detector 48 output, its modulation envelope is nearly symmetrical and has insufficient effects at the audio baseband frequency. Therefore, the limiting action of the limiter 42 need not be as large as that required for the intercarrier system. The smaller limit is also reduced. That is, 1) the audio bead field ratio generated by the possibility of image related lines and the high modulation degree of the IF image carrier, 2) the generation of harmonics mixed with the output of the VCO 50, and the distortion of the audio frequency band.

In another embodiment, the IF picture carrier at the output of the restrictor and amplifier 26 is combined with the mixer 44 at point B and thereby used as its switching signal input. The circuit between points A and B can be eliminated. However, since the IF amplification channel 20 is included in the voice detection system, the Nyquist induced ICPM is not eliminated and this embodiment is not suitable for detecting wide bandwidth composite audio signals. However, IF speech carriers produce relatively little attenuation and can maintain an improvement in the actual signal to noise when detecting relatively narrow bandwidth audio signals, such as, for example, monophonic audio signals.

Separate IF amplifiers 40 and 46 are used in the embodiment shown in FIG. 1 while a common IF amplifier is used to amplify the IF picture and audio carrier signals supplied to the output of buffer 34, which will be described next. .

Referring to FIG. 2, a television carrier signal received by the antenna 108 is applied to a television tuner 110 comprising a radio frequency (RF) amplifier 112, a mixer 114 and a local oscillator 116. . The tuner 110 selectively converts the RF picture and audio carrier signals of the selected television channel for intermediate frequency (IF) carriers, which are 45.75 MHz and 41.25 MHz, respectively, in an NTSC device. An IF carrier is available for tuner output terminal 118. A 45.75 MHz picture carrier is a basic amplitude modulated (AM) signal that contains composite video information. On the other hand, the 41.25 MHz IF voice carrier is a frequency modulated (FM) signal. Conventional color television signal processing circuits, including IF filtering and amplifying channels 120, video detectors 122, and video signal processors 124, are kinescopes (not shown) to reproduce color images of broadcast scenes. (R), green (G), and blue (B) color video signal terminals 118 to respond to the IF picture carrier. Thus, the tuned limiter and amplifier stage 126 provides a moderately filtered, amplified and limited IF image carrier modified signal to the video detector 122 for synchronous detection of the composite video signal from the IF image carrier. The IF image carrier output of the limiter and amplifier 126 is automatically untuned through a 90 ° phase mobile network 130 to supply an AFT control signal at the output of the AFT detector 128 applied to the local oscillator 116. AFT) detector is also applied. In this method the frequency and phase of the local oscillator signal applied to the mixer 114 vary with the frequency and phase of the received RF image carrier. The configuration and operation of the video detector 122, the limiter and the amplifier 126, the AFT detector 128, and the phase shift network 130 are W. It is described in more detail in U.S. Patent No. 4,263,611 issued April 21, 1981 by G. Gibson and other co-inventors.

IF channel 120 has IF picture carrier P positioned as low as 6 db on the significantly higher frequency slope of passband response 132 and IF voice carrier S is nearly 25 db on the very low frequency slope of passband response 132. It has a conventional passband response 132 located low. As such, residual sideband video information can be detected from the IF speech carrier without significant crosstalk. As explained, the IF passband response characteristic 132 induces an accidental “Nyquist” carrier phase modulation (ICPM) for the IF picture carrier due to the asymmetric attenuation of the IF picture carrier sideband.

In the present speech detection apparatus, the IF speech carrier is processed in a passage separated from the IF image carrier processing circuit and the composite audio signal is detected by the frequency shift phase locked loop (PLL) 157 of FIG. According to the present invention, suitable amplification of the IF picture and voice carriers suitable for use in the PLL 157 is provided by a common IF amplifier. In particular, the IF picture and voice carriers supplied at terminal 118 are applied to a single IF amplifier 138 via buffer amplifier 134 and filter 136. The filter 136 is a relatively narrow (e.g., 3 db bandwidth of 1 MHz) passband portion that is nearly symmetrical and concentrated at the IF image carrier frequency P for selecting and passing the image and audio carriers to the IF amplifier 138 without relatively attenuating them. 137a), and have a double vein (two peak) bandpass frequency response that includes a relatively narrow (e.g., 3 db bandwidth of 1 MHz) passband portion 137b that is nearly symmetrical and focused on the IF voice carrier frequency. The filter 136 maintains the relative size of the image and audio carriers and has a generally designed identification element tuning circuit or surface acoustic wave filter, such as FI322 manufactured by Toko Shiba Yura Electric Corporation Limited. IF amplifier 138 includes an automatic gain control circuit for gain control of amplifier 138 to linearly amplify the image and audio carriers and set its output signal to a predetermined level.

From the output of the IF amplifier 138, the buffer amplifier 140 and the limiter 142 apply an amplitude limited and thus almost unmodulated image carrier to the input terminal 143 of the mixer 144. Filter 146, buffer amplifier 148, and limiter 150 apply almost IF voice carriers to input terminal 151 of phase detector 152 only. Limiters 142 and 150 are simply provided with Schottky barrier diodes connected in parallel and vice versa. The filter 146 has an identification element trap circuit having a tuned center frequency to provide a frequency response 147 having a maximum attenuation at the IF picture carrier frequency P and a minimum attenuation at the voice carrier frequency S.

A varactor tuned voltage controlled oscillator (VCO) 154 having a frequency difference between the IF image and voice carrier, i.e., 4.5 MHz for NTSC devices, at terminal 145, mixer 144 at Motorola Semiconductor Products It has a dual balanced analog multiplier circuit, such as MC1496 manufactured by Incorporated, which operates in a switching mode responsive to an amplitude limited IF picture carrier from limiter 142 (which acts as a switching control signal). The mixer 144 mixes the IF picture carrier and the VCO output signal and supplies at its output a frequency converted signal representing the frequency and phase difference between its input signal. The frequency difference between the input signals of mixer 144 is 41.25 MHz. The 41.25 MHz output signal of the mixer 144 is supplied to the second input terminal 153 of the phase detector 152, which also includes the MC1496 integrated circuit. Phase detector 152 supplies an output signal with varying amplitude directly related to the phase difference of its input signal and operates as a frequency modulated (FM) type demodulator for its applied FM modulated IF voice carrier and complex at its output. Supply a baseband audio signal (according to the undesirable signal resulting from the detection process).

The low pass filter 156 connected to the output of the phase detector 152 filters the output signal of the phase detector 152 and supplies a control signal for controlling the frequency of the VCO 154. Mixer 144, phase detector 152, VCO 154 and low pass filter 156 include a frequency shifted phase locked loop 157. The low pass filter 156 has a sufficient bandwidth to select a composite baseband audio signal (while rejecting an undesirably high frequency undesirable signal) and supplies the composite audio to the stereo decoder 158. If stereo programming is provided as a composite audio signal, decoder 158 decodes the composite audio signal and supplies left and right stereo signals to speakers 160 and 162, respectively. If stereo programming is not supplied, the decoder 158 supplies only a monophonic signal to the speaker.

In operation, the limiter 142 responds to the highest amplitude signal applied to its input. In television broadcasting and cable television distribution devices, the transmitted audio carrier is attenuated by at least 8 db, depending on the amplitude of the image carrier. The filters 136 maintain their magnitudes so that the limiter 142 operates to select only the image carriers from the output of the amplifier 138 and applies them to the mixer 144 with little modulation. In other words, a band pass filter or trap type filter circuit may be used instead of the limiter 142. However, they are difficult to embed easily in integrated circuit chips, for example diode limiter circuits, and do not remove amplitude modulated video information.

As described above, the mixer 144 frequency converts the IF image carrier applied to its input and the frequency difference of its input signal (e.g. 41.25 MHz) to the input terminal 153 of the phase detector 152. Supply an output signal with a corresponding frequency. Unnecessary output signals corresponding to the feedback of the sum of the input signal and the input signal frequency are also supplied to the output of the mixer 144, so that these signals are supplied to the decoder 158 by a relatively narrow bandwidth of the low pass filter 156. Being reached is prevented.

The filter 146 supplies attenuation of the image carrier with respect to the voice carrier, and the limiter 150 supplies almost only the voice carrier to the input terminal 151 of the phase detector 152 in response to the larger amplitude voice carrier. The amplitude of the output voltage of the phase detector 152 is equal to the phase difference between those input signals. This output voltage is filtered and supplied by the low pass filter 156 as a control voltage to the VCO 154. The output frequency of the VCO 154, and consequently the 41.25 MHz converted signal from the mixer 144, changes directly in relation to the amount of control voltage, and the phase difference between the signals at the input of the phase detector 152 decreases in a direction to decrease. It is. When the input signal phase of the detector 152 is at right angles (90 °), the minimum amplitude control voltage is applied to the VCO 154. Due to the feedback nature of the loop 157, the control voltage at the output of the low pass filter 156 corresponds to the audio information when it is cropped, and the frequency of the converted difference signal at the output of the mixer 144 is the IF voice carrier. It is configured to be equal to the average frequency of and its phase is perpendicular to the phase of the IF speech carrier. Thus, phase detector 152 operates as FM detection for the frequency that demodulates the IF speech carrier.

Prior to the tuner 110, the phase detector 152 is canceled by passing some commonly FM to the converted difference signal, which is delivered to the image and audio carriers by a tuner local oscillator 116 or TV accessory, such as a cable TV converter.

Since a single IF amplifier is used to provide a picture and audio carrier suitably amplified by the frequency modified PLL 157, the complexity and cost of the circuit is reduced compared to the previously described embodiment.

The amplitude levels of the input signals to mixer 144 and phase detector 148 are configured such that their input devices first respond to the frequency and phase of the input signal and not to their amplitude.

Thus, AM rejection inherent in IF image carrier modulation is obtained.

Moreover, although the sum frequency signal (i.e. 82.5 MHz) will be provided at the output of the phase detector 152, it has an insufficient effect at the audio baseband frequency because its modulation envelope is nearly symmetrical. Therefore, the limiting operation of limiter 142 need not be as large as required for the intercarrier system. The less restrictive is the possibility of the presence of line and field ratio audio buzz associated with the picture caused by the high degree of modulation of the IF picture carrier and the harmonics that cause distortion in the audio frequency band by mixing with the output of the VCO 154. Reduced by occurrence.

The speech detection device from the IF amplifier 138 described above to the decoder 158 (including the filter 146) can be advantageously configured on a single integrated circuit. Limiters 142 and 150 are composed of diode circuits. Filters 136 and 146 can be constructed using relatively inexpensive tuned circuits because the amount of attenuation they have to supply is less. However, in another embodiment, to compensate for the additional signal rejection previously supplied by the limiter 150, the limiter 150 may be deleted, and the filter 146 provides a corresponding increase attenuation at the picture carrier frequency P. must do it.

This can easily be done by adjusting its tuned circuitry. Moreover, filter 146 consists of a relatively narrow (ie 500 KHz) band pass filter with a frequency response 164 focused on voice carrier frequency S and provides actual attenuation at picture carrier frequency P.

Other embodiments described above describe various techniques for attenuating the image carrier and preventing it from reaching the input terminal 151 of the input terminal 152 of the phase detector 152. If the image carrier is not properly attenuated at the input terminal 151, the feedback of the image carrier from the output of the mixer 144 to the input terminal of the detector 152 is mixed with the image carrier at the input 151 and is within the audio frequency range. The audio information is distorted by supplying a zero beat output signal.

FIG. 3 shows another embodiment of the television receiver sound detection system of FIG. Elements having the same configuration and operation have the same reference numerals of the elements corresponding to FIG. The filter 166 is provided between the output of the mixer 144 and the input terminal 153 of the phase detector 152 to substantially attenuate the image carrier and pass the audio carrier.

As such, no actual means for supplying a voice carrier to the input terminal 151 of the phase detector 152, such as the filter 146 and / or the limiter 150, is required. The filter 166 is of a trap circuit type having a frequency response characteristic 167 that supplies the maximum signal cancellation at the image carrier frequency P and the minimum attenuation at the voice carrier frequency S. In other words, the filter 166 may be configured with a bandpass that provides a substantial attenuation at the image carrier frequency P with a relatively narrow (i.e. 500 KHz) passband frequency response characteristic 168 with a center frequency at the voice carrier frequency S. have.

In the embodiments described above, the IF picture carrier supplied at the output of the television tuner is treated with a separate IF channel for generating video information. Audio information is detected by a frequency shifted phase locked loop (PLL) comprising a mixer and a phase detector respectively responsive to the image and audio carriers supplied at the output of the tuner.

According to another embodiment of the present invention, the system has a single phase locked loop circuit that detects video signal noise defects and recovers audio information from a television signal, which can be easily configured as an integrated circuit.

Referring to FIG. 4, the television broadcast signal received by the antenna 208 is applied to the tuner 210. As shown in FIG. Tuner 210 selectively converts radio frequency (RF) picture and voice carriers of TV channels selected at their respective intermediate frequencies, namely 45.75 MHz and 41.25 MHz, in an NTSC system. The IF carrier is available at the tuner output terminal 212. An IF picture carrier is basically an amplitude modulated (AM) signal that contains composite video information.

Meanwhile, the IF voice carrier is a frequency modulated (FM) signal. Meanwhile, a conventional color television signal processing circuit comprising an IF filtering and amplifying channel 214 and a video detector 216 responds to an IF picture carrier at terminal 212 for detecting a composite baseband video signal. The baseband video signal is applied to an input terminal 217 of a comb filter that distinguishes chromaticity (C) and luminance (Y) from the composite baseband video signal and supplies respective signals to terminals 219 and 220. . A more detailed description of comb filter 218 will be described later. Luminance and chromaticity signals are processed in order to generate a color signal supplied to each gun of the kinescope (not shown) in a conventional manner. In the conventional method, in order to display an image on the screen of the kinescope, the resultant electron beam is continuously flattened on the horizontal image line.

IF channel 214 has IF image carrier P positioned as low as 6 db on the higher frequency slope of passband response characteristic 221 and IF audio carrier S as nearly 25 db on the lower frequency slope of passband 221. Low passband response 221 characteristics. As a result, residual sideband video information can be detected from the IF voice carrier without significant interference. As described, the IF passband response characteristic 221 induces "Nyquist", which is the incident carrier phase modulation (ICPM) of the IF image carrier, due to asymmetric attenuation of the IF image carrier sidebands.

In this receiver, the frequency conversion phase lock loop (PLL) 222 substantially reduces the Nyquist induced ICPM to process the IF speech carrier in a path distinct from the IF image carrier processing circuit for audio information detection. From the buffer amplifier 224 terminal 212 to the image and audio band pass filters 226 and 228 combine the IF image and audio carriers, respectively. The image band pass filter 226 is symmetrical and has a relatively narrow (i.e., 3 db bandwidth of 1 MHz) pass band response characteristic concentrated at an IF image carrier frequency P (i.e. 45.75 MHz in NTSC devices) that passes almost IF image carrier signals only. 227). IF amplifier 230 and limiter 232 moderately amplify and limit the IF picture carrier and finally apply to one input of a limited and nearly unmodified IF picture carrier signal mixer 234.

Voice passband filter 228 is symmetrical and is a fairly narrow (i.e., 3 db bandwidth of 1 MHz) passband concentrated at an IF voice carrier frequency S (i.e. 41.25 MHz in NTSC systems) for processing only IF voice carrier signals. Direct sidebands are applied as input to one of the phase detectors 238. IF amplifiers 230 and 236 can be easily configured and each has an integrated circuit IF amplifier such as TA7607 manufactured by Tokyo Shibaura Electric Corporation Limited. Limiter 232 is simply provided with an opposite pole Schottky barrier diode connected in parallel. The conventional automatic gain control circuit (AGC) 240 supplies an AGC control voltage to the IF amplifiers 230 and 236 and controls the gain to control gain so as to set their output signal to a predetermined level. That is to say, to the output of amplifier 230).

A varactor tuned voltage controlled oscillator (VCO) 242 having a normal oscillation frequency equal to the frequency difference between the IF image and audio carriers (e.g., 4.5 MHz in NTSC systems) supplies a second input to the mixer 234. do. Mixing 234 consists of a dual balanced analog multiplier circuit such as, for example, MC1496 made by Motorola Semiconductor Products, Inc., which outputs a frequency-converted signal representing its frequency and phase between its input signal and its output. It operates as a switching mode in response to an amplitude limited IF picture carrier by mixing the IF picture carrier signal and the VCO output signal to supply to the. The frequency difference between the input signals of mixer 234 in an NTSC device is 41.25 MHz. The 41.25 MHz output signal of the mixer 234 is supplied as a second input to the phase detector 238 provided with the MC1496 integrated circuit. Phase detector 238 supplies an output signal having a varying amplitude directly related to the frequency difference of its input signal to provide a frequency for the FM modulated IF speech signal coupled with its first input from amplifier 236. It operates as a modulated (FM) type demodulator and supplies a composite baseband audio signal (an undesirable signal resulting from the detection process) to its output.

The low pass filter 244 coupled with the output of the phase detector 238 filters its output signal and supplies a control signal for frequency control of the VCO 242. Mixer 234, phase detector 238, VCO 242, and lowpass filter 244 form a loop 22 with frequency shifted phase. The low pass filter 244 has a bandwidth sufficient to select a composite baseband audio signal, but has a bandwidth narrow enough to reject higher frequencies for the undesirable signal and supplies the composite audio information signal to the stereo decoder 246.

If stereo programming is provided as a composite audio signal, decoder 246 decodes the composite audio signal and provides left and right stereo signals to speakers 248 and 250, respectively. If stereo programming is not provided, the decoder 246 will only provide a monophonic signal to the speaker.

In operation, the function of mixer 234 frequency converts the IF picture carrier signal applied to its input and provides an output signal with a frequency corresponding to the frequency difference (eg 41.25 MHz) of its input signal. The undesirable output signal corresponding to the feedback of the sum of the input signal and the input signal is attenuated by the relatively narrow bandwidth of the low pass filter 244. The output voltage amplitude of the phase detector 238 is equal to the phase difference between its input signals. This output voltage is filtered and amplified by the low pass filter 244 as a control voltage to the VCO 242. The output signal frequency of the VCO 242 and consequently the 41.25 MHz converted signal appearing from the mixer 234 are in a direction in which the amplitude of the control voltage changes in direct relation and reduces the phase difference between the signals at the input of the phase detector 238. .

When the input signal phase of the detector 238 is at right angles (90 °), the minimum amplitude control voltage is applied to the VCO 242. Therefore, due to the feedback nature of the loop 222, when it is closed, the control voltage at the output of the filter 244 is such that the frequency of the difference signal converted at the output of the mixer 234 is equal to the average frequency of the IF speech carrier and its phase. It is perpendicular to the phase of this IF audio carrier. As such, phase detector 238 acts as an FM detection for demodulating the IF speech carrier to frequency to generate a signal that, when filtered by filter 244, corresponds to audio information.

When the mixer 234 responds to the picture carrier signal, any common mode FM delivered to the picture and audio carrier by a television accessory or local oscillator, such as a cable television converter, prior to the tuner 210, is frequencyd by the mixer 234. The phase detector 238 cancels out the transmission of the converted signal.

As described above, the frequency-converted speech carrier is selected from the output of tuner 210 by bandpass filter 228 and amplified by amplifier 236. The amplifier 236 controlled by the AGC circuit 240 tends to adjust the voice carrier to have a constant long duration average amplitude and to remove amplitude waves that normally appear unrelated to noise. The time of the AGC device is not fast enough to limit the appearance of amplitude waves due to impulse noise.

As such, except for amplitude waves induced by impulse noise, the amplitude of the modulated speech carrier is nearly unchanged. Since the impulse noise wave of the PLL is relatively insensitive to amplitude waves, the limiting does not require passing between the amplifier 236 and the detector 238 to eliminate the impulse noise wave.

As a result, the output of amplifier 236 is advantageous for detecting amplitude waves of modulated voice carriers, driving coupling control signals, and reducing video signal defects induced by impulse noise. In addition, because the bandwidth of the filter 228 is wide enough to pass the voice carrier and some of its sidebands, it appears to be a relatively shorter signal delay than the delay in the video signal path and allows sufficient time for video signal compensation. . The combined control signal generation circuit 252 including the synchronous detector 254 to detect the wave induced by the impulse noise can be advantageously embedded in a predesigned structure with a minimum configuration in the required integrated circuit and the audio detector It can be easily built by the structure method of PLL.

Since the output signal of the mixer 234 has the same frequency as the average frequency of the frequency-modulated voice carrier but has a quadrature phase which is a 90 ° phase shift network 256, the output signal of the mixer 234 is shifted and synchronized. It is applied to one input of sex detector 254. As such, when an FM voice carrier is applied to the other input of the detector 254, the phases of its input signals are in synchronism with each other (i.e., in a phased array) and in a well known manner such a sync detector 254 can be Detect amplitude variations in voice carriers. Detector 254 also has a dual balanced multiplier such as can be found in the MC1496 integrated circuit described above.

The amplitude variation seen at the output of the detector 254 is applied to the first input terminal of the comparator circuit 258. The DC threshold signal from the power source 260 is applied to the second input of the comparator 258. Each time the output signal of detector 254 passes the threshold level, the comparator results in a change in output state. The output signal of the detector 254 will be a slowly varying AC signal with a relatively small amplitude, except when impulse noise or other dramatic signal disturbances occur, in which case the output signal is the RF circuitry and noise of the tuner 210. Other species associated with impulse interactions have relatively large and rapid output signal changes that are negative or positive. As such, it is required to connect the threshold signal of the opposite polarity to the second comparator circuit in order to detect deviation of the sync detector output signal of one polarity. Furthermore, the two comparators are logically interconnected and represent as output signals to inform the response characteristic correlated with the impulse noise ringing characteristic.

The output signal of the comparator is regulated by the pulse generator 262 and through the terminal 263 to an enable and disable switch 264 for influencing video signal correspondence in the comb filter 218. Is authorized. The pulse generator 262 may be a monostable multivibrator that generates pulses of defined amplitude and persistence (typically in the range of 1-10 microseconds). The test data shows that the impulse noise duration is typically in the 1-3 microsecond range and the 3 microsecond control pulse is sufficient to compensate for most noise impulses. On the other hand, it is required to end the video correspondence at a specific time interval during the horizontal retrace period. In this case, the pulse generator is an edge trigger flip-flop that is triggered (set) by a comparator output signal, such as a horizontal synchronization pulse, and off (reset) by a separation time pulse.

The comb filter 218 is experimented with a charge transfer technique and is integrated with the CCD sum portion, the sample and hold (S / H) circuit, and the 1-H (limit horizontal image line) delay line to interlaced luminance (Y) and chroma signal. The comb filter 218 performs the filter function required to comb the video signal for isolation. a. Levin's name is described in US Patent No. 4,158,209, issued June 12, 1979. The CCD delay line is a practical storage element which is mostly connected in series by continuously converting sample data from one element to the next at a predetermined ratio. This signal can be trapped nondestructively from one or more storage locations to realize special signal delays. Adder 266 averages two samples of the video signal trapped in one half of the chroma subcarrier cycle before and after the 1-H delay. The trapped and delayed signal is adjusted by the adder 266 so that the trapped and delayed signal is shifted by 180 ° with respect to the remaining untrapped delayed signal in the CCD device so that it has an appropriate chromaticity phase for the operation of the corresponding signal. Subcarrier cycle) has its chromaticity signal. The 1-H delayed signal is useful at the output of adder 266 and is applied to terminal 269 via amplifier 268. A useful signal at terminal 269 is coupled to comb filter input terminal 270. A more detailed description of a circuit for phase inverting a chroma signal is shown in US Pat. No. 4,272,785, issued June 9, 1981 by J. S. Fr. Instead of the example, the comb filter circuit 218 is manufactured by Tokyo Shibaura Corporation Limited, such as the T3928 integrated circuit.

The real-time video signal is applied to the terminal 217 of the comb filter and to the input connection 271 of the CCD register via a switch 264. If the signal combination, i.e., the noise, is determined in the real time signal and the 1-H delay signal from the terminal 269 is temporarily replaced by the position switch 164 with the real time signal supplied from the video detector 216, the comb filter Call the 1-H delay signal supplied to the input terminal 270.

Thus, the signal delayed in advance by one horizontal phase line is cycled again in the CCD register.

The foregoing demonstrates an advantageous technique of corresponding delay signals for real-time signals in video signal processing paths of color receivers utilizing special types of comb filters. Replacement devices for use in black and white television receivers include 1-H delay lines and switches. In that case, the delay line has its input terminal connected for receiving a real-time video signal, and the switch selectively supplies a real-time signal or a 1-H signal directly from the delay line to another processing circuit. In this type of device, the video signal does not cycle again in the delay line but only passes depending on the delay of one image line. For color signals, the delay corresponding to many integer horizontal periods can effectively use the same basic result. For example, in NTSC color signals, the delay circuit has a 2-H delay line, in which case it is necessary to invert the chromatic component phase of the video signal in order for the chromatic component to have an appropriate phase to act as a corresponding signal.

As such, what has been described in relation to this embodiment relates to an apparatus in which impulse noise or defect detection in a video signal is advantageously performed with a PLL audio signal detector for audio information detection. The embodiment shown in FIG. 4 uses separate IF amplifiers 230 and 236, while a common IF amplifier is used for amplifying the IF picture and voice carrier signals supplied to the output of buffer 224 as described above. Can be. Also, although the PLL 222 shows a mixer 234 for driving a 41.25 MHz frequency-converted signal for application to the phase detector 238 and the synchronous detector 254, the normal frequency of the VCO output signal is It can be changed to a frequency such as the voice carrier frequency and can be combined directly with detectors 238 and 254, as indicated by the dashed line in FIG. 4 without the frequency being converted by mixer 234. Of course, in other embodiments, mixer 234 is omitted and not useful for removing any common mode FM of the image and audio carriers.

Moreover, it should be understood that in the described embodiment, the combined control signal can be used for the control to be replaced by the delay video signal, but the combined control signal can be used together with other noise suppression or compensation circuitry. For example, the defect control signal can be used to be an automatic gain control device of a television receiver that is not sensitive to noise impulses (impulse noise) in the video signal.

Finally, the NTSC frequencies mentioned herein are illustrative only and can be scaled appropriately to operate at other frequencies in NTSC, PAL or SECAM television devices.

Claims (16)

The intermediate frequency signal has an image and audio carrier modulated with video and audio information, respectively, a PLL for generating a carrier demodulation reference signal, a phase detector followed by a low pass filter, and a controllable oscillator. In an audio demodulator for a television receiver, controlled by an output signal of the low pass filter, wherein a signal including audio information is combined from the PLL, the output signal of the controllable oscillators 50, 154, 252 is a first input to the image carrier. Is applied to the second input terminal of the frequency converter circuits 44, 144 and 234, and the voice carrier is applied to the first input terminal of the phase detectors 48, 152 and 238 to receive from the output signal of the frequency change circuit to the second input terminal. Demodulation voice information is combined from the output of the low pass filter (52, 156, 244). TV demodulator as set. The frequency converter circuit (44,144,234) of claim 1, wherein the nominal frequency of the oscillator output signal is equal to the frequency difference between the image and audio carriers. And a mixer (44,144,234) having a second input coupled to receive the oscillating output signal and an output for providing a signal at a nominal frequency equal to the frequency of the voice carrier. 3. The television demodulator according to claim 2, wherein the amplifier means (34, 36, 40, 42) responds to the IF signal to apply an almost unmodulated image carrier to the mixer (44). 4. The first bandpass filter (38) according to claim 3, wherein the first bandpass filter (38) has a symmetrical passband concentrated at a voice carrier frequency, an input / output stage coupled to receive the IF signal, and the first IF amplifier (46) A television sound demodulator, characterized in that it is coupled between an output end of the one filter network (38) and a first input end of the phase detector (48). 5. The second bandpass filter (36) according to claim 4, wherein the amplifier means has a second bandpass filter (36) and a second bandpass filter having a symmetrical passband concentrated at an image carrier frequency and an input / output stage coupled to receive the IF signal. And a second IF amplifier (40) coupled between the output end of the head 36 and the first input end of the mixer. 6. A television voice demodulator according to claim 5, characterized by an amplitude limiter (42) coupled between the output end of the second IF amplifier (40) and the first input end of the mixer (44). 7. The audio information according to claim 1, 2 or 6, wherein the audio information includes a plurality of audio channels, and the multi-channel decoder circuit 54 includes an input coupled to an output of the low pass filter 52 and an abnormality of the plurality of audio channels. And a television audio demodulator having an output stage for providing a signal. 3. The television voice demodulator according to claim 2, wherein the phase detector and the speech mixer each have a double balanced analog multiplier. 3. The filter of claim 2, wherein the first filter 136 has an input coupled to receive the IF signal and is concentrated at the frequency of each of the image and audio carriers to supply the image and audio carriers to its output. Humped amplitude vs. frequency response 137a, 137b with peak amplitudes S, P, amplifier 138 coupled to the output of the first filter 136 to amplify the image and audio carriers. Has an input stage and provides the amplified carrier to an output stage of the amplifier 138, and a second filter 146 is coupled to one of the first and second input stages 151 and 153 of the phase detector 152, And the second filter 146 has an amplitude-to-frequency response characteristic for providing a predetermined attenuation at the frequency of the image carrier and is substantially less than the predetermined attenuation at the frequency of the voice carrier. . 10. The television voice demodulator according to claim 9, wherein said second filter (146) is coupled between an output end of said amplifier (138) and said first input end of said phase detector. 10. The television sound demodulator according to claim 9, wherein said second filter (146) is coupled between an output end of said mixer (144) and a second input end of said phase detector (152). 12. The method according to claim 9, 10 or 11, characterized in that the selection means 140, 142 are coupled between the output end of the amplifier 138 and the input end of the mixer 144 to almost provide the image carrier at its output. TV voice demodulator. 13. A television sound demodulator according to claim 12, wherein said selecting means (140, 142) comprises a diode limiter circuit (142) for supplying an image carrier which is hardly modulated at the input of said mixer (144). 11. The apparatus of claim 10, wherein the first selection means (148, 150) provides additional attenuation between the output of the second filter (146) and the first input of the phase detector (152) to provide a predetermined attenuation at the frequency of the voice carrier. And a filter (146) to provide less than the level of. 15. The method according to claim 14, characterized in that the second selecting means (40, 142) is coupled between the output of the amplifier (138) and the input of the mixer (144) to supply almost only the image carrier to the mixer (144). TV voice demodulator. 15. The television sound demodulator according to claim 14, wherein said first and second selection means (148,150,140,142) comprise diode limiters (148,150).

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