KR910001146B1 - Electro-magnetic flow meter - Google Patents

Abstract

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Description

Electronic flowmeter

1 is a block diagram showing an embodiment of the present invention to excite based on a commercial frequency.

FIG. 2 is a block diagram showing an embodiment in which the flow rate feedback mode is changed from the embodiment of FIG.

3 is a block diagram showing a third embodiment of the present invention which compensates for these zero points based on signal processing at a commercial frequency.

FIG. 4 is a block diagram showing a fourth embodiment of the present invention in which the configuration of the embodiment of FIG. 3 is simplified.

FIG. 5 is a block diagram showing a fifth embodiment of the present invention in which the correction point of the zero signal is changed with respect to the embodiment shown in FIG.

FIG. 6 is a block diagram of an embodiment corresponding to FIG. 3 mainly comprising a commercial frequency signal processing system mainly designed for a low frequency signal processing system and improving response characteristics thereof.

FIG. 7 is a block diagram showing an embodiment corresponding to FIG. 4, which is further improved with respect to FIG.

FIG. 8 is a block diagram showing an eighth embodiment of the present invention when still with a square wave. FIG.

FIG. 9 is a waveform diagram for explaining the operation of the embodiment shown in FIG.

FIG. 10 is a waveform diagram illustrating the initial responsiveness of the embodiment shown in FIG.

FIG. 11 is a block diagram showing the ninth embodiment of the present invention in which the rate of change of noise is detected and the output of the two frequency excitation and the low frequency excitation is added at an arbitrary ratio.

FIG. 12 is a block diagram showing a tenth embodiment of the present invention showing a special configuration of the embodiment shown in FIG.

FIG. 13 is a block diagram showing a twelfth embodiment in which the switching part is improved with respect to the embodiment shown in FIG.

FIG. 14 is a block diagram showing a thirteenth embodiment of the present invention in which the magnitude of the noise is detected and the output of the two frequency excitation and the low frequency excitation is switched.

FIG. 15 is a block diagram showing a fourteenth embodiment of the present invention in which the distribution of the amplification degree of the circuit is changed to prevent saturation due to noise such as an amplifier.

FIG. 16 is a block diagram showing a fifteenth embodiment of the present invention that allows for a quick return to normal operation.

FIG. 17 is a block diagram showing a sixteenth embodiment of the present invention using a microcomputer. FIG.

FIG. 18 is a timing chart for explaining the operation of the embodiment shown in FIG.

FIG. 19 is a flowchart showing the signal processing procedure shown in FIG. 18; FIG.

20 is a calculation diagram showing the calculation in the flowchart art diagram of FIG.

FIG. 21 is a flowchart art showing the rate-mitting procedure in FIG.

FIG. 22 is a timing chart for explaining the operation of the embodiment having an excitation waveform different from that of the embodiment shown in FIG. 17; FIG.

FIG. 23 is a flowchart showing the signal processing procedure shown in FIG.

FIG. 24 is a calculation diagram showing the calculation in the flowchart art of FIG.

* Explanation of symbols for main parts of the drawings

10: conduit 11a, 11b: electrode

12: female coil 13,15,16: resistance

14: constant current source 16: low frequency constant current source

17: preamplifier

20,28,37,49,51,57,62,75,77: Multiplier

21,30 Amplifier 22,31 Demodulator

23,32,40,56: Voltage / frequency converter

24,33,43,52,67,68,70,73,83,85,87,90,92: Low pass filter (low pass filter)

26 low frequency signal processing system 27 commercial frequency signal processing system

34,61,71,88: High pass filter (high pass filter)

38: frequency separation circuit 42,48,55: zero correction point

44: zero detection amplifier 45,46,47,54 amplification system

50: deviation amplifier 53, 58: zero detection circuit

59,64: response correction point 60,63: response detection amplifier

65: excitation circuit 69: timing circuit

72,84: absolute value circuit 74: subtractor

76: addition circuit 78, 80: comparator

79a, 91: monostable circuit 79b: OR circuit

81: variable amplifier 82: compensation amplifier

83,89: Noise detection circuit 86: Time constant change circuit

93: selection circuit 94: power state detection circuit

95,96: Analog / Digital Converter 97: RAM

98: ROM 99: Bus

100: CPU 101: clock generator

102: divider 103: timing output port (To)

104: excitation circuit 105: digital / analog converter

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an electromagnetic flowmeter for applying a magnetic field to a fluid to be measured and measuring the flow rate thereof, and more particularly to an electromagnetic flowmeter having an improved excitation method and a signal processing method accordingly. .

Background Art Commercial electromagnetic flowmeters have conventionally adopted a commercial frequency excitation method that uses a commercial power supply to excite. The excitation method of commercial frequency can be manufactured quickly and cheaply. Ⓑ Its advantage is that it is not affected by low frequency irregular noise (hereinafter referred to as flow noise) that increases with the fluid velocity generated in slurry fluid or low conductivity fluid. There is a drawback that the zero point fluctuates when left unattended for a certain amount of time.

For this reason, the low frequency excitation method which excites at the low frequency of 1/2 or less of a commercial frequency is employ | adopted. If the low frequency excitation method is known, there is an advantage that a stable zero flow meter at zero point can be obtained. However, since the excitation frequency is low, it is close to the frequency of the flow noise, which is easy to be influenced by the flow noise, and this effect is remarkable especially when the flow velocity is increased. In addition, in order to alleviate the effects of flow noise, damping has a drawback of slow response.

In addition, recent electronic flowmeters tend to save energy. In particular, energy saving is an essential requirement in a two-wire type electromagnetic flowmeter that simultaneously supplies power and transmits signals by two wires. In such a case, it is necessary to reduce the electromotive force per unit flow rate. For example, in the conventional low frequency excitation method, what is about 0.5 mV / m / s is reduced to about 10 μV / m / S when the two-wire system is used. When the generator electromotive force is reduced by one or more digits as compared with the related art, the influence of the flow noise is relatively increased. Therefore, there is a limit to the energy saving by the low frequency excitation method.

The excitation by the commercial frequency has the advantage that the response speed is fast and unaffected by the flow noise, but there is a drawback that the zero point is unstable.

On the other hand, in the low frequency excitation, the zero point is stable, but there is a drawback that is susceptible to flow noise, and any excitation method adopts a stable zero point and is not affected by the flow noise, and also has a fast response speed. There is a problem that a flow meter cannot be obtained and the obstacle of energy saving cannot be eliminated.

SUMMARY OF THE INVENTION In view of the above-described prior art, the present invention aims to provide an electronic flowmeter which responds quickly to changes in flow rate, is free from flow noise, and is stable at zero viscosity.

The main constitution of the present invention which achieves this object is an excitation means for supplying a magnetic field having two different frequencies of a first frequency (high frequency) and a lower second frequency (low frequency), and excitation means by this excitation means. First demodulation means for discriminating the signal voltage generated corresponding to the flow rate according to the first frequency and outputting a first output, second demodulation means for discriminating and demodulating the signal voltage according to the second frequency, Low-pass filtering means for outputting the second output by low-pass filtering the output of the demodulation means with a large time constant, and performing predetermined calculations (addition, zero correction, response correction) using the first and second outputs It is to output the flow output.

EMBODIMENT OF THE INVENTION Hereinafter, embodiment of this invention is described along drawing. 1 is a block diagram showing an embodiment of the present invention that is excited based on a commercial frequency.

This embodiment shows a case where the feedback loop of the transducer of the electromagnetic flowmeter is formed separately on the high frequency side and the low frequency side, respectively. In the same drawing, reference numeral 10 denotes a conduit of a transmitter of an electromagnetic flowmeter, and an insulating lining is installed on the inner surface thereof. 11a and 11b are electrodes for detecting signal voltages. (12) is the excitation coil, and the magnetic field generated by this is applied to the fluid to be measured. A constant current of a commercial frequency flows through the excitation coil 12 through a constant current source 14 through a resistor 13, and a low frequency constant current source through the resistor 15 through the excitation coil 12 at the same time. According to (16), for example, a low frequency constant current of about 50/8 Hz flows superimposed. As a result, magnetic fields of two different frequencies are applied to the fluid to be measured, one of commercial frequency and one eighth of the commercial frequency.

On the other hand, the signal voltage is detected by the electrodes 11a and 11b and output to the preamplifier 17. In the preamplifier 17, the common mode voltage is eliminated and the impedance change is made and output to the coupling point 19 via the output terminal 18.

At the coupling point 19, a demodulator including a low pass filter (low pass filter) having a small time constant, which is amplified by the amplifier 21 by taking a deviation between the output of the preamplifier 17 and the output of the multiplier 20 ( 22) synchronous rectification or sample-holdered. The smoothed direct current output is converted into a pulse frequency signal having a constant pulse width by the voltage / frequency converter 23, fed back to the multiplier 20, and outputted to the low pass filter 24 and smoothed, thereby outputting the output ( V _L ) is output to the addition point 25. The multiplier 20 is composed of a switch, for example. One end of the switch is applied with a low frequency comparison voltage generated at both ends of the resistor 15, the switch is opened and closed with an output pulse of the voltage / frequency converter 23, and the voltage generated at the other end thereof is coupled to the coupling point 19. Output to. In addition, a low frequency comparison voltage from the resistor 15 is applied to the demodulator 22.

The amplifier 21, the demodulator 22, the voltage / frequency converter 23, the low pass filter 24, and the multiplier 20 constitute a low frequency signal processing system 26 which processes low frequency signal voltage therefrom, The signal corresponding to the low frequency excitation of the flow rate signal of the measuring fluid is processed and output to the addition point 25 as an output V _L. The time constant in the low frequency signal processing system 26 takes the time constant of the low pass filter 24 large and delays the response.

On the other hand, the commercial frequency signal processing system 27 is connected between the output terminal 18 and the addition point 25 of the preamplifier 17 in parallel with the low frequency signal processing system 26.

The output voltage of the output stage 18 of the preamplifier 17 and the output voltage and the deviation of the multiplier 28 are taken at the coupling point 29 and amplified by the amplifier 30. The output of the amplifier 30 becomes a current voltage smoothed by synchronous rectification or sample mode as a reference voltage based on the comparison voltage of the commercial frequency generated in the resistor 13 in the demodulator 31. This DC voltage is converted into a pulse frequency signal having a constant pulse width by the voltage / frequency converter 32 and returned to the multiplier 28. The output voltage of the voltage / frequency converter 32 is smoothed by the low pass filter 33 to be a DC voltage, which is output as the output V _H to the addition point 25 via the high pass filter (high pass filter) 34. Output The addition point 25 adds the output V _L and the output V _H , and outputs the combined output V _C to the output terminal 35.

With such a configuration, in the normal operation with small flow rate fluctuations, the commercial frequency processing system 27 does not respond due to the presence of the high pass filter 34, and the low frequency signal processing system 26 whose zero point is stable is mainly used. The output V _L is output as the synthesized output V _C , while the influence of the low pass filter 24, in which a large time constant is selected for the flow noise, is reduced so that the influence of the output V _L fluctuates. In addition, since the commercial frequency processing system 27 has a high excitation frequency, the frequency difference with the flow noise existing in the low frequency region is large, and the influence on the output V _H does not appear.

In other words, in a normal operation in which the flow rate fluctuation is small, it is possible to provide an electromagnetic flowmeter which is free from the influence of flow noise while ensuring a stable zero point.

Next, when the flow rate fluctuates rapidly, the low frequency signal processing system 26 does not respond because of the large time constant of the low pass filter 24, but the time constant of the commercial frequency signal processing system 27 is small and the high pass filter is small. Since the output is via 34, the output V _H is immediately output as the combined output V _C in response.

In the case where the flow rate is zero, no flow noise occurs, and thus there is no influence. The commercial frequency signal processing system 27 does not have a zero point drift due to the presence of the high pass filter 34, and thus outputs V _H. Is maintained at zero, and the output V _L of the stable low frequency processing system 26 at the zero point is output as the combined output V _C.

Further, the overall amplification of the low frequency signal processing system 26 and the overall amplification of the commercial frequency signal processing system 27 are approximately equal, and the time constants of the low pass filter of the entire low frequency processing system 26 and the commercial frequency signal are approximately equal. If the time constants of the high pass filter of the processing system 27 are made substantially the same, the response of the combined output V _C at which the flow rate changes rapidly and reaches the normal flow rate is smoothly operated.

FIG. 2 is a block diagram showing a second embodiment of the present invention which is modified from the embodiment shown in FIG. This example shows a case where the same feedback loop is used as the feedback loop of the transducer of the electromagnetic flowmeter. In the following description, the same code | symbol is attached | subjected to the part which has the same function, and the description is abbreviate | omitted.

Each constant current from the low frequency constant current source 16 and the commercial constant current source 14 is supplied to the excitation coil 12 via the resistor 36, and the resistor 36 generates a comparative voltage in which a low frequency and a commercial frequency are combined. Doing.

On the other hand, a deviation is obtained at the coupling point 19 between the output of the preamplifier 17 and the output of the multiplier 37 and input to the amplifiers 21 and 30, respectively. The output of the amplifier 21 has only low frequency components separated by the frequency separation circuit 38, and is a DC flow signal input to the demodulator 22 as a comparison voltage and generated in response to excitation of low frequency at the output terminal of the demodulator 22. Occurs. This flow rate signal is output as an output V _L1 to the addition point 39 via the low pass filter 24. The output of the amplifier 30 has only commercial frequency components separated by the frequency separation circuit 38, and is input to the demodulator 31 as a comparison voltage, and is a direct current generated in response to the excitation of the commercial frequency at the output terminal of the demodulator 31. A flow signal is generated. This flow rate signal is output as an output V _H1 to the addition point 39 via the high pass filter 34.

In the addition point 39, the combined output V _C1 obtained by combining the output V _L1 and the output V _H1 is converted into a pulse train having a duty width of a constant pulse width by the voltage / frequency converter 40 to thereby multiply the multiplier 37. ) as soon return to at the same time outputs the output (V ₀₎ to the output terminal 41. The multiplier 37 is composed of a switch or the like, which is opened and closed by an output pulse of the voltage / frequency converter 40, and returns the comparison voltage of the complex frequency generated across the resistor 36 to the coupling point 19. Let's do it.

Also in the above structure, it operates similarly to the case of FIG. 1 by selecting the time constant of the low pass filter 24 large.

3 is a block diagram showing a third embodiment of the present invention. In this embodiment, based on the commercial frequency signal processing system 27, a low frequency zero detection circuit is added to this to ensure stability of the zero point.

The output V _{H '} of the low pass filter 33 of the commercial frequency processing system 27 is output to the zero correction point 42 and also to the low pass filter 43.

The output of the low pass filter 43 is applied to the non-inverting input terminal (+) of the zero detection amplifier 44, and the output of the low pass filter 24 is applied to the inverting input terminal (-) so that these deviations are zero detection. It is computed in the amplifier 44. Thus, the zero detection amplifier 44 obtains at its output the zero signal ε ₁ corresponding to the deviation of the zero point of the commercial frequency signal processing system 27. The zero correction point 42 subtracts the output V _{H '} from the zero signal ε ₁ and outputs it to the output terminal 35.

Since the time constant of the low pass filter 24 is largely selected, the low pass filter 43 is largely selected in accordance with this in order to make the response speed the same.

Moreover, in the above structure, the amplifier system 45 which consists of the amplifier 21, the demodulator 22, the voltage / frequency converter 23, and the multiplier 20, the amplifier 30, the demodulator 31, the voltage It is necessary to make the amplification degree of each of the amplification system 46 composed of the / frequency converter 32 and the multiplier 28 equal.

Since the amplification system 46 performs signal processing at a commercial frequency, the response is fast without being affected by the flow noise, while the variation of the zero point is slow. On the other hand, since the amplification system 45 performs low frequency signal processing, it is influenced by flow noise, but the zero point is stable. However, since the low pass filter 24 has a large time constant, the flow noise does not affect the output of the low pass filter 24. Therefore, if the deviation between the output obtained through the low pass filter 43 having a large time constant and the output of the low pass filter 24 is calculated by the zero detection amplifier 44, the output of the amplification system 46 is calculated. Although the response is slow, a zero signal ε ₁ indicating the deviation of the zero point occurring in the amplification system 46 appears.

On the other hand, a zero response point of the commercial frequency signal processing system 27 fluctuates, but a zero response point ε ₁ corresponding to the change of the zero point is input to the zero correction point 42 and the subtraction thereof is zero. Since it is taken at the correction point 42, the output stage 35 has a quick response, thus eliminating the flow noise, and obtaining a flow rate signal from which the zero point variation is eliminated.

4 is a block diagram showing a fourth embodiment of the present invention.

In the embodiment shown in FIG. 3, the linearity of the amplification system 45 and the amplification system 46 need to be taken the same. In the embodiment shown in FIG. 4, the configuration is simplified by eliminating this necessity. Let's.

The amplification system 47 is composed of an amplifier 30, a demodulator 31, a voltage / frequency converter 32, a multiplier 28, and a zero correction point 48. The output of the multiplier 28 and the zero signal [epsilon] ₂ are input to the zero correction point 48, and the difference thereof is taken at the zero correction point 48 and output to the coupling point 19.

The multiplier 49 is supplied with a comparison voltage of the output of the voltage / frequency converter 32 and the frequency from the resistor 15, and modulates the low frequency modulated voltage modulated by the comparison voltage to one end of the input of the deviation amplifier 50. Is authorized. The deviation amplifier 50 takes the difference between the modulation voltage and the output of the preamplifier 17 and outputs it to the demodulator 22. Since the demodulator 22 is applied with a low frequency comparison voltage from the resistor 15, a DC voltage corresponding to the low frequency component of the signal voltage is obtained at its output. This DC voltage is converted into a frequency in the voltage / frequency converter 23 through a low pass filter 24 having a large time constant, and multiplied by the comparison voltage of the commercial frequency from the resistor 13 at this frequency. Is modulated by and is output to the zero correction point 48 as a zero signal ε ₂ representing the variation of the zero point of the commercial frequency. Since the zero correction point 48 is subtracted from the output of the commercial frequency of the multiplier 28 by the zero signal ε ₂ modulated at the commercial frequency, the zero point is corrected through the low pass filter 52 at the output terminal 35. The flow rate signal is obtained.

The zero detection circuit 53 is constituted by the deviation amplifier 50, the demodulator 22, the low pass filter 24, the voltage / frequency converter 23, the multiplier 51, and the like.

Since the zero detection circuit 53 is used only for suppression of the zero point and operates as a kind of deviation amplifier as a whole, linearity is not important. In addition, since the zero point at the commercial frequency normally changes slowly, the response of the zero detection circuit 53 may be late, thereby smoothing the flow noise.

5 is a block diagram showing a fifth embodiment of the present invention. This embodiment changes the correction point of the zero signal ε ₂ with respect to the embodiment shown in FIG.

The output of the demodulator 31 of the amplification system 54 and the zero signal ε ₃ are input to the zero correction point 55 and subtracted and output to the voltage / frequency converter 56. The voltage / frequency converter 56 is selected to be equal to the amplification degree of the voltage / frequency converter 32.

The multiplier 57 is supplied with the output of the voltage / frequency converter 56 and the low frequency comparison voltage from the resistor 15, and applies the low frequency modulation voltage modulated by the comparison voltage to one end of the input of the deviation amplifier 50. . The deviation amplifier 50 takes a deviation between the modulation voltage and the output of the preamplifier 17 and outputs it to the demodulator 22. The demodulator 22 outputs a DC voltage corresponding to the low frequency component among the signal voltages as a zero signal ε ₃ to the zero correction point 55.

The zero detection circuit 58 is composed of a multiplier 57, a deviation amplifier 50, a demodulator 22, a low pass filter 24, and the like.

FIG. 6 is a block diagram of an embodiment corresponding to FIG. 3 mainly using a commercial frequency signal processing system because the response characteristics are mainly improved by using the low frequency signal processing system.

The output of the stable low pass filter 24 at the zero point of the low frequency signal processing system 26 is output to the response correction point 59 and is applied to one end of the input of the response detection amplifier 60. At the other end of the response detection amplifier 60, the output of the commercial frequency signal processing system 27 with a fast response is applied, taking a deviation from this and the output of the low pass filter 24 and interposing the high pass filter 61. To the response correction point 59.

The response correction point 59 outputs to the output stage 35 in consideration of the output of the high pass filter 61 and the output of the low pass filter 24.

The output of the low frequency signal processing system 26 has a stable zero point but a late response. On the other hand, the output of the commercial frequency signal processing system 27 is unstable at the zero point but gives a quick response.

Therefore, the output obtained by the response detection amplifier 60 with the deviation between the output of the low pass filter 24 and the output of the low pass filter 33 contains a direct current, but this is via the high pass filter 61. Since the obtained response compensation signal (V _C ) does not contain a DC component and does not damage the zero stability line and shows a fast response, the output of the response compensation circuit 59 indicates that the late response of the low pass filter 24 is a response compensation signal. Compensated by (V _C ), the response is fast and the zero point is stable output.

FIG. 7 is a block diagram showing an embodiment mainly comprising a low frequency signal processing system corresponding to FIG. 4 mainly using a commercial frequency signal processing system further improved with respect to FIG.

The output of the voltage / frequency converter 23 is applied to the multiplier 62, and the multiplier 62 modulates it with the comparison voltage of the commercial frequency obtained from the resistor 13, and modulates the modulated output of the input of the response detection amplifier 63. Will output at one end. The response detection amplifier 63 calculates and outputs a deviation between the modulation output applied to one end of the input and the output of the preamplifier 17 applied to the other end thereof. The output of the response detection amplifier 63 is demodulated by the demodulator 31 by the comparison voltage of the commercial frequency obtained from the resistor 13 so that the response compensation signal V _{C '} corresponding to the commercial frequency component of the signal voltage is the response compensation point. Is output to (64).

The low frequency output from the low pass filter 24 is applied at the response correction point 64. The output is output to the voltage / frequency converter 23 by correcting the response with the response compensation signal V _{C '} .

Thus, when the response compensation signal V _{C '} is applied to the inside of the low frequency signal processing system 26, the response detection composed of the response detection amplifier 63, the demodulator 31, the multiplier 62, and the like is performed. The amplification degree of the circuit is not an error factor even if it is not stable.

8 is a block diagram showing an eighth embodiment of the present invention when an excitation is performed by a square wave.

The exciting coil 12 is supplied with an _exciting current I _f from the exciting circuit 65, and the exciting circuit 65 is configured as follows.

The reference voltage E ₁ is applied to the non-inverting input terminal (+) of the amplifier Q ₁ via the switch SW ₁ , and the output terminal thereof is connected to the base of the transistor Q ₂ . The emitter of the transistor Q ₂ is connected to the common COM via the resistor R _f and is connected to the inverting input terminal (−) of the amplifier Q ₁ . An excitation voltage ( _ES ) between the common (COM) and the collector of the transistor (Q ₂ ) is connected to the series circuit of the switch (SW ₂ ) and the switch (SW ₃ ), and the switch (SW ₄ ) and the switch (SW) connected in parallel thereto. ₅ ) through a series circuit. The exciting coil 12 is connected to the connection point of the connection point of the switch (SW ₄₎ (SW ₅₎ of the switch _{(SW 2) (SW 3)} . The timing signals S ₁ , S ₂ , and S3 control opening and closing of the switches SW ₁ , SW ₂ , SW ₅ , and SW ₃ , SW _4, respectively.

On the other hand, the signal voltage is detected by the electrodes 11a and 11b and output to the preamplifier 17. In the preamplifier 17, common mode voltage is removed and impedance conversion is performed, and is output to the coupling point 66 via the output terminal thereof. The signal voltage at the connection point 66 is the switch (SW ₇₎ a, or an inverting amplifier (Q ₃₎ and switch the low-pass filter (67 having a smaller time constant, respectively via a series circuit (SW ₈₎ by interposition ) Is applied.

The signal voltage at the connection point 66 is the switch (SW ₉₎ for it, or an inverting amplifier (Q ₄₎ and switch the low-pass filter to have a smaller time constant, respectively via the series circuit of (SW ₁₀₎ interposed Is applied to (68). The switches SW ₇ , SW ₈ , SW ₉ , and SW ₁₀ are opened and closed by timing signals S ₇ , S ₈ , S ₉ , and S ₁₀ from the timing circuit 69, respectively. . The output of the low pass filter 67 is through a low pass filter 24 having a large time constant, and the output of the low pass filter 68 is a series circuit of the variable gain amplifier Q ₅ and the high pass filter 34. Are added at the addition point 25 via the low pass filter 70 and are output to the output terminal 35 via the low pass filter 70.

In addition, a variable gain amplifier (Q ₅₎ is controlled to be equal to the magnitude of the output voltage (V _H) of the output voltage (V _L) and a high-pass filter 34 of the low-pass filter 24.

In the above configuration, the low frequency loop formed by the coupling point 66 and the addition point 25 via the low pass filter 24, the coupling point 66 and the addition point 25 via the high pass filter 34. Each constant is selected so that the sum of the transfer functions of the respective loops with the high frequency loop formed by the sum is 1. In practice, the time constants of the low pass filter 24 and the high pass filter 34 are the same, and the magnitude of the signal voltage of each loop is adjusted by adjusting the gain of the variable gain amplifier Q ₅ .

Next, the operation of the embodiment shown in FIG. 8 will be described with reference to the waveform diagram shown in FIG.

The timing signal S ₁ is repeatedly turned on and off as shown in FIG. 9A, whereby the reference voltage E ₁ is applied to the non-inverting input terminal (+) of the amplifier Q ₁ . To be turned off. On the other hand, the low frequency switch SW ₂ (SW ₅ ) and the switch SW ₃ by the timing signal S ₂ [FIG. 9 (b)] and the timing signal S ₃ (FIG. 9 (c)). Since (SW ₄ ) is turned on each other, as shown in FIG. 9 (d), the excitation current I _f in which the low frequency (period: 2T) and the high frequency (period: 2T) are combined flows.

Since the signal voltage at the coupling point 66 is sample-holed with the timing signals S ₇ (S ₈ ) shown in Figs. 9 (e) and (f), the voltage shown in Fig. 9 (g) It is obtained on the output side of this switch (SW ₇ ). The voltage smoothed by the low pass filter 67 is output to the addition point 25 via the low pass filter 24.

The signal voltage at the connection point 66 is therefore sampled by the ninth degree (h) timing the illustrated timing (i) the signal (S ₉₎ (S _10), the output side of the switch (SW ₉₎ The signal voltage shown in FIG. 9 (j) is output, and the signal voltage is adjusted in the variable gain amplifier Q ₅ and outputted to the addition point 25 via the high pass filter 34.

Each signal voltage added at the addition point 25 is smoothed in the low pass filter 70 and output to the output terminal 35. In this case, if the transfer function of the low pass filter 24 is 1 / (1 + T ₁ S) and the transfer function of the high pass filter 34 is T ₂ S / (1 ÷ T ₂ S), Select each time constant (T ₁ ) (T ₂ ) so that the sum equals 1 as T ₁ = T ₂ .

When each time constant is selected to achieve such a relationship, when the signal voltage is changed into a step shape, as shown in FIG. 10A, the output signal voltage V _L and the high pass filter of the low pass filter 24 are shown. The output signal voltage V _H of 34 changes. Therefore, the addition output V _{O obtained} by adding these represents a step shape change without error as shown in FIG. 10B.

On the other hand, when the sum of the transfer functions is not 1, when the signal voltage changes to the step shape, as shown in FIG. 10C, the output signal voltage VL and the high pass of the low pass filter 24 are shown. The output signal voltage V _{H '} of the filter 34 changes. Therefore, the addition output V _{O 'obtained} by adding these represents a change including an error ε ₁ as shown in FIG. 10D.

The low pass filter 24 and the high pass filter 34 may be any type of pass filters regardless of the first pass filter as long as the sum of these transfer functions is one. In addition, if the precision of the step response is not required, the sum of the transfer functions does not have to be set correctly.

FIG. 11 is a block diagram showing the ninth embodiment of the present invention, taking advantage of the two-frequency and low-frequency excitation.

In electromagnetic flowmeters, in addition to flow noise, differential noise is mixed into the electrodes in various circuits by capacitive coupling, etc., which causes a slow variation (steam fluctuation) in the case of a high frequency excitation. On the other hand, the low frequency excitation has the advantage that the zero point is stable and does not respond to uneven noise, while the frequency band responds to flow noise in the same region as the frequency band of the flow noise.

On the other hand, in the case of a two-frequency excitation having two frequencies of low frequency and high frequency as shown in Fig. 8, the zero point is stable, and as can be seen from the description above, there is an advantage that it is also stable against flow noise. Since it has a high-frequency signal processing system, it is stable in the long term even for undifferentiated noise, but it is weak in medium-term fluctuations.

Therefore, if there is no flow noise or fine noise, or if there is fine noise, stable signal can be obtained even if signal processing is performed by the low frequency signal processing system alone. You can get a stable zero output. In addition, when the flow noise and the fine powder noise are together, the fluid is flowing, so the two-frequency excitation side of the zero point may be changed even if the zero point fluctuates between zero and moderate time due to the undifferentiated noise. There is no variation and it is averaged over time so that it is practically no error.

Therefore, in the embodiment shown in FIG. 11, the magnitude of the flow noise is detected by the output of the low frequency side, and the optimum ratio is obtained by varying the addition ratios of the low frequency side and the 2nd frequency side accordingly.

e△

E₂로 선정되어 있으므로 신호(S_L)의 변화분(△S_L)이 커짐에 따라서 곱셈기에 인가되는 전압은 작아진다. 곱셈기(75)는 신호(S_L)와 감산기(74)의 출력과의 곱을 취하여 비교가산회로(76)의 일단에 출력한다.From the output of the low pass filter 67 on the low frequency side, a signal S _L containing slurry noise or the like is detected, and the variation ΔS _L is extracted by the high pass filter 71 to extract the absolute value circuit 72. Output to. The absolute value circuit 72 in the minute change (△ S _L) of the absolute value | temporarily applied as an output (e △) in the subtracter 74 by taking it through a low-pass filter (73) | △ S _L. At the other end, the reference voltage E ₂ is applied, and these voltage differences are applied to the multiplier 75. Here, the output e △ is 0

e △

Voltage applied to the multiplier according to this larger change minute (△ S _L) of the signal (S _L) because it is named the E ₂ is small. The multiplier 75 multiplies the signal S _{L by} the output of the subtractor 74 and outputs it to one end of the comparison addition circuit 76.

On the other hand, the multiplier 77 takes the product of the combined output V _C from the addition point 25 and the output eΔ of the low pass filter 73 and applies it to the other end of the addition circuit 76. If you express this relationship as an expression,

V _O = KV _C + (1-K) S _L K is a ratio controlled by the output e "Δ.

The addition circuit 76 and outputs it as a flow rate output (V _O) to an output terminal (35) by by adding the respective output through a low-pass filter 70 from the multiplier 75, 77.

Accordingly, the signal output of the (S _L) changes minute (△ S _L), the multiplier 75 is larger be increases the output of the low pass filter 73 of the will be made smaller, whereas the output of the multiplier 77 is therefore larger , The ratio of the output of the low frequency side to the output of the two frequency side becomes small.

In this manner, according to the present embodiment, an optimum output can be obtained by detecting fluctuations in slurry noise or the like on the low frequency side and automatically changing the ratio between the output of the two frequency side and the output of the low frequency side.

FIG. 12 is a block diagram of the tenth embodiment of the present invention, which simplifies the configuration of the embodiment shown in FIG.

The output eΔ of the low pass filter 73 is applied to one end of the input of the comparator 78 to which the reference voltage E ₃ is applied to the other end of the input, and is a low frequency signal at the output of the comparator 78. Switch between (S _L ) and the switch (SW ₁₁ ) to which the composite output (V _C ), a signal on the two-frequency side, is applied. When the output e △ of the low pass filter 73 exceeds the reference voltage E ₃ , the switch SW ₁₁ is switched to the synthesized output V _C , and when not exceeded, the switch is switched toward the low frequency signal S _L. .

FIG. 13 is a block diagram showing a twelfth embodiment of the present invention in which hysteresis is added during switch switching with respect to the embodiment shown in FIG.

This embodiment applies the output of the comparator 78 and the output via the monostable circuit 79a to the input terminal of the OR circuit 79b so that the switch SW ₁₁ is controlled by the output.

In this way, even if the output of the comparator 78 changes, the switch SW ₁₁ is not switched for the time of the predetermined pulse width of the pulse generated in the monostable circuit 79a, so that chattering can be prevented.

FIG. 14 is a block diagram showing a thirteenth embodiment of the present invention in which the magnitude of the output of the low frequency side is detected to switch between the two frequency side and the low frequency side.

11 to 13 detect the change in the low frequency signal and use it to switch the switch SW _{11. In} this embodiment, however, the absolute value of the low frequency signal is detected. will be. The signal S _L is applied to the other end of the input of the comparator 80 to which the reference voltage E ₄ is applied to one end of the input, and the switch SW ₁₁ is switched to the low frequency side or the second frequency side as its output. When the signal S _L is greater than the reference voltage E ₄ , the switch SW ₁₁ is switched to two frequencies to output the combined output V _C.

FIG. 15 is a block diagram showing a fourteenth embodiment of the present invention in which the distribution of the amplification degree of the circuit is changed to prevent saturation due to noise such as an amplifier.

Electromagnetic flowmeters tend to reduce the excitation power in order to save energy, but the smaller the excitation power, the smaller the generated signal voltage. Thus, in order to compensate for this, the amplification degree of the circuit tends to be increased. On the other hand, since slurry noise and the like have a constant size irrespective of the excitation power and the like, they tend to be weakened by noise such as slurry noise. Thus, in this embodiment, the distribution of the amplification degree in the circuit is changed in accordance with the noise to prevent saturation of the circuit.

A variable amplifier 81 is installed between the output terminal of the preamplifier 17 and the coupling point 66, and a compensation amplifier 82 is installed between the addition point 25 and the low pass filter 70 of the output. . Then, by the noise detected by the noise detection circuit 83 amplified by the variable amplifier 81 and the compensation amplifier 82, the product of the corresponding amplification degree of the variable amplifier 81 and the compensation amplifier 82 is controlled to be constant. .

The variable amplifier 81 is configured as follows. The common inverting point COM is connected to the non-inverting input terminal (+), the output terminal of the preamplifier 17 is connected to the inverting input terminal (-) of the amplifier Q ₆ via the resistor R1, and the output terminal Between the series circuit of the resistor R ₁ and the switch SW ₁₂ , the series circuit of the resistor R ₂ and the switch SW ₁₃ , and the direct connection circuit of the resistor R ₃ and the switch SW ₁₄ are parallel to each other. Is connected. These switches SW ₁₂ , SW ₁₃ , and SW ₁₄ are switched to control signals S ₁₂ , S ₁₃ , and S ₁₄ from the noise detection circuit 83, respectively.

The noise detection circuit 83 is configured as follows. The output of the preamplifier 17 is calculated by the absolute value circuit 84 and output to the low pass filter 85. The output of the low pass filter 85 is the other end of the comparator Q ₇ to which the reference voltage E ₅ is applied to one end of the input and the other end of the comparator Q ₈ to which the reference voltage E ₆ is applied to one end of the input. Is applied to each, and its size is determined. Each output of the comparator Q ₇ (Q ₈ ) is applied to each input terminal of the NOR gate Q ₉ , and these NORs are calculated at the NOR gate Q ₉ to output a control signal S ₁₄ to the output terminal thereof. do. In addition, each output of an output of the comparator Q ₇ and the output of the comparator Q ₈ and the output of the comparator Q ₈ inverted at the inverter Q ₁₁ is applied to each input terminal of the NOR gate Q ₁₀ , and these NORs are NOR gates. It is computed in the (Q _10), and outputs a control signal (S ₁₃₎ to its output. In addition, the control signal S ₁₂ is obtained as the output of the comparator Q ₇ . These control signals detect the magnitude of the noise generated at the output terminal of the preamplifier 17, switch each switch according to the magnitude, and change the amplification degree of the variable amplifier 81.

In addition, the compensation amplifier 82 is configured as follows so as to be the variable amplifier 81. Non-inverting input terminal (+) is an amplifier (Q _{6 ')} connected to the common potential point (COM) - at the same time as connection via an inverting input terminal of the (), the resistance (R _i output terminal of the preamplifier _17') that Between the output terminal is a series circuit of resistor (R _{i '} ) and switch (SW _12' ), a series circuit of resistor (R _{2 '} ) and switch (SW _13' ), and resistor (R _{3 '} ) and switch (SW _{14 '} ) series circuits are connected in parallel, respectively. These switches SW _{12 '} , SW _13' , and SW _{14 '} are switched by the control signals S ₁₂ (S ₁₃ ) and S ₁₄ from the noise detection circuit, respectively. By these switching, the compensation amplifier 82 is varied so as to compensate for the amplification degree of the variable amplifier 81, and each constant of the element is selected so that the overall amplification degree is kept constant. In this way, the amplification degree is changed corresponding to the magnitude of the noise to prevent saturation of the circuit due to the noise.

FIG. 16 is a block diagram showing a fifteenth embodiment of the present invention that enables a quick return to normal operation.

Since the two-frequency excitation has a low pass filter with a large time constant on the low frequency side, it can be used until it operates in the normal state when the power is turned on or when the noise is eliminated and it returns to the normal state from the abnormal state. The problem is that it takes time. Thus, this embodiment solves this problem.

The time constant change circuit 86 is configured as follows. Between the output terminal of the low pass filter 67 having a small time constant and the addition point 25, the transistor Q ₁₂ , which is short-circuited by the switch SW ₁₅ , is connected in series, and the output terminal is connected to the common potential point ( The capacitor C ₁ is connected to the COM. The time constant of the low pass filter 87 is varied by the transistor Q ₁₂ and the capacitor C ₁ whose internal resistance is controlled by the control signal S ₁₅ .

A capacitor C ₂ is connected in series between the output terminal of the variable gain amplifier Q ₅ and the addition point 25, and is short-circuited by a switch SW ₁₆ between the output terminal and the common potential point COM. The transistor Q ₁₃ whose internal resistance is controlled by the control signal S ₁₅ is connected. The high pass filter 88 is constituted by the transistor Q ₁₃ and the capacitor C ₂ .

The noise detection circuit 89 is configured as follows. The output of the preamplifier 17 is a reference voltage to one end of the input (E ₇₎ is applied to a comparator (Q ₁₄₎ and one reference voltage to the input (E ₈₎ The other end of each input of the authorized comparator (Q ₁₅₎ Is applied to each input of the OR gate Q ₁₆ . The output of the OR gate Q ₁₆ is applied to the other end of the input of the comparator Q ₁₇ to which the reference voltage E ₉ is applied to one end of the input via the low pass filter 90, and the output end thereof is a monostable circuit. It is output to 91. The output of the monostable circuit 91 is inverted by the inverter Q ₁₈ and output to the low pass filter 92. The low pass filter 92 is composed of a resistor R ₄ and a capacitor C ₃ , and both ends of the resistor are short-circuited with a diode D ₁ , and a control signal S ₁₅ is applied to an output terminal of the low pass filter 92. Get

When the noise of the output of the preamplifier 17 is small, the output of the OR gate Q ₁₆ is in the "L" level, and the output of the comparator Q ₁₇ is also in the "L" level. Therefore, the output of the monostable circuit 91 is at the "L" level, but the output of the low pass filter 92, that is, the control signal S ₁₅ is at the "H" level. This state is applied to the transistors Q ₁₂ (Q ₁₃ ) via the selection circuit 93 to keep their internal resistance high.

Next, when large noise exceeding the reference voltage E ₇ (E ₈ ) appears at the output of the preamplifier 17, the output of the low pass filter 90 is at the “H” level. Therefore, as long as there is noise, the output of the comparator Q ₁₇ maintains the "H" level.

However, when this noise disappears, the output of the comparator Q ₁₇ falls to the "L" level. Therefore, the downstable edge is detected by the monostable circuit 91, and a pulse having a predetermined length of "H" level is applied to the output. Output Since this pulse is inverted in the inverter Q ₁₈ and applied to the low pass filter 92, the output control signal S ₁₅ drops to the "L" level once, and then the time constant of the low pass filter 92 As soon as you decide to recover to "H" level.

Therefore, when the noise disappears, the internal resistance of the transistors Q ₁₂ (Q ₁₃ ) decreases for a predetermined time and then immediately rises thereafter, so that the return to the normal state after the noise disappears quickly.

The power supply state detection circuit 94 is configured as follows. E ₁₀ is a power source and is applied to the resistor R ₅ and the capacitor C ₄ via a switch SW ₁₇ . The voltage at the connection point between the resistor R ₅ and the capacitor C ₄ is applied to the other end of the input of the comparator Q ₁₉ to which the reference voltage E ₁₁ is applied at one end of the input. The output of the monostable circuit 91 and the comparator Q ₁₉ is input to each input terminal of the OR gate Q ₂₀ , and the switch SW ₁₅ (SW ₁₆ ) of the time constant change circuit 86 is output to the output of the OR gate. Open and close the).

When the power supply voltage is applied, the voltage at the connection point of the resistor R ₅ and the capacitor C ₄ is at the "L" level, so that the output of the comparator Q ₁₉ is at the "H" level and the switch SW ₁₅ (SW ₁₆ ). Is in the on state, but after a predetermined time determined by the time constants of the resistor R ₅ and the capacitor C ₄ , the output of the comparator Q ₁₉ is inverted to the "L" level so that the switch SW ₁₅ ( SW ₁₆ ) is turned off.

Therefore, since the time constant of the time constant change circuit 86 is small for a predetermined time from the time of power-on, it returns to normal operation quickly.

In the normal return after the noise is extinguished, pulses of the "H" level from the monostable circuit 91 to the predetermined time are supplied to the switches SW ₁₅ and SW ₁₆ via the OR gate Q ₂₀ . Turn the switch on.

In the selection circuit 93, the voltage at the connection point between the control signal S ₁₅ , the resistor R ₅ , and the capacitor C ₄ is applied to the input terminal of the comparator Q ₂₁ , respectively, and to the output of the comparator Q ₂₁ . Control the switch SW ₁₈ . It is on the ① side when the power is turned on, and is switched to the ② side in the normal state.

FIG. 17 is a block diagram showing a sixteenth embodiment of the present invention using a microcomputer.

The output of the preamplifier 17 is converted into a digital signal in the analog / digital converter (A / D _L ) 95 and the analog / digital converter (A / D _H ) 96, respectively, and the bus 99 It is stored in the RAM 97 through it. The ROM 98 stores a predetermined arithmetic program and initial data, and is calculated according to the arithmetic procedure stored in the ROM 98 under the control of the CPU 100, and the result is stored in the RAM 97.

Reference numeral 101 denotes a clock generator, which is divided at 1 / n by the divider 102 and supplied to the CPU 100 and the analog / digital converter 96 as a system clock Sh.

The CPU 100 determines the waveform of the excitation current I _f1 (or I _f2 ) in the timing signal output port TO 103 via the bus 99 in accordance with an arithmetic program stored in the ROM 98. Output the timing. The timing signal output port 103 outputs a timing signal S ₂₂ (S ₂₃ ) (S ₂₄ ) (S ₂₅ ) for switching the excitation current in accordance with this timing. These timing signals S ₂₂ , S ₂₃ , S ₂₄ , and S ₂₅ switch the switches SW ₂ , SW ₃ , SW ₄ , and SW ₅ of the excitation circuit 104.

The timing signal output port 103 outputs the timing signal Sl to the analog-digital converter 95 in accordance with the timing specified by the CPU 100, and samples the output of the preamplifier 17.

On the other hand, a predetermined operation is executed in the CPU 100 by using the data stored in the RAM 97 by the operation program stored in the ROM 98, and the result of the operation is stored in the RAM 97 and at the same time as the bus. Through the 99 and the digital / analog converter 105, the output stage 106 outputs the flow rate output.

Next, the timing diagram shown in FIG. 18, the flowchart diagram shown in FIG. 19, the calculation diagram shown in FIG. 20, and the flowchart diagram shown in FIG. 21 are shown in FIG. The operation of one embodiment will be described.

The system clock Sh obtained at the output of the frequency divider 102 shown in FIG. 17 is a waveform shown in FIG. 18A, which is supplied to the CPU 100. In step 1 of FIG. 19, the CPU 100 is bused by a predetermined arithmetic program stored in the ROM 98 in synchronization with the interruption stop timing (Fig. 18 (g)) of the system clock Sh. A timing output indicating the switching timing of the excitation waveform is output to the timing signal output port 103 via 99.

In step (2), the timing signal output port 103 receives this switching timing and receives the timing signal S ₂₅ (Fig. 18B), S ₂₄ [Fig. 18C), and S ₂₃ [ Fig. 18 (d)] and S ₂₂ [Fig. 18 (e)] are output to the switches SW ₅ (SW ₄ ) (SW ₃ ) (SW ₂ ) of the excitation circuit 104, respectively. The excitation circuit 104 receives these timing signals and outputs the excitation current I _f1 of the waveform shown in FIG. _18F to the excitation coil 12. As shown in Fig. 18 (h) (i), the excitation waveform is a waveform in which the timing number (i) constitutes one cycle with 0 to 15 and repeats it. It is shown as. The excitation waveform has a waveform obtained by multiplying a low frequency waveform with a high frequency waveform.

Next, the process proceeds to step 3. Steps 3 to 6 show the procedure for reading out data from the analog / digital converters 96 and 95.

In step 3, the data input from the analog-digital converter 96 for each cycle in synchronization with the system clock Sh (Fig. 18 (a)) is transferred to the bus (Fig. 18 (j)). 99 is stored in the predetermined data area Hi of the RAM 97 under the control of the CPU 100.

Next, the process proceeds to step 4, and it is judged whether or not the read-out timing number i is 0. If not, the process proceeds to step 6, and if it is 0, the process proceeds to step 5.

In step 5, the data inputted from the analog / digital converter 95 is sampled by the timing of the timing signal Sl (Fig. 18 (k)) output from the timing signal output port 103. As shown in Fig. 18 (l), the predetermined data area of the RAM 97 is controlled by the CPU 100 via the bus 99 ..., L ₁ (n-1), L ₁ ( n), L ₁ (n + 1), ..., and proceeds to step (8).

Next, in step 7, the analog / digital transducer 95 is inputted by the sample timing by the timing signal S1 (degree 18 (k)) output from the timing signal output port 103. The predetermined data area of the RAM 97 is controlled by the CPU 100 via the bus 99 as shown in FIG. 18L. . . , L ₁ (n-1), L ₁ (n), L ₁ (n + 1),. . . . Is stored, and the flow advances to step 8.

In step 8, it is determined whether the timing number i is an odd number, and if it is an odd number, the procedure proceeds to step a, and if it is not an odd number, the procedure shifts to step 12.

In step 9, high frequency demodulation operation is performed. In the demodulation operation, the data Hi stored in the RAM 97 is used and stored in the ROM 98 under the control of the CPU 100 at the timing shown in FIG. 18 (m). It calculates by the arithmetic formula shown in the column of the high frequency demodulation operation e _Hi shown, and stores the result in RAM97. By this demodulation operation, the electrochemical noise generated in the electrodes 11a and 11b is eliminated, and the differential noise is kept at a constant value so that it is not an error factor. In Fig. 20, the constant composed of A is expressed as A = Tc / (Tc + " Tc) when Tc is a derivative or integral constant and " ΔTc is a calculation cycle shown in Fig. 18 (f).

In step 10, it is a step which performs a latelimide calculation. This step is not necessarily required, but is described herein for convenience of explanation. This operation is a calculation for limiting this amplitude to a predetermined value because noise of a large amplitude is mixed when the operation on the high frequency side is good in the two-frequency excitation.

This explanation is made using FIG. In FIG. 21, the upper limit of noise is determined in step A. FIG. In this case, the result of adding the upper limit width (e _R ) to the previous calculation result (e _Hi -2) on the high frequency side is compared with the calculation result (e _Hi ) of the current time. It is assumed that the process proceeds to step C, and if the current value is large, the process proceeds to step B.

In step B, the value obtained by adding the late limit width e _R to the previous and subsequent calculation result e _Hi -2 is output as the current value, and the amplitude is limited.

In step C, the lower limit of the noise is determined. Here, the value obtained by subtracting the rate limit width e _R from the previous calculation result (e _Hi -2) on the high frequency side is compared with the calculation result (e _Hi ) of the present time. If the value of this time is large, it is determined that the process proceeds to step D.

In step D, the value obtained by subtracting the rate limit width e _R from the previous calculation result e _Hi -2 is output as the current value and the amplitude thereof is limited.

Next, the process proceeds to step 11. In this case, the high frequency filter (F _Hi ) on the high frequency side is executed. In the filter operation, the high-pass filter shown in FIG. 20 stored in the ROM 98 under the control of the CPU 100 using the data e _Hi stored in the RAM 97 and the result of the previous filter operation. The calculation is performed using the formula shown in the column of (F _Hi ), and the result is stored in the RAM 97.

Next, the process proceeds to step 12. In step 12, it is determined whether the timing number i is 0 or 8, and if it is 0 or 8, the process proceeds to step 13, and if it is not 0 or 8, the process proceeds to step 15.

In step 13, low frequency demodulation operation is performed. In the demodulation operation, the data stored in the RAM 97 ... L ₀ (n-1), L ₀ (n), L ₀ (n + 1), ... L ₁ (n-1), Fig. 20 stored in the ROM 98 under the control of the CPU 100 at the timing shown in Fig. 18 (n) using L ₁ (n), L ₁ (n + 1), ... The operation is performed using the arithmetic expression indicated in the column of one low frequency demodulation operation e _Li , and the result is stored in the RAM 97. In Fig. 20, the constant B is represented by B = ΔT / ("ΔT + T).

In step 14, the low frequency filtering F _{Li at} the low frequency side is executed. In the filter operation, the data (e _LO ) (e _LB ) stored in the RAM 97 and the result of the previous filter operation are used to store FIG. 20 stored in the ROM 98 under the control of the CPU 100. The calculation is performed using the arithmetic formula shown in the column of the displayed low-pass operation F _Li and the result is stored in the RAM 97.

In step 15, it is determined whether the timing number i is an odd number, and if it is an odd number, the process proceeds to step 16; In step 16, addition operation is performed. As shown in FIG. 20 stored in the ROM 98 under the control of the CPU 100, using the result of the high pass operation F _Hi and the result of the low pass operation F _Li stored in the RAM 97 The calculation is performed using the calculation formula indicated in the column of one addition operation e _A , the result is stored in the RAM 97, and the process proceeds to step 18.

In step 18, the process waits until the next interruption stop timing, and when the next interruption stop timing arrives, the procedure from step 1 to step 18 is executed again.

22 to 24 illustrate signal processing when the waveform of the excitation current is changed. Since it is basically the same as that of FIGS. 17-22, the difference with these is demonstrated.

In FIG. ₂₂ , the waveform of the timing signal S ₂₂ is different from the waveform of FIG. For this reason, the waveform of the excitation current F _f2 is different from the waveform of FIG. 18, and it becomes the addition type waveform which added the low frequency waveform and the high frequency waveform. Therefore, the order of signal processing is slightly different.

The difference from FIG. 19 in the flowchart art shown in FIG. 23 differs from the judgment in step 8, the calculations in steps 9, 11, 13, and 14, and step 15. FIG. It is only judgment.

The determination of step 8 and step 15 is a determination required in relation to the calculation in steps 9, 11, 13, and 14.

The calculation in step 9 is performed by using the data stored in the RAM 97 by the calculation formula indicated in the high frequency demodulation operation e ' _{Hi of} FIG. 24 stored in the ROM 98 (m). Is calculated at the timing shown in Fig. 2), and the result is stored in the RAM 97 again via the bus 99.

The calculation formula in step 11 is computed using the data stored in RAM 97 by the calculation formula shown in the high-pass operation F ' _{Hi of} FIG. 24 stored in ROM 98, and as a result, Is stored in the RAM 97 again via the bus 99. The operation displayed in step 13 is the data stored in RAM 97 by the operation expression shown in the low frequency demodulation operation e ' _{Li of} FIG. 24 stored in ROM 98, ..., L ₀ ' ( n-1), L ₀ '(n), L ₀ ' (n + 1), .... L ₁ '(n-1), L ₁ ' (n), L ₁ '(n + 1), use .... is calculated at a timing shown in Fig. 22 (n), a result (e _'Li) is via a bus 99 and stored in the re-RAM (97).

The calculation in step 14 is performed using the data stored in RAM 97 by the calculation formula shown in the low pass operation F ' _{Li of} FIG. 24 stored in ROM 98, and as a result, Is stored in the RAM 97 again via the bus 99.

In step 16, the results of the high filter operation F ' _Hi and the low filter operation F' _Li are added and these are added to obtain an addition output e _O.

In the above description, the frequency of the excitation current has not been described. However, since the excitation flowmeter handles a minute signal, the power input overlaps the signal voltage. Therefore, the frequencies on the high frequency side do not match the integer multiples of the commercial frequency, and the frequencies on the low frequency side are selected to be even (even) of the commercial frequency, respectively. The adsorbate is generated and removed by the low pass filter at the rear stage.

As described above, according to the present invention, by combining the output of the low frequency signal processing system including the low frequency filtering means and the output of the high frequency signal processing system including the high frequency filtering means as synthesis means for synthesizing by the addition means, It is possible to provide an electromagnetic flowmeter that responds to the lawyer quickly and is not affected by flow noise and has a stable zero viscosity.

Claims (17)

In an electromagnetic flowmeter for applying a magnetic field to a fluid to be measured and measuring the flow rate thereof, an excitation means (12, 14, 16/12, 65) for supplying a magnetic field having two different frequencies of a second frequency lower than this of the first frequency (65). And first demodulation means 31 / Q ₄ , SW ₉ , SW _{10 /e.Hi} (step 1 / Q ₄ , SW ₉ , SW ₁₀ /e.Hi) for discriminating and outputting the signal voltage generated by the excitation means corresponding to the flow rate according to the first frequency 9), e _Hi '(step 9), high frequency filtering means 34 / F _Hi (step 11), F _Hi ' (step 11) / 88) for high-pass filtering the output of the first demodulation means, and the signal Second demodulation means (22 / Q ₃ , SW ₇ , SW ₈ / e _Li (step 13), e _Li ′ (step 13)) for discriminating and demodulating the voltage according to the second frequency. sum using the low pass filter means for low-output aftermath (24 / F _Li (step 14), F _Li '(step 14) / 87) and the high-band wake means and each output of said low pass filter means for synthesizing logarithmically Means (25,39 / e _A (step 16), an electronic flow meter, characterized in that it includes the e _O (step 16). 2. The apparatus according to claim 1, further comprising noise detection means (71, 72, 73) for detecting the fluctuation of the signal voltage from the output of said second demodulation means, an output of said combining means (25) and an output of said second demodulation means; And a rate control means (77) for changing the addition ratio by the output of the noise detection means. 2. The apparatus according to claim 1, wherein the comparing means (80) for comparing the absolute value of the output of said second demodulating means with a level setting voltage, and when said signal voltage is less than or equal to a predetermined value, switches to said second demodulating means and sets a predetermined value. And a switching means (SW ₁₁ ) for switching to said synthesizing means when exceeded. 2. The apparatus according to claim 1, wherein the comparison means (78) for comparing the absolute value of the time variation of the output of the second demodulation means with the level setting voltage is switched to the second demodulation means when the signal voltage is less than or equal to the predetermined value. And a switching means (SW ₁₁ ) for switching to said synthesizing means when a predetermined value is exceeded. 5. An electromagnetic flow meter according to claim 4, wherein hysteresis means (79a) is provided on an output side of said comparison means. 2. The apparatus according to claim 1, wherein the first demodulation means (step 9 (e _Hi , e _Hi ')), the high frequency filtering means (step 11 (F _Hi , F _Hi ')), and the second demodulation means (step 13). (e _Li , e _Li ')), the low pass filtering means (step 14 (F _Li , F _Li' )) and the synthesizing means (step 16 (e. _A , e _O )) as a calculation means using a microcomputer Electron flow meter, characterized in that. The flowmeter according to claim 1, wherein the sum of the transfer function of the high pass filter means and the transfer function of the low pass filter means is selected to be approximately 1. The electromagnetic flowmeter according to claim 1, wherein the first frequency is selected as a frequency which does not coincide with an integer multiple of the commercial frequency, and the second frequency is selected as one of the even fractions of the commercial frequency. The method of claim 1, wherein the high-band wake means 34 or the low pass filter means gain composition means (Q ₅₎ the high-Filter means 34 to which the gain control so that the I whole size of each output of the 24-side Or an electronic flow meter mounted on the low-pass filter means (24). The variable amplifier means 81 which variably amplifies the signal voltage and outputs the signal voltage to the first and second demodulation means, and a rear end of the synthesizing means. Compensation amplification means 82 for compensating for this change in amplification degree, and noise detection means 83 for detecting any of the rate of change or magnitude of noise contained in the signal voltage. And an amplification degree of the variable amplifying means (81) and the compensation means (82) by an output. 2. A rate limiting means (step 10) is installed between the first demodulation means (step 9) and the high frequency filtering means (step 11) to limit the variation of the signal voltage to a predetermined modulation width. An electronic flow meter, characterized in that. The time constant of the high frequency filtering means (88) and the low frequency filtering means (87) is made small by detecting the time when the noise contained in the signal voltage falls within a predetermined allowable range. After that, the noise detecting means 89 outputs a time constant control signal having a large time constant, and the time constants of the high frequency filtering means 88 and the low frequency filtering means 87 are changed by the time constant control signal. An electronic flow meter comprising a time constant changing means (86). The power supply detector 94 according to claim 1, which detects a time point at which the power is turned on and outputs a time constant control signal having a small time constant from a time point at which the power is turned on, and the time constant control signal is used to output the time constant control signal. And a time constant changing means (86) for changing a time constant of the high pass filtering means (88) and the low pass filtering means (87). An electromagnetic flowmeter for applying a magnetic field to a fluid to be measured and measuring the flow rate, comprising: excitation means (12, 14, 16) for supplying a magnetic field having two different frequencies of a first frequency and a lower second frequency; Signal processing means (46, 47, 54) for discriminating the signal voltage generated by the excitation means and corresponding to the flow rate according to the first frequency and outputting the signal voltage as the first signal, the voltage related to the first signal and the The flow rate signal is corrected using zero detection means 44, 53, 58 which take a deviation from the second frequency component of the signal voltage and detect it as a zero signal, and the zero signals E ₁ , E ₂ , E ₃ . An electronic flow meter comprising zero correction means (42, 48, 55). An electromagnetic flowmeter for applying a magnetic field to a fluid to be measured and measuring the flow rate, comprising: excitation means (12, 14, 16) for supplying a magnetic field having two different frequencies of a first frequency and a lower second frequency; Second demodulation means 22 for discriminating and outputting the signal voltage generated by the excitation means corresponding to the flow rate according to the second frequency, and low pass means 24 for low-pass filtering the output of the second demodulation means. And first demodulation means 31 for discriminating and demodulating the signal voltage according to the first frequency, and detecting a deviation between the output of the first demodulation means and the output of the low pass filter (low pass filter). Response correction means (60) for calculating the difference between the response compensation signal (60) and the response compensation signal obtained by outputting the response amplification means via the high pass filter (61) and the low pass filter (24); Electron flowmeter characterized in that provided. An electromagnetic flowmeter for applying a magnetic field to a fluid to be measured and measuring the flow rate, comprising: excitation means (12, 14, 16) for supplying a magnetic field having two different frequencies of a first frequency and a lower second frequency; Second demodulation means 22 for discriminating and outputting the signal voltage generated by the excitation means corresponding to the flow rate according to the second frequency, and low pass filtering means 24 for low-pass filtering the output of the second demodulation means. ), Response detection amplifying means 63 for amplifying a deviation between the flow rate output and the signal voltage, and the output of the response detecting amplifying means is a response compensation signal Vc 'obtained by demodulating according to the first frequency and the An electromagnetic flowmeter, comprising a response correction means 64 for correcting the response using the output of the low-pass filtering means 24, and outputting a signal related to the output of the response adjusting means 64 as the flow rate signal. . An electromagnetic flowmeter for applying a magnetic field to a fluid to be measured and measuring the flow rate, comprising: excitation means (12, 14, 16) for supplying a magnetic field having two different frequencies of a first frequency and a lower second frequency; The first demodulation means 31 which discriminates the signal voltage generated by the excitation means corresponding to the flow rate according to the first frequency and outputs the first output, and the mutual voltage is discriminated according to the second frequency The second demodulation means 22 for demodulating, the low pass filtering means 24 for low-passing the output of the second demodulating means with a large time constant and outputting a second output, and the first and second outputs An electromagnetic flowmeter, wherein the flow rate output is output by performing a predetermined operation.

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