KR910000880B1 - Flux control for induction motor drive using load commutated inverter circuit - Google Patents

Abstract

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Description

Induction motor operation control system and its control method

1 is a diagram of a suitable embodiment of a power circuit for supplying power to an induction motor in accordance with the present invention.

2 is a graph of typical LCI system requirements for a VAR as may be supplied by the system of the present invention.

3, 4 and 5 are vector diagrams useful for understanding the present invention.

6-10 are diagrams of variable or controllable VAR generators applicable to the system of the present invention.

11 is a diagram illustrating the invention in its overall basic control concept.

12 is a diagram illustrating, in a suitable embodiment, the control of a variable VAR generator of the system of the present invention.

13 and 14 are detailed diagrams illustrating possible devices of the system part of FIG. 12 shown in block form.

15a, 15b, 15c, 16a, 16b, 18 and 19 are useful vector diagrams for understanding FIG.

17 is a basic block diagram of another suitable embodiment of the present invention.

20 is a basic block diagram of another suitable embodiment of the present invention.

21 is a basic block diagram showing a digital device of the present invention.

* Explanation of symbols for main parts of the drawings

10: Load Switching Inverter Circuit (LCI) 11: AC / DC Converter

12: DC / AC converter 14: DC link

16 inductor 24 induction motor

25 winding 30 capacitor circuit

40: VAR generator

The present invention relates generally to the control of an electric motor, and more particularly to an apparatus for controlling an induction motor using a load commutated inverter (LCI) for supplying power to the electric motor.

In motor drive, it is common to use a power converter system to provide power to the motor from a source. There are various types of power converters, but when used at high voltages and power, the converters are often comprised of thyristors in bridge arrangements. Bridged thyristors are selectively gated to vary or control the power supplied to the motor, which is commonly known as phase control. In an AC motor drive with adjustable speed, two converters are commonly used, the first of which converts alternating current (AC) power from a source (i.e. power line) into direct current (DC) power. Used. The second converter connected to the first converter by the DC link circuit is used to convert the supplied DC power into variable frequency AC power supplied to the motor. The first converter (supply source side) is controlled to vary the amount of current supplied to the motor, while the second converter (load side), commonly called an inverter, is used to vary the frequency of power supplied to the motor.

The source-side converter for a typical three-phase system consists of six thyristor bridges whose phase is controlled to vary the output current or voltage. The load side converter, or inverter, is composed of either a load switching inverter or a forced switching inverter. Switching here means making the thyristor of the bridge non-conductive.

As is known in the art, in order to non-conduct a die thruster, the current in the die thruster must be reduced to a value of zero by applying a reverse voltage to the die thruster. This is usually not a problem for the source side converter. This is because the converter basically acts as a rectifier, and the gating on of one die Lister has the effect of switching the challenging die lister since it develops a reverse voltage to the AC source (i.e. power line) to which the converter is connected. However, in a load-side converter supplied with DC power, some reactive-volt-ampere (VAR) must be present to achieve switching. In the load switching inverter, the VAR is derived from the load. In a force-switched inverter, there is a device such as a capacitor, in which charge is appropriately charged and charges charged in the capacitor at an appropriate time are used to switch the inverter.

In driving a variable speed AC motor, a low voltage (i.e., 100 volts or less) is appropriately driven by using a cage type induction motor, and the induction motor of this type is combined with a drive wave inverter, a pulse width modulated inverter or a current source inverter. It is supplied by the same forced switching inverter. Induction motors are suitably used here because they are simple and robust. The advantage of the induction motor is to overcome the disadvantage that the forced switching inverter must have a switching circuit. However, the technique is usually limited to inverters in which a single main or power thyristor switching device is provided in each inverter leg portion. This is because of the technical difficulties associated with parallel or series thyristors that require a higher strategic cost. By way of example, it is very difficult to forcibly and simultaneously forcibly switch over a number of diistors contained in series in one leg portion of a bridge in which the required voltage exceeds the voltage required for a single thyristor. Therefore, it is practical to use a load switching inverter to supply synchronous motors at higher voltages (ie, 100 volts or more).

Load-switched inverter systems having both the source-side and load-side converters described above are considered capable of properly passing the effective current from one AC source to another (or load-side) while extracting delayed reactive current from the source and load side. do. Therefore, in driving an electric motor, it is usually used to drive a synchronous motor other than an induction motor. This is because a synchronous motor having a rotor excited by a DC current is used as a delayed reactive current source for the load side of the LCI to effectively convert. LCI is not commonly used as a suitable source for driving induction motors. This is because the motor requires a delayed invalid current source at the terminal of the motor like the LCI. Thus, while load-switched inverters or LCIs are cheap and able to handle high voltages, they cannot be suitably used in simpler and cheaper induction motors.

Accordingly, it is an object of the present invention to drive an AC induction motor using a load switching inverter circuit.

Another object of the present invention is to drive an induction motor by using a combination of a load switching inverter circuit for supplying current to the motor and a capacitor circuit for supplying a predetermined reactive current or VAR to the system and a variable VAR generator. will be.

Another object of the present invention is to drive an induction motor using a combination of load switching inverter circuit and capacitor circuit and a variable VAR generator, wherein the load switching inverter and variable VAR generator switch the load switching inverter circuit. It is controlled to maintain the desired motor torque and flux and the required VAR to supply the appropriate exciting current to the motor.

The above and another object is to provide a load-switched inverter circuit comprising an AC induction motor having a winding powered from a multiphase source, a controllable AC / DC converter connected to the source and a DC / variable frequency AC converter connected to the motor. By providing a system connected to said source via LIC). The two converters are connected to each other by a DC link circuit. The first feedback control passage in response to the command signal and the feedback signal indicative of the motor operating parameters is used to control AC / DC (source side) operation, while the second feedback passage in response to the command signal is connected to the motor. Used to control the operation of the DC / AC (load side) converter.

On the other hand, the fixed capacitor circuit is connected between the motor windings to provide an advanced VAR to the system, and the variable VAR generator connected between the motor windings supplies an additional VAR to the entire system in response to the VAR command signal. The VAR command signal is generated by the third feedback control passage and responds to defined operating parameters of the LCI circuit and the motor.

Hereinafter, the present invention will be described in detail with the accompanying drawings.

1 shows a power circuit for driving an induction motor according to a general preferred embodiment of the present invention.

The power source (three phases) indicated by terminals L ₁ , L ₂ and L ₃ is connected to the load switching inverter circuit shown by number 10. The load switching inverter circuit LCI includes a source side converter 11 connected to the source side converter 12 by a DC link 14 including an inductor 16 to supply a DC IDC. Each of the converters 11 and 12 has six thyristors connected in a bridge arrangement in a known manner as shown. It should be noted that each of the six die Listers is intended to represent a single or multiple die Lister connected in parallel or in series depending on the power requirements of the entire system. Induction motor 24 with winding 25 is connected to LCI 10 by three connecting lines 18, 20, 22 such that controlled current I _L is supplied to motor 24, in particular the load-side inverter ( 12 is shown as the wire connection of the motor windings, but the same as the delta connection can be used as well. The primary function of the LCI 10 is to provide effective power to the induction motor 24, but as can be appreciated in some cases, there is also a delayed excitation current component at the output of the LCI.

The capacitor circuit 32 connected to the lines 18, 20 and 22 is composed of three capacitors 34, 36 and 38 connected by wire connection. These capacitors may be connected in delta connection depending on engineering practicality. Although the capacitor is shown as having a fixed value, a switched capacitor circuit with various capacitors switched in the circuit as a function of the operating parameters of the motor, eg speed, may be used. The main function of the capacitor circuit is to provide a leading reactive current to excite the induction motor, to provide a portion of the required VAR and to perform the switching of the LCI 10. The VAR generator 40 is also connected to both ends of the winding of the motor. The primary function of the VAR generator is to maintain the magnetic flux level of the motor by supplying an additional reactive current that increases or decreases the current supplied from the capacitor circuit and the LCI 10.

As shown in FIG. 1, the VAR generator 40 includes a circuit 39 that is actually connected to lines 18, 20, and 22. As shown in FIG. Possible operation of the circuit 39 will be described with reference to FIGS. 6 to 10. In addition, the AC / DC power source 42 indicated by the dotted line is connected to the terminals L ₁ , L _2, and L ₃ by lines 44, 46, and 48, and suitably connects the inductor 43. Power is supplied to the circuit 39 through the included DC link (41). Although it may be desirable to have more complete control of the generator output, the power source is not required except when the VAR generator must provide active power to the entire system. As described below, although not particularly limited, the VAR generator 40 is intended to represent only the circuit 39 or to represent the circuit 39 and power source 42 together. When the power source is included, the power source 42 can be a typical bridge converter as shown by the converter 11 of the LCI 10 and the control of the two elements of the generator 40 will be described in more detail below. As shown, they are fundamentally identical to the two LCI converters.

FIG. 2 is a graph showing the excitation system control range for the appropriate motor excitation and switching VAR at typical loads. The motor torque required here is almost proportional to the square of the motor speed. 2 shows a unit speed along the abscissa and a unit VAR along the ordinate. Typical examples are shown as 1 per unit capacitor circuit and 0.25 per unit phase-to-ground VAR controller. That is, the VAR generator 40 of FIG. 1 shows that it can provide 0.25 per unit VAR at the rated voltage and frequency, either in ground or in phase form. The dashed line 50 in FIG. 2 represents the required total VAR of the system in the speed range of 0.2 to 1 per unit. The dashed line 52 represents a VAR that can be provided to the system by the capacitor circuit of FIG. When the capacitor is alone, the time to have the correct VAR in the system is approximately 0.7 per unit speed. The solid line represents the limit of possible VARs of the combined capacitor and VAR generator. As shown in FIG. 2, if the VAR generator can only supply a true VAR, it is shown that a system with a capacitor circuit as shown can only operate within the range of 0.2 to 0.7 per unit speed. Conversely, if only delay VAR can be used to supplement the capacitor circuit 32, operation is allowed in the unit speed range 0.7 to 1.0. Thus, the VAR generator can provide both fast and ground VARs such that the system demand line is between two solid lines of 0.33 to 1.0 speed. Thus, the motor can be operated in a wider speed range.

The vector diagrams of FIGS. 3, 4 and 5 in accordance with FIG. 2 are provided to facilitate understanding of the operation required for the system of the present invention. As described above, the load switching inverter of the LCI sends an active current from an AC source to the load side. Since induction motors and load switching inverter circuits require a ground reactive current source at the terminals, LCI is not considered to properly drive induction motors.

As shown in FIG. 1, in order to better understand the convention of multiple currents at the motor terminals of a basic power circuit, FIG. 3 is shown with reference. In the diagram, the reference vector is the motor air gap voltage represented by the vertical vector V _M. For easy understanding of the description, the motor losses are ignored and the motor supply voltage and the motor terminal voltage are considered equal. The motor current in the diagram can be shown at any angular position. If the motor current is in phase with the motor voltage vertically, the motor becomes a power generator. If the current is vertical and directed downward, the motor absorbs power, providing mechanical force in normal operation of the motor. If the current is 90 ° above the voltage (turned to the right in FIG. 3), the current tends to reduce the air gap magnetic flux in the motor. This is called the vertical linear current for the synchronous motor / generator. If the current is 90 ° above the voltage (turned to the left in FIG. 3), the current increases the void flux.

The synchronous motor can operate with current at any vector position on the diagram. However, induction motors are only excited from the stator terminals and therefore must be excited by a current with a component horizontal to the left (the direction to make left). Thus, the allowances for the motor and generator operation of the induction motor are most of the left half of the vector diagram as indicated and classified as "induction motor operating range".

The load-switched inverter circuit is operated by a thyristors or other controlled switching devices at a time that is delayed by a controllable amount relative to the gating time (the gating time at which the diester is conducted to the diode) causing a pre- rectification action. The gating time is typically expressed as a zero delay reference time and the delay by the time indicated by the electrical angle is represented by an alpha (α) angle. The current supplied from the LCI to the motor is delayed by α on the load side of the LCI at the phase position determined by the gating. The angle α in the vector diagram of FIG. 2 is measured clockwise (ie, in the same direction as the motor current) from the vertical reference value V _M. The reference value for each α is the same as the reference value for the motor current. This is because the operation of the LCI as a diode bridge corresponds to receiving the pure active power from the motor and operating the motor as a generator. Therefore, the normal range of LCI from α = 0 to a value slightly less than 180 ° is almost all of the lower right half of the vector diagram in FIG. From FIG. 3, it can be seen that the LCI is not appropriate to be provided to the induction motor. This is because there is no operating region of the LCI that includes the operating region of the induction motor. The exciting current component I _LE provided by the LCI is depolarizing.

As mentioned above, the entire system of the present invention has three current sources. Here, the first current source I _L comes from the LCI. The second current source is I _C , coming from the capacitor circuit 32 of FIG. 1, and the third current I _X coming from the variable VAR generator 40. Each of the currents is limited to a specific range in FIG. 3 by the characteristics of the current source and the sum of the three currents must be equal to the total motor current.

The current I _C of the capacitor circuit is horizontal to the left as shown in FIG. 3, and the phase angle increases the magnetic flux. The current angle of the accelerating switching inverter connected to the power source can be any value. The function of the force-switched inverter is that the VAR generator current I _X can be switched at any moment and operated in any vector position. In the case of a VAR generator without a separate power source, the effective power cannot be accommodated or decomposed, and the current is horizontally parallel to the left (capacitive) or right (inductive) polarity as shown in the diagram of FIG. Should be.

4 shows how the currents of three current sources are combined to meet the requirements of an induction motor. The motor current I _M is shown in a vector position representing the motor operation. The LCI current I _L has the same effective component (vertical) as I _M and is close to the position of the motor current in the vector position (within the allowable range of the LCI current). The capacitor circuit current I _C is in the allowable vector position, and when it is sized by the voltage and frequency of the motor, it has the following relationship.

Where V _M = motor line for neutral voltage

W = motor stator frequency

C = effective line for neutral capacitance

Capacitor current vector I _C is shown in FIG. 4 at two positions on the vector diagram. One is shown in its original position and the other is shown in combination with I _L at the bottom of the diagram. The current I _X from the VAR generator is in the horizontal direction and has the magnitude and polarity to complete the vector diagram so that the current required for the motor at the required torque, I _M ,

Becomes 4 shows the case where the capacitor current is too small to supply the required reactive current.

5 shows vector positions for various currents when the capacitor circuit is very large. In this case, control of the VAR generator 40 reverses the phase position and adjusts the amplitude to provide the reactive current where the sum of the three currents is required as before. Since the magnitude of the capacitor current is determined by Equation (1) above, and is represented as a function of the motor voltage and the motor speed, there is no way to directly control the magnitude of the capacitor current. Thus, the variable voltage VAR generator current is adjusted at each operating point, causing an error in reactive current. Thus, the optimal choice for the capacitor circuit minimizes the power requirements of the VAR controller.

6-10 show with reference to possible configurations for the circuit 37 and variable VAR controller 40. In each case, the illustrated VAR controller is connected to lines 18, 20 and 22. In each case, multiple switching or gating elements of various controllers are placed under the control of appropriate control means, as described above, so that the amount of VAR provided may vary depending on the capabilities of the various circuits.

6 shows a first possibility for the VAR generator circuit 39. The circuit shown relates to a line switching converter having six thyristors in a three phase three leg arrangement. The connections by lines 18, 20 and 22 between the thyristors 60 and 61, 62 and 63 and 64 and 65 are made of the thyristor junction in each of the three legs. The DC conductor of the conventional converter bridge is connected to the terminating impedance inductor 66. The form of FIG. 6 can only provide ground VARs, and the amount of VARs supplied is represented by the gyring function of the dyistor.

7 shows a capacitor circuit switched as a possible VAR generator. It should be appreciated here that the VAR generator can only provide true VARs. As shown, three lines 18, 20, and 22 are connected to the delta circuit arrangement, each leg of the arrangement being in the counterbalance of a gating element, such as, for example, the thyristors 72 and 73. Connected. By selectively gating or challenging the thyristors of the multiple leg portions of the delta arrangement, the bridge or generator of FIG. 7 can supply a variable phase VAR to the system.

8, 9 and 10 illustrate a number of forced conversion converters that can provide fast and ground VARs. The converter is considered to be more preferred for use in the present invention where motor operation is required in an extended range. 8 shows a control switching inverter or an inverter which is switched continuously automatically. Each of the three legs with respect to the three phases is arranged in series and includes a first die Lister 74, a first diode 75, a second diode 76, and a second die lister 77. Each of the three legs has a first diode, a first diode 75, a second diode 76, and a second die Lister 76 arranged in series with each other having a polarity to be conducted in the same direction. Capacitors, such as capacitors 78 and 79, are switched between the thyristors of adjacent legs. The bridge device, similar to the device shown in FIG. 3, ends at an inductive reactance 80.

8 illustrates the form included for the sake of completeness. The VAR generator of this type is connected to lines 18, 20 and 22 by a suitable isolation transformer 82 having a Y-type primary winding 84 and a delta-type secondary winding 86. This is intended to represent another variant of the connection that is included only for understanding specific cases where isolation and voltage level adjustments are required in accordance with good design practicality and can be applied to all other embodiments.

9 is a forced switching inverter of the second type, also commonly referred to as a McMurray inverter. The circuit and its operation are described in more detail in US Pat. No. 3,207,974, entitled “Inverter Circuit,” registered September 21, 1965. The inverter shown in three phases has three identical parts and only the part connected to the line 18 is shown in the figure. The inverter leg portion includes a first die Lister 88 connected in series with a second die lister 86 between DC booths. The two thyristors are commonly called power transistors. The pair of diodes 90 and 92 are connected in parallel with the thyristors 86 and 88 and are reversed in polarity. Two additional thyristors 94 and 96, referred to as current dielisters, have a capacitor 98 and an inductor 100 between the cathode of the dielist 94 and the anode and cathode of the diode 90. And when connected in series to the thyristor 86, are connected between the DC booths. By proper gating of the plurality of thyristors, the capacitor 98 is properly charged and prevented to convert the leg portion thyristors in the manner known and described in the patents mentioned above. The forced switching inverter ends on the DC terminal in the capacitor 102.

10 is another type of forced switching inverter. The inverter is commonly referred to as a gate turn off inverter and can be operated in pulse width modulation or square wave mode. The symmetric three-phase form is shown again, where a first gate turn-off switch 110 connected in reverse parallel with the diode 112 is included in one leg. In a similar form, the negative portion of the leg includes a gate turn off switch 114 connected in reverse parallel with the diode 116. The inverter ends on the DC terminal in capacitor 120.

In the division, the signal marked with * is the required value or command value obtained, and the signal without * is the measured value or calculated value. As an example, I _L ^* represents the value of the required LCI output current while I _L is the actual LCI output current.

11 shows a system showing the basic form of the present invention. A power unit as shown in FIG. 1 is included with the associated control unit. Similarly, the supply side converter 11 is connected to a three-phase AC power source indicated by terminals L ₁ , L ₂ and L ₃ . In addition, the supply side converter 11 is connected to the load side converter 12 by a DC link 14 including an inductor 16. The induction motor 24 is connected to the load side converter 12 by the conductors 18, 20, and 22. As shown in FIG. Capacitor circuit 30 includes capacitors 34, 36, 38 and is connected through windings of motor 24 via lines 18, 20, 22. The VAR generator 40 is connected in parallel with the capacitor circuit 32 through the motor winding.

를 가지는 속도 조정회로(139)에 공급된다. 회로(139)의 출력은 전동기 작동 명령 신호인데, 여기서는 T^*로 나타내지며, 토오크 명령신호이다. 공급원측 콘버터에 있어서, 다이리스터 점화를 제어하는 상부 제어 통로를 먼저 살펴보면, T^*신호는 전류 기준 회로(140)에 인가되며, 상기 회로는 출력으로 신호 I_L ^*을 공급하기 위해 한계치를 가지는 절대치 회로이다. 상기 출력 신호는 명령 입력 신호로 사용되어 제어회로(142)를 조정하며 점화시켜서, 라인(144)상에 출력 신호를 제공하여 콘버터(11)의 다이리스터의 점화를 제어한다.Three feedback control passages responsive to the motor operation command signal are associated with the illustrated power circuit and used to control the source side, converter 11, load side inverter 12 and VAR generator 40. Thus, according to a suitable embodiment of the present invention, a feedback signal proportional to the actual speed of the motor is supplied first. The signal is represented by the stator frequency W as described below, but what is described here is the signal N extracted from the tachometer 130, which is connected to the motor 24 by a dashed line 132 as shown. It is. The tachometer 130 supplies an output signal N on line 130 that is proportional to the actual speed of the motor. The speed signal N represents a predetermined motor speed and is summed in the summation junction 136 together with the signal N ^* extracted from suitable means such as the operator control section 138. The difference between N and N ^* signals determined by junction 136 is the transfer function

It is supplied to the speed adjusting circuit 139 having a. The output of the circuit 139 is an electric motor operation command signal, which is represented by T ^* here and is a torque command signal. In the source-side converter, first looking at the upper control passage that controls the thyristors ignition, the T ^* signal is applied to the current reference circuit 140, which has an absolute value with a threshold for supplying the signal I _L ^* to the output. Circuit. The output signal is used as a command input signal to adjust and ignite the control circuit 142 to provide an output signal on line 144 to control the ignition of the thyristor of converter 11.

In a similar manner, a T ^* signal is applied to the phase angle reference circuit 146, which is a shaping amplifier with a threshold and provides an output signal α ^* . The α ^* signal is used as a command input and applied to the ignition control circuit 148. The output signal on line 150 controls the firing of the thyristors of load-side converter 12. Control of these two passages to the controllers of the load side and source side converters is known in the art. For a more complete description of this, a patent was filed on May 15, 1984 entitled "Flux Feedback Ignition Control for Load Change Inverters," and was filed by Applicant David L. Repeat et al., US Pat. No. 4,449,087. The patent is described herein by reference.

에 응답하여 VAR 발생기(40)의 작동을 제어한다. 제11도에 있어서의 스위치(158)와 관련되어 회로 노드(157)에서

신호를 제공하는 두 방법이 도시되어 있다. 제1방법은 예정된 값의

기준 신호를 입력 단자(159)에 공급하며, 스위치(158)가 하부 위치에 놓여질 때 상기 신호는 노드(157)에서

신호로 사용된다. 제2방법은 속도 조정기(139)로부터의 T^*신호를 입력으로 하며, 작동 블록(152)에 있어서 스위치가 상기 위치에 놓여질 때, 노드(157)에 나타나는

신호를 발생시킨다. 작동 블록(152)은 비선형 이득 및 한계치를 가지는 절대치 회로로 나타나 있다.

신호는 두 통로를 통해 인가되어서 전동기 여자 전류 I^* _ME에 대한 명령을 하게 된다. 상기 두 통로는 함께 자속조정기(161)를 형성한다. 제1통로는 블록(163)을 통과하는 피드-포워드 통로이다. 제1근사치로서, 여자전류 I^* _ME는 상기 통로에 의해 전동기 여자 리액턴스 X_M에 대한 자속 명령비로 세트된다. 제2통로는 블록(164)을 통과하는 궤환 통로이며, 여기서

신호가 합산 정션(160)에 한 입력으로서 인가되며, 상기 통로의 제2입력은 실제 전동기 자속의 절대치를 나타내는 신호 |

| 신호이다. |

|신호의 발생은 전동기와 연관된 자속 코일로부터 추출됨으로 변환될 수 있다. 그러나, 도시된 실시예에 있어서, |

|신호는 회로(162)에 의해 발생되며 상기 회로는 전동기 단자 전압을 합산하며 정수의 절대치의 합을 결정하여 실제 자속의 신호 표시를 나타낸다(

신호를 발생시키는 제2방법은 제14도에 더욱 상세하게 도시되어 있다).The third feedback passage is a passage of primary interest in the present invention. The passage is a magnetic flux command signal indicative of a predetermined motor magnetic flux level.

In response, the operation of the VAR generator 40 is controlled. At the circuit node 157 in association with the switch 158 in FIG.

Two ways of providing a signal are shown. The first method is to

A reference signal is supplied to the input terminal 159, which is at node 157 when the switch 158 is placed in the lower position.

Used as a signal. The second method accepts a T ^* signal from the speed regulator 139 and appears at node 157 when the switch is placed in this position in actuation block 152.

Generate a signal. Operation block 152 is represented by an absolute value circuit with nonlinear gain and threshold.

The signal is applied through two paths to command the motor excitation current I ^* _ME . The two passages together form a flux regulator 161. The first passage is a feed-forward passage through block 163. As a first approximation, the excitation current I ^* _ME is set by the passage to the magnetic flux command ratio for the motor excitation reactance X _M. The second passage is the feedback passage through block 164, where

Signal is applied as an input to the summation junction 160, the second input of the passage being a signal representing the absolute value of the actual motor flux |

| It is a signal. |

The generation of the signal can be converted into extraction from the magnetic flux coils associated with the motor. However, in the illustrated embodiment, |

The signal is generated by circuit 162 which sums the motor terminal voltage and determines the sum of the absolute values of the integers to represent the signal representation of the actual magnetic flux.

The second method of generating the signal is shown in more detail in FIG. 14).

, |

|신호 사이의 차는 블록(164)에 인가되며, 상기 블록(164)은 속도 조정기(139)와 동일한 타입의 전달 함수를 가진다. 상기 함수는

로 나타낼 수 있다. 자속 조정기의 출력은 속도 조정기(139)와 동일한 타입의 전달함수를 가진다. 상기 함수는

로 나타낸다. 자속 조정기의 출력은 적절하게 전동기를 작동시키기 위해 전동기 전류의 소정의 여자성분에 비례하는 명령 신호 I^* _ME가 된다. 상기 성분은 제3도에 도시된 바와같이 공급자속과 동상이 된다. I^* _ME신호는 한 입력으로서 3입력 합산 정션(166)에 인가되며, 인가되는 제2입력은 I_LE로 표시되는 신호이다. 상기 I_LE신호는 함수블록(168)의 출력이며, 상기 블록은 전류 조정기(140)로부터의 신호 I_L ^*과 위상각 조정기(146)로부터의 사인 α^*의 적을 제공한다. 상기 I_LE는 부하 전환 인버터(제3도 참조)에 요구되는 무효 전류를 나타낸다. 합산 정션(166)에 대한 제3입력은 신호 |I_C|이며, 상기 신호는 캐패시터 회로(32)에 의해 시스템에 제공되는 전류의 절대치에 비례한다. 상기 전류 신호는 후술하는 바와같은 어떤 적절한 방법으로 추출될 수 있다. 또, 제11도에 도시된 바와같이, 캐패시터 회로(32)를 전동기에 접속시키는 라인의 전류를 감지하는 3개의 전류 변압기(170,172,174)를 사용하여 감지된 값으로부터 생성시킬 수도 있다. 센서 출력은 함수 블록(176)에 접속되어 있으며 상기 블록은 각 캐패시터 전류의 절대치를 결정하며, 상기 블록의 출력인 |I_C| 전류 신호를 제공하는 상기 값을 합산한다.Taken from junction 160,

, |

The difference between the signals is applied to block 164, which has a transfer function of the same type as the speed regulator 139. The function is

It can be represented as. The output of the flux regulator has the same type of transfer function as the speed regulator 139. The function is

Represented by The output of the flux regulator becomes the command signal I ^* _{ME which} is proportional to the predetermined excitation component of the motor current in order to properly operate the motor. The component is in phase with the supplier as shown in FIG. The I ^* _ME signal is applied to the three input summing junction 166 as one input, and the second input applied is the signal represented by I _LE . The I _LE signal is the output of the function block 168, which provides the product of the signal I _L ^* from the current regulator 140 and the sine α ^* from the phase angle regulator 146. _ILE represents the reactive current required for the load switching inverter (see FIG. 3). The third input to summing junction 166 is signal | I _C |, which is proportional to the absolute value of the current provided to the system by capacitor circuit 32. The current signal can be extracted in any suitable manner as described below. In addition, as shown in FIG. 11, the capacitor circuit 32 may be generated from the sensed values using three current transformers 170, 172 and 174 which sense the current in the line connecting the motor. The sensor output is connected to a function block 176, which determines the absolute value of each capacitor current, which is | I _C | The above values providing the current signal are summed.

The output of junction 166 is I _X ^* , which is a command for the VAR generated current. The regulator and ignition control 180 provide a signal on line 182 to control the gating of VAR generator 40 to provide appropriate VAR correction as described above. That is, by the signal value from line 166, the VAR generator is controlled to adjust the ignition time (phase angle) of the thyristors or gates of the appropriate VAR generator to control the excitation current I _X (VAR).

In this respect, the present invention is to control the generation of VAR as a function of the motor flux. Although not specifically shown in the figure, the feedback signals I _LE and | I _C | may be omitted to connect the output of the flux regulator directly to the regulator and the ignition controller 180 to control operation. The problem with this simplified system is that it is subject to the positive feedback effect of the capacitor circuit 32 by the motor excitation. The feedback effect can be immediately explained by observing that the current from the capacitor circuit 32 that generates the motor magnetic flux is directly related to the motor voltage, as can be seen in equation (1). Increasing the motor voltage causes an increase in the capacitor current to increase the motor excitation current. An increase in the excitation motor current causes an increase in the magnetic flux, further increasing the motor voltage. In other words, a positive feedback is generated that creates a condition of instability. From the models of the motor 24 and capacitor circuit 32 connected as shown, it can be seen that the positive feedback loop for the motor exciting current I _ME has the overall gain given by the following equation.

Where L _M is the equivalent magnetization inductance of the motor.

W ₁ is the stator frequency that is proportional to the speed of the motor plus slip.

C is the effective capacitance of the capacitor circuit 32.

S is the Laplace conversion operator.

t ₂ is a time constant that typically has a value of about 1 second.

t ₁ is typically a time constant shorter than t ₂ .

From the above equation, it can be seen that the positive feedback is proportional to the square of the velocity. At low speeds the effect of the loop is not critical, and without the control the motor is not excited. At a particular speed, i.e., in the range of 0.5 per unit, the total loop gain reaches one. The speed is the speed with respect to the motor self-excitation by the capacitor circuit. At higher speeds, the positive feedback excites the motor to saturation level. At lower speeds the magnetic flux attenuates without the VAR generator.

I _LE and | I _C | shown in FIG. The signal deviates from the positive feedback effect and generates a signal I _X * representing the amount of current required from the VAR generator to generate a commanded motor excitation current, ie the magnitude of the excitation component I _ME of the motor current. Changes in certain factors that determine the excitation component of the motor current cause a change in the motor flux that is delayed by the motor excitation time constant t ₂ . Thus I _LE and | I _C | By including the feedback signal, a change in the excitation component of the current is detected, and the correction is carried out before the motor flux changes significantly and the positive feedback loop described above is broken. Since the time constant t ₂ is about 1 second, the control loop should be faster than approximately 1 radian per second. This is usually required at the current technical level.

12 illustrates one form of a variant embodiment of the invention, which is the same general type of system as FIG. The general part here is duplicated in FIG. The control of the load switching inverter, i.e., the converters 11 and 12, is the same as the method described above. The difference in FIG. 12 with respect to FIG. 11 lies in the control of the VAR generator 40.

In particular, in FIG. 12, the load side converter 12, the DC link inductor 16, the lines 18, 20 and 22, the induction motor 24 and the capacitor circuit 32 are shown. VAR generator 40, suitably of the forced switching type, supplies variable excitation current I _X to lines 18, 20 and 22, and thus to motor 24. As shown in FIG. In a manner similar to that of FIG. 11, the flux regulator 161 is provided and the output command signal I _ME ^* as an input is supplied to the summation junction 200. The second input supplied to the summation junction 200 is a signal extracted from the I _ME resolver 202 and represented by I _ME on the line 210. The resolver 202 receives a signal indicative of the terminal voltage of the motor 24 via line 204. The resolver 202 also receives a signal indicative of the phase rectification supplied to the motor through suitable sensing elements such as current transformer 206 and line 208. The I _ME output of the resolver 202 represents the magnitude of the actual motor excitation current.

The output of the summation junction 200 representing the difference between the predetermined motor excitation level and the actual motor excitation level is applied to a simple integration function (block 212), the output of which is a signal I _X representing the amount of excitation current. ^{Is *} . The VAR generator 40 here generates a signal to properly excite the motor. The I _X ^* signal is applied to two function blocks, which are composed of an adjuster and an ignition controller 180 of the regulator. Decoupling the adjuster is a general form of adjustment, the control of which is suitable for use when the VAR generator is supplied with a separate power source indicated by dashed line 42 shown in FIG. The I _X ^* signal is supplied to a function block 214 which is essentially an absolute value circuit, which now provides a signal to adjust the output magnitude of the VAR generator 40 to ignite the controller 218. The detection (polarity) of the VAR generator output is determined by the output of the function block 216, which provides an output representing +90 or -90 depending only on the polarity of the I _X ^* signal. The signals from both blocks 214 and 216 are applied to the ignition control 218 which, in response, controls the ignition angle of the VAR generator thyristors. The VAR generator output is therefore adjusted as a function of the actual excitation current I _ME of the motor.

The I _ME control loop as shown in FIG. 12 does not include any internal motor parameters except for the small effect of motor leakage reactance, and the response speed is limited only by the VAR generator itself. Thus, it is easy to select a gain constant Ke at block 212 that causes a fast settling loop to maintain a predetermined excitation current level in the motor. In essence, the advantage of the control loop for adjusting the motor excitation current is to break the positive feedback loop by correcting the error upon excitation before the error occurs in the magnetic flux.

Although described in the above embodiments, the output of the flux loop, not shown, is defined as the current command for the VAR generator. However, the flux is not directly proportional to the excitation current of the VAR generator. It is proportional to the motor's excitation current (proportional to self saturation). Therefore, if the output of the flux regulator is defined by the command I _X to the VAR controller 40, the control is nonlinear. However, if the output of the flux loop is defined as a command for the motor excitation current I _ME ^* and the excitation current is measured for feedback (twelfth) and the error is defined as I _X ^* , then the flux control is linearized.

Similarly, the flux loop output consists of the motor excitation current command I _ME ^* , and the command for I _X ^* corrects all other components of I _ME as shown in Figure 11 and _sends the rest of I _ME ^* to the VAR controller. The flux loop is linearized if generated by defining the command I _X ^* .

The loop of FIG. 11 or FIG. 12 reduces the forced or dynamic range required for variable VAR generator 40. When some change occurs, for example, when a change occurs in the excitation component of the motor current from the LCI, the motor flux starts to change at a time constant of 1 second. If the change is not immediately corrected and remains until the change in the motor flux can be detected by the flux loop, the correction of the flux error is required to correct three components of the motor excitation current. The first component required for calibration is a change in the electric motor excitation current I _ME . The second component required for the calibration is for the capacitor current that changes because the magnetic flux is allowed to change. The third component is to provide a transition forced current to satisfy the flux level. Since the detection of the three components is usually the same, these components are combined to determine the dynamic range required for the VAR generator. If all errors are errors that are motor excitation current errors and it is detected that the excitation current is corrected before the flux is converted, the only change that needs to be compensated is the excitation current disturbance.

FIG. 13 illustrates one possible embodiment of the I _ME resolver 202 of FIG. The resolver 202 has three phase components 220, 222 and 224, which generate phase excitation currents I _ME1 , I _ME2 and I _ME3 , respectively.

Each of the components 220, 222 and 224 is essentially the same except for the connection only in the entire system, so only the component 220 will be described in detail here.

Figure 13 uses the following equation for each phase.

Where · and X represent the vector sum, and the second equation is valid only in a symmetric three-phase system.

For phase component 220, a signal proportional to the current in phase 1, first phase, is provided from sensor 260 to a suitable vector multiplier 226. A voltage signal is provided to differential amplifier 228 via line 204 from phase 2 and phase 3 of motor 24. The output of amplifier 228 is equal to the phase-to-phase voltage V ₂₃ . The voltage signal is applied to the multiplier 226 as a second input, and the output of the multiplier becomes a vector of phase voltage and phase current in phase. The output is supplied to divider 230. The output of the amplifier 228 is applied to absolute value circuit 232, the output of the circuit is applied to the low pass filter 234. The output of the filter is a signal which is proportional to the absolute value of the signal V _23. The V ₂₃ signal is applied to the second input of the divider 230. The output of divider 230 is proportional to the motor excitation current with respect to phase 1 (I _ME1 ).

The I _ME1 signal is applied to summing junction 236 as one input. Components 222 and 224 are similarly connected to current in phases 2 and 3, and in a similar manner, provide outputs represented by I _ME2 and I _ME3 , respectively, which form additional inputs to summing junction 236. do. The output of summing junction 236 on line 210 is signal I _ME as shown in FIG.

14 shows an I _ME resolver. The resolver shows a significantly improved resolver compared to the relatively simple system of FIG. 13 in which the required circuit arrangement is further replaced. This embodiment uses information from the three phase phase to extract pulsation free I _ME values related to the pore voltage rather than the terminal voltage. The resolver of FIG. 14 uses the following motor excitation current magnitudes.

Regarding FIG. 14, a signal I ₁ representing the current in phase 1 is extracted from the sensor 206 and used as an input and supplied to the summation junction 254. The other input supplied to summing junction 254 is extracted from sensor 206 at system phase ₃ to provide signal I ₃ . Thus, the output at summing junction 254 is the current value at another phase I ₂ . In Fig. 14, the I ₁ signal is given as I _D , and I _D is converted into linear axis current. That is, the current is regarded as magnetic flux and in phase. The current signal is used as an input of multiplier 260. The I ₂ and I ₃ signals are summed at junction 256 and the output of the junction is fed to an amplifier 258 having a gain of 0.577. The output of the amplifier is a signal represented by quadrature current I _Q and the current I _Q is supplied to multiplier 262 as one input.

_Q는 각각 표시된 자속 신호를 제2입력 신호로 가진다. 상기 지속 신호

_D및

_Q는 다음과 같이 발생한다. 위상 전압 각각은 전압분할기에 의해 차동 증폭기에 인가되어 출력단에 접지에 대한 신호를 공급한다. 즉, 위상 전압 V₁은 위상 1 및 접지 사이에 접속되어 있는 레지스터(270) 및 (272)를 포함하는 전압 분할기에 의해 전압 분하기의 중간에서 한 입력으로서 차동 증폭기(282)에 공급된다. 유사한 방법으로, 전압 V₂는 한 입력으로서 레지스터(274) 및 (276)를 통해 증폭기(284)에 인가되며, 위상 전압 V₃는 레지스터(278) 및 (280)를 가지는 전압 분할기를 거쳐 차동 증폭기(286)에 인가된다. 증폭기(282),(284) 및 (286) 각각에 대한 제2입력은 접지에 접속되어 있다. 따라서 세개의 증폭기(282), (284) 및 (286)의 출력은 접지에 대한 위상 전압을 나타내는 신호를 각각 나타낸다. 상기 신호 각각은 한 입력으로서 합산 정션(290)에 공급되어 합해지며 상기 합은 이득 0.333을 가지는 증폭기(294)에 인가된다. 노드(292)에서의 증폭기(294)의 출력은 중성-접지 전압을 나타낸다. 노드(292)에서의 상기 신호는 접지 신호에 대한 각각의 전압과 결합되어 중성 신호에 대한 위상 전압을 발생 시킨다. 즉, 증폭기(282)의 출력은 합산 정션(268)에서 노드(292)에서의 전압과 결합되어 중성에 대한 위상 1의 전압(V_IN)을 나타낸다. 유사한 방법으로, 증폭기(284) 및 (286)의 출력은 합상 정션(296) 및 (298)에서 노드(292)에서의 신호와 결합되어 위상-중성 전압 신호 V_2N및 V_3N을 제공한다. 합산 정션(268),(296) 및 (298)의 출력은 3개의 합산 정션(300),(302) 및 (304)에 입력으로서 인가된다. 상기 합산 정션(300),(302) 및 (304)에 대한 제2입력은 전동기 전류에 의한 전동기 누설 인덕턴스 전압, 즉,

에 비례하는 신호이다.The multipliers 260 and 262 denote ø _D and the linear and quadrature magnetic flux components of the motor.

_Q has each of the displayed magnetic flux signals as a second input signal. The sustain signal

_D and

_Q occurs as follows. Each of the phase voltages is applied to the differential amplifier by a voltage divider to supply a signal to ground at the output. That is, phase voltage V ₁ is supplied to differential amplifier 282 as an input in the middle of voltage division by a voltage divider comprising resistors 270 and 272 connected between phase 1 and ground. In a similar manner, voltage V ₂ is applied to amplifier 284 via resistors 274 and 276 as one input, and phase voltage V ₃ is passed through a voltage divider having resistors 278 and 280 to a differential amplifier. Is applied to 286. The second input to each of the amplifiers 282, 284 and 286 is connected to ground. Thus, the outputs of the three amplifiers 282, 284, and 286 each represent a signal representing the phase voltage to ground. Each of the signals is fed to summation junction 290 as one input and summed and the sum applied to amplifier 294 having a gain of 0.333. The output of amplifier 294 at node 292 represents the neutral-ground voltage. The signal at node 292 is combined with the respective voltage for the ground signal to generate a phase voltage for the neutral signal. That is, the output of amplifier 282 is combined with the voltage at node 292 at summing junction 268 to represent the voltage of phase 1 (V _IN ) for neutral. In a similar manner, the outputs of amplifiers 284 and 286 are combined with signals at node 292 at summation junctions 296 and 298 to provide phase-neutral voltage signals V _2N and V _3N . The outputs of summing junctions 268, 296 and 298 are applied as inputs to the three summing junctions 300, 302 and 304. The second inputs to the summation junctions 300, 302 and 304 are the motor leakage inductance voltages, i.e.

This signal is proportional to.

신호를 제공하며, I₃신호에 접속되어 있는 유사한 미분회로(310)는 유사한 신호를 합산 정션(304)에 공급된다.Thus, the second input to junction 303 is the output of differential circuit 306 which receives the I ₁ signal as an input. In a similar manner, the differential circuit 308, which is connected to the I ₂ signal, can

A similar differential circuit 310, which provides a signal and is connected to the I ₃ signal, is supplied to the summation junction 304.

_D의 직선축 성분을 나타낸다. 상기

_D신호는 증배기(260)에 대하여 제2입력을 형성한다. V_R2및 V_R3신호는 합산 정션(308)에 인가되며, 상기 정션의 출력은 0.577의 이득을 가지는 증폭기(311)에서 곱해지며, 상기 출력은 출력단에서 전동기 전압 V_Q의 구적값을 나타내는 신호를 제공한다. 상기 신호는 추가되는 적분회로(312)에 인가되며, 상기 회로의 출력은 전동기 자속

_Q의 구적축값을 나타낸다. 상기

_Q신호는 증배기(262)에 대한 제2입력으로 형성된다. 증배기(260) 및 (262)의 출력은 합산 정션(314)에 인가되며, 상기 정션의 출력은 분할 함수(316)에 한 입력으로서 사용된다.The outputs of the three summation junctions 300, 302, and 304 represent the regenerative pore voltages of the motor and are denoted by V _R1 , V _R2 and V _R3 , respectively. The signal from the summation junction 300 (the direct current component of the motor voltage) is applied to the integrating circuit 306 as an input, the output of which is the motor magnetic flux.

The linear axis component of _D is shown. remind

_{The D} signal forms a second input to multiplier 260. The signals V _R2 and V _R3 are applied to the summation junction 308, the output of the junction being multiplied by an amplifier 311 with a gain of 0.577, which outputs a signal representing the quadrature value of the motor voltage V _Q at the output stage. to provide. The signal is applied to an integrating circuit 312, where the output of the circuit is a motor magnetic flux.

The quadrature axis value of _Q is shown. remind

_{The Q} signal is formed as a second input to multiplier 262. The outputs of the multipliers 260 and 262 are applied to the summation junction 314, which is used as an input to the partition function 316.

|)에 비례한다. 상기 신호는, 제14도에 있어서, 적분기(306)로부터 먼저

_D신호를 공급하므로 추출되며 또한 증배기(318)에 대해 입력으로 된다.The second input to the divider 316 is the absolute value of the motor flux (|

Proportional to |) The signal is first released from integrator 306 in FIG.

It is extracted by supplying the _D signal and is also input to the multiplier 318.

_D ²

_D

_Q ²

|)에 비례하는 신호이다. 분할기(316)의 출력은 I_ME전류이며, 상기 전류는, 제12도에 일치하여, 라인(210)을 통해 합산 정션(200)에 공급된다. 제14도의 실시예는 제13도에 비해 다음과 같은 장점을 가진다. 먼저, 제3위상 전류가 회로로부터 추출될 때 단지 두 위상 전류만 감지된다. 위상 전압은 상술한 방법으로 공극 전압으로 재생된다. 시스템은 3상 베이스로부터 전류 및 전압 신호 모두를 유사한 2상 직선 및 구적축 시스템으로 변환시킴으로 단순화된다. 또, 재생된 공극 전압은 적분되어 공극 자속이 되며 상기 적분은 전압의 고조파를 제거하여 더욱 명확한 벡터 곱셈이 되게 한다. 마지막으로, 자속의 크기는 절대치 함수보다는 제곱값의 합의 제곱근을 사용함으로 맥동이 없게 된다.Thus, the output of the multiplier is the square of the direct current flux component

_D ²

_D

_Q ²

Is proportional to |). The output of the divider 316 is an I _ME current, which is supplied to the summation junction 200 via line 210, consistent with FIG. The embodiment of FIG. 14 has the following advantages over FIG. First, only two phase currents are sensed when the third phase current is extracted from the circuit. The phase voltage is reproduced with the void voltage in the above-described manner. The system is simplified by converting both current and voltage signals from a three phase base into a similar two phase linear and quadrature system. In addition, the regenerated pore voltage is integrated to form a pore flux, and the integration eliminates the harmonics of the voltage, resulting in a clearer vector multiplication. Finally, the magnitude of the magnetic flux is free from pulsation by using the square root of the sum of squares rather than the absolute function.

The following describes motor driving and control when the motor flux level is inherently unstable such that the magnetic flux becomes saturated or zero in the absence of active control. Control of the magnetic flux is implemented by controlling the current output of the VAR generator, and if the current command by the system always exceeds the generator capacity in either direction, the magnetic flux is at a threshold and the drive is shut down. As an example, if the VAR generator is applied to the capacitor to compensate for the magnetic flux and the motor requires more excitation than can be obtained from the current sum of the capacitor and the VAR generator, the magnetic flux begins to decrease. This reduction will further increase the current from the load-switched inverter circuit in order to attenuate the possible capacitor current (as the motor voltage decreases) and maintain the torque to further reduce the possible excitation current.

Therefore, it makes the magnetic flux attenuate overall. If the VAR generator operates in inductor mode to reach a current limit capacity that cancels the capacitor current beyond the existing requirement for excitation currents, further solo conditions occur. In this state, if the VAR generator can no longer increase the current, even a small flux increase causes an increase in the capacitor current, increasing the excitation. More flux causes the load-switched inverter circuit current to reduce the excitation current increase (provides equal torque). The magnetic flux increases repeatedly until the motor saturation consumes all excitation currents. If the motor speed is close to the rating, the saturation level of the magnetic flux is likely to generate a motor voltage higher than the inverter can withstand, so it is necessary to stop the motor drive to avoid damage.

One way to solve this problem is to provide a VAR generator with sufficient capacity so that the current limit is not reached under any state, instantaneous state. But this is not an economic solution.

The method now being described is to transfer the motor excitation control to a load switched inverter circuit when the VAR generator is out of bounds. In this respect, it is assumed that the VAR generator used is of a type capable of providing true and terrestrial VARs.

Thus, if the VAR generator is at the limit for inductive mode operation, the LCI shifts the load side angle towards α = 90 ° and increases the current level in the same way, increasing the potato at the reactive current and avoiding torque changes. . The system is shown in Figures 15a, 15b and 15c. The figure shows a vector diagram of a motor current using laws and terms similar to those used in FIGS. 3, 4 and 5.

FIG. 15A shows the case where the VAR generator output current I _X can sufficiently cancel the capacitor current I _C exceeding the limit in the inductive direction. Here, full operation is performed within a stable range.

FIG. 15B shows the effect when there is a slight increase in speed but no change in torque. Capacitor current I _C increases with increasing speed and VAR generator output current I _X does not increase to offset increment I _C. Uncorrected errors cause the flux to saturate.

FIG. 15C shows an advanced embodiment, the apparatus of which is shown in FIG. In Fig. 15C, the LCI load side converter angle is changed by the value of Δα, and the LCI current I _L increases by ΔL _L. The net effect is that the motor torque is maintained (the vertical component of I _C remains unchanged) and the flux control is maintained (the sum of all horizontal components coincides with the horizontal component of I _M ).

Since the operation of the system is very satisfactory within the inductive limits, it is desirable to use it as a conventional motor system operating mode. That is, if the capacitor current slightly exceeds the amount of pure excitation current required, the VAR generator can be turned off or omitted and only driven on the load switching inverter.

If the VAR generator operates within capacitive current limits, it is shown in Figures 16a, 16b and 16 (c). In FIG. 16A, the possible capacitor current is small and the VAR generator increases in capacitive current until the current limit is reached. Once again, the system is stable because it satisfies the system's excitation current requirements.

Figure 16b shows a slight reduction in speed without change for a given motor torque. Here, the capacitor current I _C is reduced by the frequency attenuation. The VAR generator current I _X cannot increase, resulting in an incorrect error as indicated. This lack of excitation predicts the decay of the magnetic flux.

Figure 16c shows the effect of the improved embodiment. The LCI current is reduced until the sum of the available capacitor current and the VAR generator current meets the motor excitation requirements. No change in LCI phase angle is commanded. This is because the phase angle is already commanded to a value that generates the minimum reactive current in the potato direction. The torque provided is reduced by the illustrated ΔT. In this case, there is no way to satisfy the desired torque level, but in most cases the synthesis system will slow down or slow down the motor by reduced torque rather than shutting down to damp the magnetic flux so that no torque is produced at all. It is preferable. The useful limit of the VAR is reached regardless of the presence of the VAR generator at the operating point which prevents the transfer of the commanded torque. If the VAR generator can supply an advanced VAR, more torque can be generated before reaching the operating point. However, this indicates that a system operable without a VAR generator can be obtained, although at a reduced torque level.

15A-15C and 16A-16C clearly illustrate certain operating modes of embodiments of the present invention. An apparatus for obtaining the predetermined result is shown in FIG. FIG. 17 is slightly different but has the characteristics of the system shown in FIG. 12 (deformer of FIG. 11). In FIG. 17, the output of the I _ME regulator 212 is denoted by I _E ^* , in FIG. 12 the output is denoted by I _X ^* , and the output is a signal that directly controls the VAR generator. In FIG. 17, signal I _E ^* is applied to three new function blocks 400, 402 and 404, and excitation current correction proportional to the change in the excitation current required by motor 24. It is defined as a command.

The function block 400 provides a linear amplification function defined in both directions, the output of which is an I _X ^* signal, the signal being two adjustment functions 214 for controlling the VAR generator as described above. And 216. The limit at the block 400 output is designed to match the limit of the VAR generator. The I _E signal also forms an input to function block 402, which provides an output signal only at the upper limit of the output of block 400. Thus, if the I _E * command signal exceeds the capacity of the VAR generator in the capacitor direction, a spillover occurs through function block 402. The spillover passes through a gain block 406 that adjusts the limit of the programmable limiting circuit 408, reducing the load switching inverter circuit current to attenuate the demagnetizing effect required by the I _E ^* signal. This is the operation shown in FIG. 16C. The gain of block 406 is set to provide the same excitation current effect that would result from an increase in I _E ^* if the VAR generator could respond to the increase. This maintains the crossover frequency of the electric motor excitation current (I _ME ) loop that is changed by the transfer of the magnetic flux from the VAR generator control to the LCI circuit control of the magnetic flux.

The limiting circuit 408 has the characteristic of immediately responding to a small signal from the gain block 406. The minimum signal appears at the output of gain 406, and the clamp or limit of block 408 is instantaneously lowered to the torque command signal T ^* present. Thus, another output from block 406 reduces the signal I _L ^* from current reference block 140 ′ in a linear form. Regardless of the above characteristics, the small error passing through block 406 has no effect on the I _L ^* signal if the signal is not already near the maximum limit.

In this regard, as shown in FIG. 17, the current reference block 140 'and the phase angle reference block 146' are slightly different from the similar function blocks 140 and 146 shown in FIG. . In FIG. 11, the current reference block has only one lower limit level, while the current reference function 140 'in FIG. 17 has a variable lower limit level. Similarly, phase angle reference block 146 in FIG. 11 includes only one single slope with a threshold, while phase angle reference block 146 'in FIG. 17 has multiple slopes. It is necessary to effectively and properly operate in accordance with the inductive limits, which are described in US patents entitled "Terminal Voltage Limit Regulators for Load Current Inverters," filed by L. C. Tupper, dated May 1, 1984. Similar to the operation described in US Pat. No. 4,446,414. The above patent is specifically incorporated herein by reference.

Regarding the inductive threshold characteristic, if the I _E ^* command signal exceeds the capacity of the VAR generator in the inductive region, i.e., at the lower output limit of block 400, spill from block 404 (over command). There is an over output, which is transformed by the nonlinear gain block 410, the output of the gain block causing a change in the operation of the current reference function 140 'and the adjusted phase angle command reference function 146'. .

In order to properly understand that the output of block 410 affects blocks 140 'and 416', it is necessary to first describe the "normal" behavior of the two blocks 140 ', 146'. . In "normal" operation, there is no spillover signal from function block 404 and therefore no output from gain block 410. This causes the lower limit in current reference block 140 'to be at the lowest level and block 146' causes the associated phase angle control value to have the highest slope. At this time, the operation is the same as that described in FIGS. 11 and 12. That is, the current reference from block 140 'provides a current command to the load switched inverter circuit, which is linearly proportional to the size of I _L ^* having the lowest current required for operation of the LCI circuit. The phase angle command generated by block 146 'is almost full inversion (α = 180 °) or almost full rectification (α = 0 °) except when the torque command T ^* is near zero. As the torque command passes through zero, it is quickly converted from one phase angle command limit to another while the current command output from current reference block 140 'is at its minimum. The trajectory of the vector I _L as a function of the torque command signal T ^* is shown in FIG.

The reason for operating in this mode is to observe the lower current limit and provide the required torque from the LCI while causing the minimum potato current. The reactive component, potato direction LCI current component, has the magnitude shown by the following equation.

The torque generating component of the LCI current is represented by the following equation.

The above-mentioned torque generating current I _LT is a linear function of LCI. Reactive current I _LE is considered undesirable in normal operation. This is because cosα is maximum when sinα is near zero when α is near zero or 180 °. When α swings around the value of sinα (near 90 °), traversal occurs at a small current. That is, at the minimum value of the output of the current reference block 140 ', the potato current i _LE is minimum.

When the system reaches the inductive limit as shown in Figures 15A-15C and there is an output from the function block 404, the LCI, the load switching inverter circuit, senses without affecting the torque generating component I _LT . It is the function of the gain block 410 to derive the increased value of the current I _LE . Thus, LCI controls the flux by providing a potato (inductive) current without distributing the torque. This is accomplished by the output of gain block 410 applied to current reference function 140 ', effectively raising the minimum current commanded by the block. Blocks 140 'and 146' include the same functions as those shown in connection with block 408, such that the minimum signal from gain block 410 is immediately lower bound at block 140 '. To rise to a current level as indicated by the output of block 408. The lower limit increases linearly with another signal from block 410 causing the current reference block 140 'to raise the current. The phase angle reference block 146 'acts similarly to change the phase angle α = 90 °.

The dotted line connecting blocks 140 'and 146' illustrates the associated action.

As shown, the same signal from gain 410, which is an increase in the minimum current commanded by block 140 ', causes the phase angle commanded by block 146' to move toward α = 90 °.

The action of blocks 140 'and 146' causes the change in LCI current magnitude and angle to cause the invalid component I _LE of the LCI output to change without distributing the torque generating component I _LT of the current. The effect of the signal on the magnitude and angle of the LCI current from block 410 is shown by the vector diagram in FIG. 19.

Blocks 140 'and 146' are selected such that the gain and nonlinearity of gain block 410 provide a gain in amperes of magnetizing current I _LE for a given increase in I _E ^* command, the gain being I _E ^*. Equivalent to the gain of the VAR generator as amperage of current I _X for increasing command. The linearity should consider the following facts. In other words, I _LE having an invalid component with respect to a voltage below the switching reactance is not at the same angle as the I _ME current having an invalid component with respect to the motor pore voltage. However, the difference is not large. The combination of all the characteristics of blocks 404, 410, 410 'and 146' is to provide the motor excitation current for gain from the I _E ^* signal, which means that the control passes the current I through the VAR generator. _The same is true whether _X is provided or current I _L is provided via LCI. This ensures that the crossover frequency of the electric motor excitation current regulator does not change, regardless of which power circuit actually controls the excitation.

Accordingly, an embodiment of the present invention as shown in FIG. 17 relates to a method for maintaining control of an induction motor exciting current I _ME , which uses a VAR generator and a load switching inverter circuit (LCI), wherein the LCI is The power circuit provides a minimum change in the power of the I _ME regulated loop as a function used to control the motor excitation current.

With respect to FIGS. 15A-15C and 16A-16C, as described above, operation is possible without the VAR controller. Thus, FIG. 17 can be modified by removing the VAR controller and control circuitry (shown in dashed block 215), where the limit of the spillover signal (block 400) is related to the zero likelihood of the VAR controller, If set to zero, the system operates in the manner described with respect to FIGS. 15C and 16C.

20 shows another embodiment of the present invention. In the embodiment of FIG. 17, the output of the summation junction 200 is the difference between the I _ME ^* signal and the I _ME signal, the difference being the difference between the commanded motor excitation current and the actual motor excitation current. In FIG. 17, the I _ME signal is shown to be extracted from the I _ME resolver 202 as described in detail in FIGS. 12, 13 and 14 degrees. As illustrated in FIG. 11, it can be seen that the capacitor current | I _C | is derived from the actual sensing of the capacitor current.

However, the current may also be calculated at this time. 11 shows a method of generating an excitation current I _LE required for LCI. In another comparison, the short circuit junction 200 of FIG. 17 corresponds to the summation junction 166 of FIG. 11, and the summation junction 166 described below is applied to I _ME ^* , I _LE and | I _C | Has a signal.

The above-described I _ME resolver senses the motor current and the motor voltage and decomposes the current into an excitation component, which excitation component is a pore flux and in phase. In FIG. 11, the capacitor circuit senses the actual current and is useful for devices such as current transformers. The embodiment shown in FIG. 20 is capable of estimating the excitation component of the motor current from a system of the type basically shown in FIG. And decomposition can be avoided.

In particular, in FIG. 20, it can be seen that the type is the same as that in FIG. 17 except for the change of the input portion for the summation junction at the output terminal of the flux regulator 200 ′ and the I _ME regulator 212 of FIG. 17. . 17 As can be seen from the road, summing junction 200 receives the I _ME ^* signal from the flux regulator 164, and I I _{ME ME} receives signals from a resolver (202). In FIG. 20, the summation junction 200 'receives an I _ME ^* signal from the flux regulator 164, and receives a signal proportional to the absolute value of the capacitor circuit current | I _C |, as with the signal I _LEN . As can be seen from the foregoing, the signal I _LEN ignores any effect of the circuits 400, 402, 406, 408, and 410 with the spillover function. Effect.

From the foregoing, the motor excitation current I _ME is obtained by subtracting the reactive current I _LE formed by the load-side converter of the LCI from the sum of the excitation currents supplied by the capacitor circuit 32 and the VAR generator 40. same.

In other words,

From the above-described equation (6), from I _LE = I _L Simα and from the equation (1), the capacitor current I _C is represented by the following equation.

(If the passive network is added to the pure capacitor shown in 32 and connected in parallel to an electric motor including other circuit elements, all network admittance is used instead of WC).

By relocating and combining as described above, the current value that needs to be supplied by the VAR generator represented by I _X * is expressed as follows.

Equation (9) is correct when the spillover form indicated by the broken line in FIG. 20 is not an active component. In this case, the excitation current correction is performed by the operation of the VAR generator, and I _E ^* becomes equal to I _X ^* . However, if there is no spillover, the current and phase angle commands for the LCI may be changed by blocks 402-410 as described with respect to FIG. 17. The effect of the spillover characteristic causes the current excitation portion from the LCI to have two components.

The first component is " naturally " generated I _LEN and the second component is represented by I _LES and is induced by the action of the spillover function. therefore,

In order for the calculated I _ME adjustment to work properly, only the I _LEN component should be used in equation (6).

In FIG. 20, the illustrated I _LEN signal is extracted as follows. The T ^* signal from the speed regulator 139 is applied to the current reference function block 430 and the angle reference function block 432. Block 430 essentially has an absolute value circuit with a lower limit. Block 432, on the other hand, is a linear amplifier with a threshold, which is nearly identical to blocks 140 and 146 as shown in FIG.

The output of block 430 denoted by I _LN ^* is applied to multiplier 434 as one input. The output of the function block 432, denoted by αN *, is applied to the sine function 436, and the output of the sine function is the sine of the input. The output of block 436 forms a second input to multiplier 434. The output of the multiplier 434 is a signal proportional to the amount of excitation current and is supplied to the motor by an LCI without spillover, the signal I _LEN being used as an input and applied to the summation junction 200 '. Junction 200 ′ receives the I _ME ^* signal from flux generator 164 as described above. The derivation of the I _LEN signal is approximately the same as for signal I _LE from block 168 in FIG.

| 신호이며, 상기 신호는 상술된 정션(160)에 인가된다.The other input to summing junction 200 is | I _C | which is proportional to the capacitor current. It is a signal. The signal is derived according to equation (1). In this regard, the terminal voltage of the motor is applied to three absolute circuits 440, 442 and 444, the output of which is supplied to summing junction 446. The output of summing junction 446 is proportional to the absolute value | V _M | of the motor terminal voltage. The signal | V _M | is used as an input and applied to the multiplier 448. The magnetic flux magnitude signal is derived by the sum absolute circuit, indicated by the respective phase integrator and number 162, as shown in FIG. The output of the sum absolute circuit 162 is |

| Signal, which is applied to junction 160 described above.

The input to multiplier 448 is signal W, which is proportional to the motor stator frequency. The W signal can be derived in any suitable way, but in the illustrated embodiment, it is shown as the output of frequency converter 452, which is one of the motor terminal voltage signals that is the voltage on line 18 at its input. Has one of

The frequency converter, patented on June 12, 1984, filed by F.H.Boettner and L.H.Walker and entitled "Method and Apparatus for Measuring Frequency of Alternating Signals" US Patent No. 4,454,470 US Patent Application No. 638,003, filed by J. Double U. Gember and L. H. Walker, filed Aug. 6, 1984, with the title "Frequency Measurement System." Is specifically described by reference.

Output of multiplier 448, | V _M | _W is applied to the gain block 454, where the gain is the output of the gain circuit 454 according to equation (1) | I _C | The signal is proportional to the capacitance (admittance) of the circuit 132 to be a signal. In another aspect, as described in FIG. 17, in the operation of the circuit, an I _ME regulator 212 that supplies a command I _E ^* to the output stage is not necessary in this case.

This is because the predetermined value of I _E ^* in FIG. 17 is directly calculated rather than being calculated by adding up the error values in I _ME .

Although the characteristics of the circuit have been described in analog form, which is the easiest to understand, one of ordinary skill in the art appreciates that it is also possible to perform the actual control functions of the present invention by means of a suitably programmed computer device. will be.

This is described in US Pat. No. 4,449,087 to which reference is incorporated. In this patent, the control of the LCI is a computer, and similarly, many of the control functions of the various embodiments described can also be implemented by computer software. One form of such control is described in FIG. 21, which uses an Intel 80286 computer. In Fig. 21, the power circuit is the same as the circuit shown in Figs. 1, 17 and 20 degrees.

Ignition of the LCI source-side converter rectifier is by means of an appropriate ignition signal, which is transmitted from the ignition circuit 502 to the converter on line 500. Ignition circuit 502 is controlled by a signal on line 504 from a suitable data processor 506 (eg, an Intel 80256 computer system). The processor 506 receives the input from the operator controller 138 as well as the voltage and current feedback signals V _FB1 , I _FB1 from the input line to the converter 11.

In a similar manner, LCI load side converter 12 receives an ignition signal from ignition circuit 510 passing through line 508. The circuit is controlled from a second data processor 514 passing through line 512. The processor 514 again receives the voltage and current feedback signals V _FB2 and I _FR2 from the output of the converter 12, as is known. The processors 506 and 514 may be interconnected by the bus 515 to communicate with each other.

The variable VAR generator 40 has power from an AC / DC converter 42 connected to terminals L ₁ , L _2, and L ₃ by lines 44, 46, and 48. Supplied via 41). In this embodiment, the VAR generator 40 is an inverter type that switches to automatic continuous operation (see FIG. 8).

Similar to that described above for the LCI, the thyristors of the generator 40 are ignited by a signal transmitted from the ignition circuit 518 through the line 516, and the point circuit is connected to the line (by the data processor 522). Controlled through 520. Processor 524, line 526, ignition circuit 528, and line 530 have similar controls for converter 42. Bus 532 provides a communication link between processors 522 and 524, while bus 534 provides a communication link between processors 506, 514, and 524. Processor 522 receives a voltage feedback signal on line 540 and a current feedback signal on line 542 from lines 18, 20, and 22 that connect the LCI circuit to motor 24.

The digital device defined by the program, as shown in FIG. 21, works in much the same way as the device described above, particularly with respect to the analog embodiment described in FIG. Therefore, according to the present invention, the power can be supplied to the induction motor by a simple load switching inverter circuit, and has the advantage of providing a relatively inexpensive and robust induction motor associated with the LCI type driving.

Although specific embodiments of the present invention have been described and illustrated above, those skilled in the art can readily make modifications. Therefore, it is intended that the present invention not be limited to the particular embodiments described and shown above, but that the appended claims should cover all modifications that fall within the spirit and scope of the invention.

Claims (16)

A system for controlling the operation of an alternating-current temperature motor (24) having a winding provided with an electric excitation from a polyphase alternating current power source, the system being connected between a power source (L ₁ : L ₂ : L ₃ ) and the induction motor (24). And a controllable load switching inverter circuit 10 having an output for supplying power and an excitation current to the motor in a sensitizing manner, and connected to an output of the motor winding and the load switching inverter circuit 10. An induction motor operation having a capacitor circuit 32 for supplying a phase excitation current to the load switching inverter circuit 10 and a means for generating an electric motor command signal T ^* (136, 138, 139); in the control system, the number for generating the exciting current correction signal (I _E ^*) representing a change in the motor exciting current required for the motor operation at a predetermined level (200,200 ') and said exciting current correction signal (I _E ^*) in response to the means (402 404), and the spillover signal, and the electric motor to said exciting current correction signal to generate a spillover signal when it exceeds a predetermined threshold And control means for controlling the output of the load switching inverter circuit (10) in response to at least one of an operation command signal (T ^* ) and the excitation current correction signal. 2. A controllable VAR generator (40) according to claim 1, further comprising a controllable VAR generator (40) connected to an output of said motor winding and said load switching inverter circuit (10) for supplying an excitation current to said motor, wherein said predetermined limit. Induction motor operating control system, characterized in that it represents a rough excitation current capacity of the VAR generator (40). The method of claim 1, wherein the means for generating the excitation current correction signal I _E ^* is a means 161 for generating an excitation current command signal I _ME ^* indicating a motor excitation current of a predetermined level, Means 202 for generating a signal I _ME indicative of the actual excitation current supplied to the motor, and means 200 for combining the excitation current command signal with the signal indicative of the actual excitation current. Induction motor operation control system. 4. The method of claim 3, wherein said means 202 for generating a signal I _ME representing an actual exciting current supplied to the motor comprises means responsive to the motor terminal voltage V _M and the current I _M. Induction motor operation control system. The method of claim 3, wherein the predetermined motor flux (

*) And actual motor flux (|

Means) 152, 157, 162 for generating a signal indicative of the difference 160 therebetween, wherein said means 161 for generating an excitation current command is provided between a predetermined motor flux level and an actual flux level. Induction motor operation control system, characterized in that in response to a signal indicating a difference of. The output of the load switching inverter circuit 10 is an output current having a magnitude and a phase angle related to a reference point, wherein the control means is configured to control the output current in response to the spillover means. Induction motor operation control system comprising means for varying size (140 '). The output of the load switching inverter circuit 10 is an output current having a magnitude and a phase angle associated with a reference point, wherein the control means is a magnitude of the output current in response to the spillover signal. And means (140 ', 146) for varying phase angles. 3. The load switching inverter circuit (10) according to claim 2, wherein the load switching inverter circuit (10) is controllable AC / DC converter (11) on the supply side and interconnected DC / AC converter (12) on the load side, which are interconnected by a DC link (14). And the system further comprises the ac / dc converter at a different level when the spillover signal is present at a first level of operation in the absence of the spillover signal in response to the motor actuation command signal T ^* . Controlling the operation of the DC / AC converter 12 in response to the motor operating command signal T ^* and a first control passage comprising means 408,140 ', 142 for controlling the operation of (11). Second control passages 146 ′ and 148 for the third control passages for controlling the excitation current supplied to the motor by the VAR generator 40 in response to the excitation current correction signal I _E ^* ( 400,214,216,218) Induction motor operation control system. 9. The apparatus of claim 8, wherein said third control passage comprises means (400) for providing a limited output signal having an upper limit indicating an approximate limit of a forward exciting current supplied to said motor by said VAR generator. Induction motor operation control system. 10. The apparatus of claim 9, wherein the means for generating the spillover signal comprises means (402) for generating the spillover signal in response to an excitation current correction signal exceeding a value corresponding to a threshold of the limited output signal. Induction motor operation control system, characterized in that. A method for controlling the operation of an alternating current induction motor having a winding supplied with an electrical excitation from a polyphase alternating current power source, the method comprising: a sensitive to the motor as an output function of a controllable load switching inverter circuit connected between a power source and the induction motor. Supplying power and an excitation current to the motor; and supplying a phase excitation current to the motor and the load switching inverter circuit by using a capacitor circuit connected to an electric motor winding and an output terminal of the load switching inverter circuit. An induction motor operation control method comprising the step of generating an electric motor operation command signal, the method comprising: generating an excitation current correction signal indicative of a change in motor excitation current required for a motor operation of a predetermined level; When the limit exceeds the Generating a peelover signal, and controlling an output of the load switching inverter circuit in response to at least one of the spillover signal, the motor operation command signal, and the excitation current correction signal. How to control induction motor operation. 12. The induction motor operating control according to claim 11, further comprising the step of supplying excitation current to said load switching inverter circuit and said motor from a controllable VAR generator having an excitation current capacity indicated by said limit value. Way. The method of claim 12, wherein the generating of the excitation current correcting signal comprises: generating an excitation current command representing a motor excitation current of a predetermined level, generating a signal representing an actual excitation current supplied to the motor; And combining the excitation current command signal and the signal representing the actual excitation current. 15. The method of claim 13, wherein generating a signal indicative of the actual exciting current supplied to the motor is performed in response to the motor terminal voltage and current. 15. The method of claim 13, wherein generating the exciting current command comprises: generating a magnetic flux signal indicative of a motor flux of a predetermined level and a magnetic flux of an actual level, and inducing a signal proportional to the difference between the magnetic flux signals. And generating said exciting current command in response to said signal indicative of a difference between said magnetic flux signal. 12. The load switching inverter circuit of claim 11, wherein the load switching inverter circuit provides an output current having a magnitude and a phase angle with respect to a reference point and is controlled to change the magnitude and phase angle of the output current in response to the spillover signal. Induction motor operation control method characterized in that.

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