KR830000267B1 - Low cost variable speed constant torque induction motor drive circuit - Google Patents

Abstract

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Description

Low cost variable speed constant torque induction motor drive circuit

1 is a functional block diagram of a motor driving circuit according to an embodiment of the present invention.

2 is a diagram of a void coil associated with a motor stator pole.

3 is a general equivalent circuit diagram of an induction motor.

4 is a circuit diagram of a power stage of a motor drive circuit in a preferred embodiment.

5 is a relationship curve of an energization period of an inverter electrostatic switch with respect to a phase of an electric motor which is a part of an electric motor driving circuit.

6 is a block diagram showing a control circuit of the motor drive circuit in the preferred embodiment.

FIG. 7 illustrates the modulation technique applied to a set of regulator electrostatic switches in the motor drive of FIG.

8A and 8B are actual circuit diagrams for the base driving of two sets of transistors used in the preferred embodiment.

9 is a circuit diagram used in the preferred embodiment for the block diagrams of FIGS.

10 is a circuit diagram of a half-wave power stage that may be used to remove a set of electrostatic switches.

BACKGROUND OF THE INVENTION The present invention relates generally to the drive of alternating current electrostatic motors, and more particularly to an improved and less expensive design for the drive of motors of low ratings of this type.

Electrostatic bridge motors are usually powered from a series power source coupled with an inverter. The prior art has been a technique for regulating an electrostatic switch of an inverter to control the current supply, for example the torque of the motor and the frequency of the inverter, ie the speed of the motor. In particular, a special inverter circuit has been designed that makes the magnetic flux of the motor suitable in the starting method and at the same time prevents the inflow of the starting current which overloads the inverter.

In general, it is not economically easy to supply inrush current to an induction motor during start-up when excited with an inverter using a variable frequency, semiconductor. If the output current of the inverter is limited to about 1 per unit by properly controlling the output current of the inverter and the motor is operated at a slip larger than the rated escape slip, the torque generated by the motor is lowered and the motor at the desired speed. Insufficient to accelerate.

This condition can sometimes occur not only when starting but also when the motor is momentarily overloaded. To prevent this, the speed is usually measured with a tachometer, the slip of the motor is calculated, and the excitation frequency is adjusted to limit the slip below the escape slip. Because of the price of the tachometer, this solution is not suitable for low cost drive.

It is an object of the present invention to provide a low cost variable speed constant torque induction motor drive without the disadvantages of conventional technical arrangements.

In the present invention, an induction motor that is supplied with current by an electrostatic regulating inverter circuit operated by a direct current power source will be described. Current is distributed between a plurality of electrostatic control switches whose conduction is controlled by a current-locked loop relative to one pole of the DC power source coupled to the inverter. Means are provided for inducing a signal indicative of the magnetic flux of the motor and for controlling the switch in relation to the signal indicative of the induced magnetic flux to stabilize the desired current supply to the induction motor. In particular, a control loop is provided for placing a coil in the gear of the motor stator to measure the voltage to respond to the air gap magnetic flux and to adjust the inverter current to maintain a substantially constant air gap magnetic flux. The inverter controls the driving of the lamp by supplying only the rated current to the motor at any given time. The inverter maintains the supply current in relation to the sensed supply voltage, and the frequency of the inverter is automatically reduced when the magnetic flux of the lamp becomes too small.

Normally, in the driving of an electric motor using the present invention, the current error frequency of the inverter is adjusted by the action of the pore voltage of the coupled induction motor, the selection being made in relation to the level of the magnetic flux of the electric motor. Since the control circuit roughly adjusts the inverter frequency and current to maintain a constant void flux, an inverter that supplies only the rated current of the motor can start the motor at any speed point and provide constant coke over a wide speed range. No tachometer required As will be found in the following description, the inverter circuit stage of the control circuit and the motor drive according to the design of the present invention is very simple and inexpensive.

The primary feature of the present invention is a method in which the current of the motor is controlled. As determined by the flux loop, the current demand is set by the master distributor to each of the three phase inverters.

Each phase is alternately asynchronously adjusted to maintain that current within the tolerance limits of the required current. In this way, the inverter is inherently protected from overcurrent and the motor always maintains the desired voltage versus frequency (constant V / F) relationship. At low speeds, the motor current is distributed evenly between stages, allowing flexible operation without the need for complex and expensive pulse width modulation designs.

FIG. 1 shows an induction motor receiving an alternating current from a three-phase inverter that changes its power in a direct current power source (Epc). The three electrostatic switches Q ₁ , Q ₂ , Q ₃ , are combined with the positive pole and Q ₄ , Q ₅ , Q ₆ are combined with the opposite pole, and only the current of the motor is supplied normally, so that the motor can be started and Adjusted to maintain a constant torque.

The operation of the induction motor M related to the control of the inverter according to the present invention can be well understood by referring to the block diagram of FIG. Coil 1 (shown in FIG. 2 in relation to the gear of the stator) is mounted on the stator in such a way that it forms a loop over the width of one pole. As shown in Figure 2, for 36 slots of a four-pole mechanism, coil 1 spans nine slots.

는 선로 6에 발생한다. 선로 6의 자속 신호는 합성점 8에 의해 선로 7상에 실린 자속 기준신호

^*와 비교된다. 선로 9상의 출력오차

^*는 회로 10에서 증폭되어서, 선로 11상에서 전류 기준 신호 Ⅰ^*를 도출한다. 인버터에서, 전류센서(sensor), 즉 중앙탭이 있는 저항체 31은 직류 전원의 양극과 거기에 결합된 트랜지스터 Q₁, Q₂, Q₃의 어느 콜랙터 사이에 제공된다.The air gap voltage sensed by the coil 1 is applied to the integrating circuit 3 through the line 2, and the integrated signal representing the magnetic flux is rectified to the circuit 5 so that the signal in the single direction indicating the motor magnetic flux is

Occurs on line 6. The magnetic flux signal of line 6 is the magnetic flux reference signal loaded on line 7 by the synthesis point 8.

Compared to ^* Output Error on Line 9

^* Is amplified in circuit 10 to derive current reference signal I ^* on line 11. In the inverter, a current sensor, i.e. a resistor 31 with a center tap, is provided between the anode of the DC power supply and any collector of transistors Q ₁ , Q ₂ , Q ₃ coupled thereto.

The current indicating signal is derived from the resistor 31 via the line 30 and is applied to the comparator 32 as a second input, along with the signal I * of the line 11. The error signal ΔI on the line 33 is a controller, the inverter regulating circuit which drives the base of the transistors Q ₁ -Q ₆ by the line 25 in accordance with the sequence defined by the distributor 17. Is applied to. The conduction state distribution frequency between the various transistors is determined by a voltage-regulated oscillator 15 that regulates the distributor with a clock signal on line 16, which is determined by a setpoint (line 12) that represents the desired lamp speed. In other words, the conduction of the switch is adjusted to maintain the value automatically between R, comparator 32, and inverter control circuit 18, so that the rated value is maintained. The peculiarity of this control is that the distributor 17 handles the switch as two separate pairs Q1, Q2, Q3 and Q4, Q5, Q6, which are connected to each polarity of the current supply, while the distributor 17 maintains 120 ° conduction for each switch. Is in

Current control is performed here for one pair, i.e., Q4, Q5, Q6, while the feedback current is induced in the other pair of supply lines, i.e. anode and pair Q1, Q2, Q3.

As a result, the regulation effect on the current and the current response are distinct. This is an obvious feature of the motor drive using the present invention, and explains how this is done in the future. The second aspect of the invention is that in motor operation, it is protected from escape when starting and overloading. Referring to FIG. 1, the signal Ø of line 6 is compared by the synthesis point 35 along with the reference signal Ømin received on line 37.

The signal Ømin represents the minimum magnetic flux level that must be maintained in the air gap. At the output, an error is applied to the integrating circuit 38 and the direct current signal on line 39 passes through the critical waveform generator 40 which generates a signal on line 41 which is zero unless Ømin reaches line 6 in circuit 5. As soon as the threshold is reached, circuit 40 forms a signal proportional to the excess under Ømin. The signal at 41 is compared with the reference frequency signal at synthesis point 13 and the error on line 14 applied to voltage-controlled oscillator 15 causes a proportional frequency reduction at distributor 17 due to a change in the set point set at line 12 for speed. .

In the foregoing, it can be considered that the signal in coil 1 of the void is used for two purposes.

a. The magnetic flux signal Ø is compared to the reference signal Ø ^* . The resulting error signal is amplified and provides the inverter current requirement I ^* . When the magnetic flux demand is satisfied, only a minimum current is required, but when the magnetic flux is low, a proportionally higher current is required. This current causes the output of the inverter to be at a higher voltage, resulting in higher void flux, thus forming a closed loop system. The current demand I ^* is clamped to provide a slightly higher current than the rated current to protect the inverter.

b. If action a) cannot sustain the required flux, Ø falls below the minimum level Ømin. The magnetic flux Ø is compared to the minimum flux level Ømin as the standard. When Ø is below Ømin, the error signal is integrated and used to lower the frequency of the inverter by lowering the input to the voltage regulated oscillator. Since the inverter frequency is lowered, the air gap magnetic flux (integration of air gap voltage) is increased.

Actions a) and b) are diagrams illustrating the equivalent circuit of an induction motor with a current source input (characterized by resistors r ₁ , inductances X ₁ and X ₂ and XØ with load r ₂ / X). It can be understood by considering. At no load, the slip S is nearly zero and the resistance r ₂ / S is quite large. Nearly all motor terminal voltages appear across the air gap. However, due to the load on the machine, S increases and r ₂ / S decreases while reducing the pore voltage and the pore flux. The motor current and then the terminal voltage are then raised by the aforementioned control loop to maintain the required magnetic flux.

However, while starting the motor, the maximum current of the inverter prevents the motor from supplying much torque because the slip is very large and there is little air gap. Action b) responds to this condition and lowers the inverter frequency so that the slip is adequate and the magnetic flux is maintained.

During normal operation, the inverter frequency F is equal to the required frequency F ^* and the current I ^* is adjusted to suit the load. The motor speed matches F, except for a relatively small slip. During startup or when the rated torque of the drive is exceeded, the inverter frequency and motor speed are lowered to maintain the minimum required flux.

As a result, the motor can be started at any speed point and recover from mechanical overload after the overload is removed, while limiting the motor current to the nominal value.

The power stage is shown in FIG. This circuit consists of a full-wave, three-phase braid used in current regulation. For clarity, the induction motor is in the form of a Y, so each switch can be thought of as being connected between one pole of the power source and the winding and one end of one phase.

However, it is clear that the present invention can also be applied directly to the delta form. Each switch Q ₁ -Q ₆ by distribution 17 is conducted during 120 ° of the fundamental frequency, as shown on the curve di in FIG.

However, each switch below is regulated for an allowable 120 ° period in order to maintain the total current of the inverter which is sensitive by 31 at the required level I ^* .

The current loop operates in the optimum response or "bang-bang" mode so that the switch is turned on when I is less than I ^* -2 △ I and turns off when I is above I ^* + 2 △ I. As a result, all switches are momentarily protected from overcurrent.

Referring to FIG. 6, a preferred embodiment of the motor drive according to the present invention is shown in detail around the logic gate 22 following the base of transistor Q ₄ described over FIG. 1 how the current is regulated in the loop.

As shown for transistor Q ₄ on FIG. 6, the lower three transistors Q ₄ , Q ₅ , Q ₆ are adjusted to regulate the inverter current, whereas transistors Q ₁ , Q ₂ , Q ₃ are not, and resistor 31 Should be connected to the anode of DC power supply. In this way, the current sensitive resistor is positioned sensitively to the transistor current.

Distributor 17 controls Q ₁ , Q ₂ , and Q ₃ (usually a switch connected to the positive pole) by a line such as 20, and passes Q ₄ , Q ₅ , Q ₆ (via lines 21 and 24 through NAND gate 22). Usually a switch connected to the cathode).

Assuming that the NAND gates do not interfere with each other's 21 distributor signals, all switches conduct fully during 120 ° as shown by curves (d)-(i) in FIG. According to FIG. 5, if the winding of the motor is still Y-shaped, the switch (virtually a transistor) appears as a pair associated with each winding, with each pair of switches connected to two different poles of a DC power source.

The switches also belong to separate sets Q ₁ , Q ₂ , Q ₃ , and Q ₄ , Q ₅ , Q ₆ associated with each pole. Each switch conducts for 120 °, ie 360/3, and displaces two sets every 60 ° multiples (−60 ° by FIG. 3).

Distributor 17 distributes the gate signal at 60 ° alternately to one or the other in the order of the switches Q ₁ , Q _6, Q ₂ , Q ₄ , Q ₃ , Q _5, and so on. As a result, the currents of windings A, B and C can be measured by I _A , I _B and I _C by the curves (a), (b) and (c) shown in FIG. With this arrangement, it is clear that the current flows continuously from one pole to the associated pair of switches. As a result, it is possible, on the one hand, to sense the current flowing between the poles associated with a set of switches, and on the other hand to regulate the current flowing to the other set of switches for controlling as a function of this sensitive current.

For this purpose, resistor 31 is connected to each switch Q ₁ , Q ₂ , Q ₃ at the anode in a common branch line.

Three other NAND gates, such as 22, are inserted between each base of the transistors Q ₄ , Q ₅ and Q ₆ coupled with the distributor 17 and the cathode, along with the other pairs.

The error [Delta] I derived at synthesis point 323 is humanized in hysteresis, which defines a dead band between the upper and lower limits (± di) via line 33. When deadbands occur between the second input of NAND gate 22 through line 24 regulates conduction of transistor Q ₄ as shown in FIG.

The same regulated current occurs with each NAND gate with transistors Q ₅ and Q ₆ (not shown in FIG. 6). Therefore, the conduction period of 120 ° with respect to control switches Q ₄ , Q ₅ , Q ₆ in FIG. 5 is in contrast to the continuous conduction periods (curves d, e, f) of unregulated switches Q ₁ , Q ₂ , Q ₃ . It is shown to be discontinuous (curves g, h, i).

Bang-bang techniques for optimal response to control are generally known.

This is mentioned in US Patent No. 3,636,430 to A. Kernick et al.

Since rectifying diodes D ₁ -D ₆ are combined with the respective switches Q _1- Q ₆ , the energy accumulated in the windings is likely to circulate through the unregulated transistors Q ₁ , Q ₂ and Q ₃ that conduct continuously as a _pair . have.

This undesirable current cannot be regulated by turning off only the regulated transistors Q ₄ , Q ₅ , Q ₆ below, so that whenever I is greater than the maximum level I max (which is actually greater than I + d I), for example a switch The current control loops by Q ₄ , Q ₅ and Q ₆ are ready to turn off all switches that do not take motor current and reach an unacceptable maximum Imax.

8A shows a typical group of base drive circuits for applying control gates to positive switches Q ₁ , Q ₂ and Q ₃ . Thus, these three identical circuits are provided in inverter control circuit 18, which is coupled to line 25 by Q ₁ , Q ₂ and Q ₃ . A separate transformer is provided on lines 51 and 22 which carry the alternating current rectified by the full-wave rectifier bridge circuit 50. The capacitor 53 is charged and the induced DC voltage establishes a standard potential at the base of the first transistor TR ₁ , which is installed as the second transistor TR ₂ and the cathode follower, and at the output between B and E points. For example, a blocking level for the gate of Q ₁ is induced and applied to the switch. In order to obtain a separate coupling, the optic connector MCT 26 causes a short circuit to be formed between the reactor R and the common mode lead line G in parallel with the capacitor 53. As a result, the zener diode Z ₁ no longer blocks the base of the transistor Q ₁ , and as described with reference to FIG. 5, energization of the transistor Q ₁ conducted for 120 ° continues.

If the cited Figure 8b, the base drive circuit rogun for "control" the transistors Q _4, Q _5, Q ₆ is shown. In the isolation transformers previously cited in connection with FIG. 8A, an alternating voltage is induced and rectified in order to supply the necessary voltage to the transistor elements of this circuit group. Q ₄ , Q ₅ and Q ₆ are controlled by three pairs of transistors (TR ₃ , TR ₃ ), (TR ₄ , TR ₄ ) and (TR ₅ , TR ₅ ), respectively. This acts as a switch responsive to the gate signal as illustrated in FIG. 6 by NAND gate 22 for a particular switch Q ₄ .

Referring to FIG. 9, the group of circuits used in the preferred embodiment for the current control loop and the minimum flux velocity correction loop is shown.

Coil 1 is connected to the input of amplifier 1-OA installed as an integrating circuit, the output of which is rectified to diode 5. The output of this rectifier is connected to junction J. The signal through line 6 at junction J enters the negative input of operational amplifier 2-OA, a linear amplifier with a limiter at synthesis point 8, to effect the circuit 10 of FIG.

Composite point 8 also receives the reference value I by line 12 at port P. ΔØ is derived from the output of 2-OA at the junction of the series network, which is the positive feedback between the anode of Zener diode Z ₁₀ and resistor R ₁₀ . At this junction, or line 11, I ^* is applied to the comparison circuit 32, which consists of the junction and the op amp 3-OA. The feedback current is applied to the resistors through 30 of each other as the differential signals enter the operational amplifier 4-OA, and are also applied to the synthesis point S ₁ there. The output of 3-OA is applied to the inverter regulating circuit through the hysteresis circuit of FIG. 6 as previously described.

The circuit of FIG. 9 also contains a group of circuits that absorb reactive currents that used to circulate in unregulated Q ₁ , Q ₂ , and Q ₃ . For this effect, as shown in FIG. 9, the sensed feedback current signal in the operational amplifier 4-OA is supplied via the line 100 to the synthesis point S ₃ at the input of the operational amplifier 102. Composite point S ₃ also receives the output at 2-OA on line 101. As will be appreciated, Zener diode Z ₁₀ acts as a threshold element between signal I at 4-OA and threshold reference signal I ₁ at 2-OA. The output of amplifier 7-OA has two states depending on whether I is greater or less than I ₁ ^* .

I ₁ ^* represents the reference signal I ^* incremented by the neural diode Z ₁₀ by a predetermined amount. The output of 102 passes through inverse logic 103. Depending on the logic state of line 106, flip-flops 104-105 are set or reset. When set, all switches Q ₁ -Q ₆ are turned OFF by suitable means. After that, it is ON again after only another 120 ° period. The control circuit is updated.

Referring to FIG. 10, there is only one set of switches that conduct during the same and adjacent time intervals from one phase to another-in this particular example a very simple motor drive circuit with transistors Q ₄ , Q ₅ and Q ₆ Is indicated. In order to regulate the motor current, these switches are also regulated in relation to the current sensitive to the neutral conductor in the DC power supply. In this typical half-wave power stage, there is a device for distributing (and recovering, if necessary) the energy accumulated in the motor windings while a switch is energized. To achieve this effect, a short circuit loop consisting of the neutral point of the motor and the voltage sink VS connected to each of the diodes D ₄ , D ₅ and D ₆ is provided.

A description is given of the actuation of an alternating current motor in which a current is supplied to an induction motor through at least one set of electrostatic control switches that are equally distributed and energized between the phases during the same time interval, which is continuous at DC voltage and is connected from one switch to the next. With this arrangement, the total current in the DC power supply is sensed and combined with the reference to regulate the motor current by regulating the energization of the switch immediately during the time interval. In addition, means are provided for dispersing or restoring energy accumulated in the motor windings during operation.

Claims (1)

In a variable speed induction lamp in which an alternating current is supplied from a direct current power source through an inverter, the first set of electrostatic switches in the inverter coupled to the positive pole of the power source and the second in the inverter coupled to the negative pole of the power source. The pair of electrostatic switches and the pair of switches selected from the respective groups for the same conduction time interval and associated with each phase of the motor and

Means for distributing the time interval continuously through each phase of the motor, such as / N (N is the number of motor phases), and by the power source to adjust the motor current to the air flux and the specified current level of the motor. Means for regulating the energization of any one set of switches in relation to the current supplied to the inverter, and means for inducing a signal indicative of the magnetic flux of the motor and at a predetermined level to reduce the frequency of the distribution means. Means for responding to a substantial reduction in said signal magnitude for said motor.

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