KR20230153144A - Temperature-compensated current source and system for measuring current using the same - Google Patents

Abstract

A constant voltage source that supplies a predetermined voltage; a compensation voltage generator connected to the constant voltage output terminal of the constant voltage source and generating a compensation voltage that is a temperature compensated voltage; an amplifier whose first input terminal is connected to the compensation voltage output terminal of the compensation voltage generator; A transconductor connected to the output terminal of the amplifier; and a resistor connected to the transconductor and the second input terminal of the amplifier, and outputting a reference current that is resistant to temperature changes through the transconductor. A current measurement system using the same is disclosed.

Description

Temperature compensated current source and current measurement system using the same {TEMPERATURE-COMPENSATED CURRENT SOURCE AND SYSTEM FOR MEASURING CURRENT USING THE SAME}

The present invention relates to a temperature-compensated current source and a current measurement system using the same. More specifically, it relates to a temperature-compensated current source that outputs a reference current that is robust to temperature changes and a current measurement system using the same.

A representative current measurement method is a current measurement method using a shunt resistance. To briefly explain this, as the current to be measured passes through the shunt resistance, a voltage drop occurs in the shunt resistance. In this case, the shunt resistance This is a method of obtaining the voltage drop V by sensing the voltage at both ends and estimating the value of the current I to be measured by dividing the voltage drop V by the shunt resistance value R according to Ohm's law.

However, according to the conventional current measurement method using a shunt resistor, the following problems existed.

First, although the actual shunt resistance value changes greatly depending on temperature changes, according to the conventional current measurement method, a fixed shunt resistance value is assumed to have a fixed value regardless of temperature changes. was substituted for the law of , and therefore, there was a problem that the value of the measured current derived from this was inaccurate.

This will be described in more detail with reference to FIG. 1 as follows.

Figure 1 schematically shows a circuit for measuring the battery current I _BAT generated from a battery pack in a battery charging and discharging system.

Referring to FIG. 1, it can be seen that a shunt resistor R _S is inserted to measure the current I _BAT supplied from the battery pack to the load.

For reference, since the shunt resistance R _S has a very small value (e.g., a value of 5 mΩ or less), the voltage across the shunt resistor can be amplified using an amplifier as shown in Figure 1, and the amplified voltage D _OUT The current I _BAT can be estimated using .

At this time, the voltage D _OUT amplified by the amplifier can be calculated according to the formula below.

D _OUT (T) = I _BAT * R _S (T) * A(T) * (2 ^N / V _ref (T))

For reference, A(T) is the voltage gain of the amplifier, T is the temperature, N is the resolution of the ADC (Analog-to-digital converter), and V _ref is the reference voltage of the ADC. For convenience of explanation, only the shunt resistance value is shown below. It changes depending on the temperature, and the voltage gain and V _ref of the amplifier will be explained assuming that they have a fixed value regardless of the temperature change.

First, the shunt resistance value at room temperature is R _S , and given that the shunt resistance value at a specific temperature higher than room temperature is 1.1R _S , the actual output voltage D _OUT of the amplifier at a specific temperature is I _BAT * 1.1R _S * It is measured to have a value of A * (2 ^N / V _ref ).

And, the value of the measured current I _BAT ^ can be calculated according to the formula below.

I _BAT ^ = (D _OUT / (R _S (T) * A(T))) * (V _ref (T) / 2 ^N )

At this time, when 1.1R _S is substituted for R _S (T) in the I _BAT ^ formula above to reflect the actual shunt resistance value at a specific temperature, the measured current I _BAT ^ is expressed in the formula below

I _BAT ^= (D _OUT / (1.1R _S * A)) * (V _ref / 2 ^N ) = I _BAT

Since I _BAT is determined according to , it can be confirmed that the actual current I _BAT supplied from the battery pack to the load is calculated correctly.

On the other hand, according to the conventional measurement method, if the shunt resistance value R _S at room temperature is simply reflected in the above I _BAT ^ formula without considering the change in shunt resistance value due to temperature change, the measured current I _BAT ^ is as follows. formula

I _BAT ^= (D _OUT / (R _S * A)) * (V _ref / 2 ^N ) = 1.1I _BAT

Accordingly, it becomes 1.1I _BAT , so it can be confirmed that the actual current I _BAT supplied from the battery pack to the load was not calculated properly.

In order to solve the problem of the measured current being incorrectly calculated due to values that change according to temperature changes (e.g., shunt resistance value, voltage gain value, etc.), conventionally, a table of actual resistance values according to temperature change was used. That is, with a look-up table created, the current value was estimated using the actual resistance value according to the actual temperature.

However, in order to know the actual resistance value according to the actual temperature using a look-up table like this, there was a problem that a temperature sensor was needed to obtain the actual temperature value.

In addition, as explained earlier, because the value of the shunt resistance is very small, the resistance of the wire connected to the shunt resistance has a great influence on the measured voltage value, which causes the problem that the current value estimated from the measured voltage value is inaccurate. It existed.

Therefore, improvement measures to solve the above problems are required.

The purpose of the present invention is to solve all of the above-mentioned problems.

Another object of the present invention is to provide a temperature-compensated current source that outputs a current that is robust to temperature changes.

Another object of the present invention is to provide a current measurement system using a temperature compensated current source.

Another purpose of the present invention is to accurately measure current without being affected by temperature changes using a temperature compensated current source.

In order to achieve the object of the present invention as described above and realize the characteristic effects of the present invention described later, the characteristic configuration of the present invention is as follows.

According to one aspect of the present invention, a temperature compensation current source includes: a constant voltage source supplying a predetermined voltage; a compensation voltage generator connected to the constant voltage output terminal of the constant voltage source and generating a compensation voltage that is a temperature compensated voltage; an amplifier whose first input terminal is connected to the compensation voltage output terminal of the compensation voltage generator; A transconductor connected to the output terminal of the amplifier; and a resistor connected to the transconductor and a second input terminal of the amplifier, and outputting a reference current that is resistant to temperature changes through the transconductor.

As an example, a temperature compensated current source is disclosed, which is connected to the transconductor and further includes a first current mirror, which is an output current mirror that copies the reference current to generate a first radiation current.

As an example, a measurement in which a second radiation current is generated by copying the reference current while changing the temperature coefficient of the compensation voltage at the compensation voltage output terminal by a bit controller for varying the transistor included in the compensation voltage generator. As a result of monitoring the difference between the 2_1 radiated current generated at the first temperature and the 2_2 radiated current generated at the second temperature through a second current mirror, which is a current mirror, the second current mirror is connected to the transconductor. , in a state where the temperature coefficient of the compensation voltage when the 2_1 radiation current and the 2_2 radiation current are close to within a preset threshold range is determined as the optimal temperature coefficient and the corresponding setting value of the bit controller is maintained, A temperature compensated current source is disclosed, characterized in that the first radiation current is generated through a first current mirror.

As an example, the compensation voltage generator includes a first transistor group to an n-th transistor group, wherein the first transistor group to the n-th transistor group are connected in series, and n is an integer of 1 or more. Each of the transistor group and the nth transistor group is connected in parallel to (i) each of the first correction transistor to the nth correction transistor, (ii) each of the first correction transistor to the nth correction transistor, and each turns ON/OFF. Each of the 1_1st bit setting transistor to the 1_m bit setting transistor, the n_1th bit setting transistor to the n_m bit setting transistor connected through each switch for control, and (iii) the first correction transistor to the above. A temperature compensation current source is disclosed, which includes each of a first non-compensation transistor to an n-th non-compensation transistor connected in series to each of the n-th compensation transistors.

As an example, a first current mirror connected to the transconductor, which is an output current mirror that copies the reference current and generates a first radiation current, further includes a first current mirror for varying the transistor included in the compensation voltage generator. By changing the ON or OFF state of the switch connected to each of the 1_1 bit setting transistor to the 1_m bit setting transistor, the n_1 bit setting transistor to the n_m bit setting transistor, by the bit controller, A 2_1 radiated current and a second 2_1 radiated current generated at a first temperature through a 2nd current mirror, which is a measuring current mirror that copies the reference current and generates a 2nd radiated current - the 2nd current mirror is connected to the transconductor As a result of monitoring the difference between the 2_2 radiated current generated at temperature, the temperature coefficient of the compensation voltage when the 2_1 radiated current and the 2_2 radiated current approach within a preset threshold range is determined as the optimal temperature coefficient, and A temperature compensated current source is disclosed, characterized in that the first radiation current is generated through the first current mirror while maintaining the corresponding setting value of the bit controller.

As an example, a temperature compensated current source is disclosed wherein the resistor includes a switched capacitor and an oscillator.

As an example, the compensation voltage generator includes a first transistor group to an n-th transistor group - the first transistor group to the n-th transistor group are connected in series, and n is an integer of 1 or more. Each of the transistor group and the nth transistor group is connected in parallel to (i) each of the first correction transistor to the nth correction transistor, (ii) each of the first correction transistor to the nth correction transistor, and each turns ON/OFF. Each of the 1_1st bit setting transistor to the 1_m bit setting transistor, the n_1th bit setting transistor to the n_m bit setting transistor connected through each switch for control, and (iii) the first correction transistor to the above. It includes a first non-compensation transistor to an n-th non-compensation transistor connected in series to each of the n-th compensation transistors, and the temperature coefficient of the compensation voltage at the compensation voltage output terminal is α _COMP , and the temperature coefficient of the switched capacitor is α _C, when the temperature coefficient of the oscillator is _α A second current mirror, which is a measuring current mirror that copies the reference current and generates a second copy current while changing the ON or OFF state of the switch connected to each of the bit setting transistor or the n_m bit setting transistor - the above A second current mirror is connected to the transconductor. As a result of monitoring the difference between the 2_1 radiated current generated at the first temperature and the 2_2 radiated current generated at the second temperature, the 2_1 radiated current and the 2_2 When the radiation current approaches within a preset critical range and the temperature coefficient of the compensation voltage satisfies the formula below α _COMP + α _C + _α A temperature compensated current source characterized by maintaining a set value of a bit controller is disclosed.

As an example, when the value of the predetermined voltage supplied from the constant voltage source changes according to temperature change, and when the temperature coefficient of the constant voltage source is α _BGR , the 2_1 radiation current and the 2_2 radiation current are When the temperature coefficient _of the compensation voltage satisfies the equation below α _COMP + α _C + _α A temperature compensated current source is disclosed, characterized in that maintaining the set value of the bit controller.

As an example, a temperature compensated current source is disclosed wherein the constant voltage source includes one of a bandgap reference, a charged capacitor, and a Zener diode.

According to another aspect of the present invention, a constant voltage source that supplies a predetermined voltage; a compensation voltage generator connected to the constant voltage output terminal of the constant voltage source and generating a compensation voltage that is a temperature compensated voltage; and a compensation voltage output terminal of the compensation voltage generator. An amplifier connected to a first input terminal; A transconductor connected to the output terminal of the amplifier; and a resistor connected to the transconductor and the second input terminal of the amplifier; a current measurement system including a temperature compensated current source that outputs a reference current that is resistant to temperature changes through the transconductor, comprising: a power supply unit; A shunt resistor located between loads or between the power supply unit and a charging unit, the charging unit supplying power to the power supply unit; the temperature compensated current source supplying the reference current to the shunt resistor; And a voltage measurement unit that measures a measurement voltage value corresponding to the voltage that drops in the shunt resistor as the power current supplied from the power unit and the reference current supplied from the temperature compensation current source pass through the shunt resistor. , a current measurement system is disclosed, characterized in that the power current supplied from the power supply unit is identified by referring to the value of the reference current and the measured voltage value measured through the voltage measurement unit.

As an example, the measured voltage value measured through the voltage measurement unit is determined as the sum of the power supply voltage value corresponding to the power current supplied from the power supply unit and the reference voltage value corresponding to the reference current supplied from the temperature compensation current source. and the multiplexing current - the multiplexing current is generated using a specific multiplexing method among Time Division Multiplexing, Frequency Division Multiplexing, and Code Division Multiplexing, and the power current and the Is a current that multiplexes a reference current - identifies the power supply voltage value and the reference voltage value corresponding to the voltage that drops in the shunt resistor as it passes through the shunt resistor using the specific multiplexing method, and A current measurement system is disclosed, characterized in that the power current is identified by referring to the power supply voltage value, the power supply voltage value, and the reference voltage value.

The present invention has the effect of providing a temperature compensated current source that outputs a current that is robust to temperature changes.

Additionally, the present invention has the effect of providing a current measurement system using a temperature compensated current source.

Additionally, the present invention has the effect of accurately measuring current without being affected by temperature changes by using a temperature compensated current source.

The following drawings attached for use in explaining embodiments of the present invention are only some of the embodiments of the present invention, and to those skilled in the art (hereinafter “those skilled in the art”), the invention Other drawings can be obtained based on these drawings without further work being done.
Figure 1 schematically shows a conventional current measurement system,
Figure 2 schematically shows a current measurement system according to an embodiment of the present invention;
Figure 3 schematically shows a temperature compensation current source according to an embodiment of the present invention;
Figure 4 is a diagram for explaining the process of determining the optimal temperature coefficient of the compensation voltage at the compensation voltage output terminal of the compensation voltage generator of the temperature compensation current source according to an embodiment of the present invention;
Figure 5 schematically shows a compensation voltage generator of a temperature compensation current source according to an embodiment of the present invention;
Figure 6 schematically shows a current measurement system including a temperature compensated current source according to another embodiment of the present invention;
Figure 7a schematically shows the process of measuring current using a current measurement system including a temperature compensated current source according to another embodiment of the present invention;
7B shows the frequency domain of the power supply voltage value D _BAT (T) and the reference voltage value D _R (T) corresponding to the measured voltage value measured by a current measurement system including a temperature compensated current source according to another embodiment of the present invention. It schematically shows the spectrum in
Figures 8a to 8c schematically show a process of modulating the first radiation current I _R through the first current mirror in a current measurement system including a temperature compensated current source according to another embodiment of the present invention.

The detailed description of the present invention described below refers to the accompanying drawings, which show by way of example specific embodiments in which the present invention may be practiced to make clear the objectives, technical solutions and advantages of the present invention. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention.

Additionally, throughout the description and claims, the word “comprise” and variations thereof are not intended to exclude other technical features, attachments, components or steps. Other objects, advantages and features of the invention will appear to those skilled in the art, partly from this description and partly from practice of the invention. The examples and drawings below are provided by way of example and are not intended to limit the invention.

Moreover, the present invention encompasses all possible combinations of the embodiments shown herein. It should be understood that the various embodiments of the present invention are different from one another but are not necessarily mutually exclusive. For example, specific shapes, structures and characteristics described herein with respect to one embodiment may be implemented in other embodiments without departing from the spirit and scope of the invention. Additionally, it should be understood that the location or arrangement of individual components within each disclosed embodiment may be changed without departing from the spirit and scope of the invention. Accordingly, the detailed description that follows is not intended to be taken in a limiting sense, and the scope of the invention is limited only by the appended claims, together with all equivalents to what those claims assert, if properly described. Similar reference numbers in the drawings refer to identical or similar functions across various aspects.

Hereinafter, in order to enable those skilled in the art to easily practice the present invention, preferred embodiments of the present invention will be described in detail with reference to the attached drawings.

First, in order to overcome the problem of the measured current having an inaccurate value due to the resistance value and wire resistance that changes depending on the temperature, which was explained previously, the current measurement method proposed by the present invention is discussed with reference to FIG. 2. I will explain briefly.

Referring to FIG. 2, it can be seen that a shunt resistor R (200) is inserted into the circuit to measure the current I in the circuit, but unlike the conventional measurement method, a current source (100) is added.

For reference, in FIG. 2, for convenience, the current source 100 is shown as being added to both the top and bottom of the shunt resistor 200. However, this means that the current source 100 is attached to either the top or bottom of the shunt resistor 200. Since it is only intended to indicate that the same result can be obtained even if added, the current source 100 does not necessarily have to be added to the top and bottom of the shunt resistor 200 in the current measurement system according to an embodiment of the present invention. .

Referring to FIG. 2, the current I^ to be measured can be calculated according to Ohm's law using the formula below.

I^ = V _I / (R(T) + 2r(T))

For reference, V _I represents the voltage that drops across the resistance as the current I in the circuit passes through the resistance R(T)+2r(T), R(T) is the shunt resistance (200), and r(T) is the shunt resistance. The resistance of the wire connected to the resistor, T, represents the temperature.

At this time, the value of the actual resistance R(T) + 2r(T) can be calculated as the formula below according to Ohm's law.

[R(T) + 2r(T)] = [Voltage drop across R(T) + 2r(T)] / [Current through R(T) + 2r(T)]

Therefore, when the current source 100 supplying the current I _R is added according to the method proposed by the present invention, R(T) + 2r(T) can be calculated as the formula below.

[R(T) + 2r(T)] = V _R / I _R

For reference, V _R represents the voltage that drops across the actual resistance as the current I _R from the current source 100 passes through the actual resistance R(T) + 2r(T).

Accordingly, the current I^ to be measured, described earlier, can be modified as shown in the formula below.

I^ = V _I / [R(T) + 2r(T)]

= V _I / (V _R / I _R )

= (V _I / V _R ) * I _R

At this time, V _I is the voltage that drops across the resistance as the current I in the circuit passes through the actual resistance R(T)+2r(T), so it can be expressed as V _I = [R(T) + 2r(T)] * I. V _R is the voltage that drops in the actual resistance as the current I _R from the current source 100 passes through the actual resistance R(T) + 2r(T), so V _R = [R(T) + 2r( T)] * I _R It can be expressed as

Therefore, the formula for the current I^ to be measured can be organized as follows.

I^ = (V _I / V _R ) * I _R

= ((R(T) + 2r(T)) * I / ((R(T) + 2r(T)) * I _R )) * I _R

= I

Referring to the above formula, even if the resistance value changes depending on the temperature and/or the reference current from the reference current source is added, regardless of this, the value of the measured current I^ is accurately assumed to have the value of the actual current I in the circuit. You can check the calculation.

However, as above, (V _I / V _R ) * I _R is converted into ((R(T) + 2r(T)) * I / ((R(T) + 2r(T)) * I _R )) * I In order to organize it into _R , the prerequisite that the current I _R generated from the current source must have a constant value regardless of temperature changes must be satisfied.

To explain why the above prerequisites must be satisfied, referring back to the formula for the measured current I^ is as follows.

I^ = (V _I / V _R ) * I _R

If the value of the current _IR generated from the current source changes depending on the temperature change, the value of the current that changes depending on the temperature cannot be known, so in the above formula, the fixed current value indicated on the current source is simply substituted for _IR .

On the other hand, V _R is a value actually measured across the shunt resistor as the current from the current source passes through the shunt resistor at a specific temperature, so it corresponds to a voltage value reflecting the current value that varies depending on temperature changes.

Therefore, if the current source outputs a current of IR at room temperature and a current of 1.1I _R at a specific temperature higher than room temperature, _VR measured at room _temperature is (R(T) + 2r(T)) * I _R has a value, and _VR measured at a specific temperature has a value of (R(T) + 2r(T)) * 1.1I _R.

Therefore, (i) substituting V _R measured at room temperature into the equation for the measured current I^, I^ = (V _I / V _R ) * I _R = (((R(T) + 2r(T)) * I) / ((R(T) + 2r(T)) * I _R )) * I _R = I, but (ii) if V _R measured at a specific temperature is substituted into the equation for the measured current I^, , I^ = (V _I / V _R ) * I _R = (((R(T) + 2r(T)) * I) / ((R(T) + 2r(T)) * 1.1I _R )) * Since I _R = I/1.1, it can be confirmed that errors occur in the measured current I^ due to temperature changes.

In the above, the reason why a current source that outputs a reference current that is robust to temperature changes is necessary in order to accurately measure the value of current is explained. Therefore, below, the temperature that outputs a reference current that is robust to temperature changes according to an embodiment of the present invention will be described. The compensation current source will first be described with reference to FIGS. 3 to 5, and then the current measurement system including the temperature compensation current source according to another embodiment of the present invention will be described with reference to FIGS. 6 to 8C.

Figure 3 schematically shows a temperature compensation current source 1000 that outputs a reference current that is robust to temperature changes according to an embodiment of the present invention.

Referring to FIG. 3, the temperature compensation current source 1000 according to an embodiment of the present invention is connected to (i) a constant voltage source 1100 that supplies a predetermined voltage, and (ii) a constant voltage output terminal of the constant voltage source 1100. , a compensation voltage generator 1200 that generates a compensation voltage that is a temperature-compensated voltage, (iii) an amplifier 1300 whose first input terminal is connected to the compensation voltage output terminal of the compensation voltage generator 1200, (iv) an amplifier 1300 ) may include a transconductor 1400 connected to the output terminal of (v) and a resistor 1500 connected to the second input terminal of the transconductor 1400 and the amplifier 1300.

In addition, the temperature compensation current source 1000 according to an embodiment of the present invention is connected to the transconductor 1400 and includes a first current mirror 1600, which is an output current mirror that copies the reference current to generate the first radiation current. may additionally be included.

For reference, (i) the output current mirror is a current mirror for generating a radiation current by copying the output current from the transconductor 1400 and supplying the radiation current to the inside of the current measurement target circuit, which will be explained later, (ii) The current mirror for measurement, which will be described in more detail later, is used from the transconductor 1400 according to various settings of the compensation voltage generator 1200 in order to determine the optimal compensation voltage temperature coefficient of the compensation voltage generator. It may refer to a current mirror for generating radiation current by copying various output currents.

For reference, the transconductor 1400 of the temperature compensation current source 1000 according to an embodiment of the present invention is an element or circuit device that outputs a predetermined current in response to a predetermined voltage being input. Below, The description will be made using a transistor (eg, NMOS transistor) as an example, but this is only for convenience of explanation and the present invention is not limited thereto.

In addition, the constant voltage source 1100 of the temperature compensation current source 1000 according to an embodiment of the present invention includes one of a bandgap reference, a charged capacitor, and a Zener diode. can do.

Additionally, the resistor 1500 of the temperature compensation current source according to an embodiment of the present invention may include at least a portion of a fixed resistor with a fixed value and a variable resistor with a variable value.

For example, the resistor 1500 of the temperature compensation current source 1000 according to an embodiment of the present invention may include a switched capacitor and an oscillator, as shown in FIG. 3. At this time, the oscillator may be a crystal oscillator, but the present invention is not limited thereto. For reference, even without additional explanation of the switched capacitor, a person skilled in the art will be able to clearly understand the operation of the temperature compensation current source 1000 according to an embodiment of the present invention.

As another example, the resistor 1500 of the temperature compensation current source 1000 according to an embodiment of the present invention may include a fixed resistor having a fixed resistance value. For reference, a fixed resistor with a fixed resistance value usually has a larger temperature coefficient than a variable resistor including a switched capacitor and an oscillator.

As above, the temperature compensation current source 1000 according to an embodiment of the present invention may include various elements and/or circuit devices. However, these elements and/or circuit devices, such as resistors, have a unique temperature coefficient, and the value of resistance may change depending on temperature changes.

Therefore, the temperature compensation current source 1000 according to an embodiment of the present invention is configured to output a reference current with a constant value even if the value of elements and/or circuit devices, such as resistance, changes according to temperature changes. It may include a compensation voltage generator 1200.

Below, in order to output a constant value of current regardless of temperature change, the optimal compensation voltage, which is the voltage at the compensation voltage output terminal of the compensation voltage generator 1200 of the temperature compensation current source 1000 according to an embodiment of the present invention, is described. The process by which the temperature coefficient is determined will be described with reference to FIG. 4.

Referring to FIG. 4, the temperature coefficient of the compensation voltage V _COMP at the compensation voltage output terminal is changed by the bit controller 400 for varying the transistor included in the compensation voltage generator 1200, and the reference current is copied. The difference between the 2_1 radiation current generated at the first temperature and the 2_2 radiation current generated at the second temperature can be monitored through the second current mirror 300, which is a current mirror for measurement that generates 2 radiation currents. For reference, the second current mirror 300 may be connected to the transconductor 1400.

At this time, the first temperature and the second temperature may be determined according to the environment in which the current measurement system is used, and the number of bits of the bit controller 400 may be determined according to design needs.

Then, the monitoring result is referred to, and the temperature coefficient of the compensation voltage when the 2_1st radiation current and the 2_2nd radiation current are close to within the preset threshold range is determined as the optimal temperature coefficient, and the setting value of the bit controller 400 corresponding thereto is determined. Accordingly, by ensuring that the temperature coefficient of the compensation voltage at the compensation voltage output terminal has the optimal temperature coefficient, the temperature compensation current source 1000 according to an embodiment of the present invention can output a reference current that is robust to temperature changes.

For example, assuming that the first temperature is -40 degrees Celsius, the second temperature is 125 degrees Celsius, the number of bits of the bit controller 400 is determined to be 7, and the preset threshold range is set to 0, the bit controller 400 ), the results of monitoring the difference between the 2_1st radiation current and the 2_2nd radiation current in each case can be expressed as in the table below. For reference, since the number of bits is 7, the control value set by the bit controller 400 may have a value from 0 to 127.

control value 2_1 radiated current at the first temperature (-40 degrees Celsius) 2_2 radiation current at the second temperature (125 degrees Celsius) Difference between 2_1 radiation current and 2_2 radiation current 0 I _{-40,0,mirrored} I _{125,0,mirrored} △ ₀ One I _{-40,1,mirrored} I _{125,1,mirrored} △ ₁ ... ... ... ... 80 I _{-40,80,mirrored} I _{125,80,mirrored} △ ₈₀ ... ... ... ... 127 I _{-40,127, mirrored} I _{125,127, mirrored} △ ₁₂₇

In the table above, when the control value set by the bit controller 400 is 80 and the difference between the 2_1 radiation current and the 2_2 radiation current is monitored as 0, the 2_1 radiation current and the 2_2 radiation current are preset. Since this corresponds to the case where it approaches within the critical range, the temperature coefficient of the compensation voltage when the control value set by the bit controller 400 is 80 can be determined to be the optimal temperature coefficient.

Accordingly, the control value set by the bit controller 400 is set to 80 so that the temperature coefficient of the compensation voltage at the compensation voltage output terminal has an optimal temperature coefficient, so that the temperature compensation current source (1000) according to an embodiment of the present invention ) can output a reference current that is robust to temperature changes.

Below, the compensation voltage generator 1200 of the temperature compensation current source 1000 according to an embodiment of the present invention will be described in more detail.

The compensation voltage generator 1200 of the temperature compensation current source 1000 according to an embodiment of the present invention may include a first to nth transistor group.

At this time, the first transistor group to the n-th transistor group may be connected in series (if n is 2 or more), and each of the first transistor group to the n-th transistor group is (i) a first correction transistor to an n-th correction transistor. Each of the transistors, (ii) a 1_1 bit setting transistor to a 1_m bit setting transistor connected in parallel to each of the first correction transistor to the nth correction transistor, but connected through respective switches for controlling the ON/OFF of each. , to the n_1th bit setting transistor to the n_mth bit setting transistor, respectively, and (iii) each of the first non-correction transistor to the nth non-correction transistor connected in series to each of the first correction transistor to the nth correction transistor. You can. For reference, n is an integer greater than 1.

In the above, the correction transistor is a transistor connected in parallel to the bit setting transistor that controls the ON/OFF state of the switch to change the control value, and the non-compensation transistor is a transistor in which the bit setting transistor is not connected in parallel. , the names 'compensation' and 'non-compensation' are used, respectively, and the same transistor can be used for the compensation transistor, non-compensation transistor, and bit setting transistor included in each transistor group.

For example, a PMOS transistor with the same specifications can be used for a correction transistor, a non-compensation transistor, and a bit setting transistor.

However, the present invention is not limited to this, and a plurality of different types of transistors may be used as the correction transistor, non-compensation transistor, and bit setting transistor included in each transistor group.

And, similar to what was previously described with reference to FIG. 4, the 1_1 bit setting transistor to the 1_m bit setting transistor, through the bit controller 400 for varying the transistor included in the compensation voltage generator 1200. By changing the ON or OFF state of the switch connected to each of the n_1 bit setting transistor to the n_m bit setting transistor, the reference current is copied and the second copy current is generated through a second current mirror, which is a measurement current mirror. As a result of monitoring the difference between the 2_1 radiation current generated at temperature 1 and the 2_2 radiation current generated at the second temperature, a thermometer of the compensation voltage when the 2_1 radiation current and the 2_2 radiation current are close to within a preset threshold range The number can be determined as the optimal temperature coefficient and the corresponding set value of the bit controller (i.e., the specific ON/OFF state of the switch connected to each bit setting transistor inside the compensation voltage generator) can be maintained.

Figure 5 shows the compensation voltage generator 1200 of the temperature compensation current source according to an embodiment of the present invention in more detail.

Referring to FIG. 5, the compensation voltage generator 1200 of the temperature compensation current source according to an embodiment of the present invention includes a first transistor group 1210 and a second transistor group 1220, and the first transistor group ( It can be seen that 1210) and the second transistor group 1220 are connected in series.

At this time, the first transistor group 1210 is connected in parallel to (i) the first correction transistor 1211 and (ii) the first correction transistor through respective switches to control each ON/OFF. It can be confirmed that it includes the 1_1 bit setting transistor 1212_1 to the 1_4 bit setting transistor 1212_4, and (iii) the first non-compensation transistor 1213 connected in series to the first correction transistor 1211.

For reference, the description of the second transistor group 1220 is similar to the description of the first transistor group 1210, so overlapping descriptions will be omitted.

And, similar to what was explained with reference to FIG. 4, the 1_1 bit setting transistor to the 1_4 bit setting transistor, through the bit controller 400 for varying the transistor included in the compensation voltage generator 1200. While changing the ON or OFF state of the switch connected to each of the 2_1 bit setting transistor to the 2_4 bit setting transistor, the 2_1 radiation current generated at the first temperature and the 2_1 radiation current generated at the second temperature through the second current mirror 2_2 The difference in radiation current can be monitored.

Referring again to FIG. 5, since each transistor group 1210 and 1220 includes four bit setting transistors (i.e., the number of bits is 4), the range of control values that can be set by the bit controller 400 is In a situation of 0 to 15, if the control value when the difference between the 2_1st radiation current and the 2_2nd radiation current approaches within the critical range is 5, the temperature coefficient of the compensation voltage when the control value is 5 is the optimal temperature coefficient. By determining, (i) the switch connected to the 1_1st bit setting transistor (1212_1), the 1_3th bit setting transistor (1212_3), the 2_1st bit setting transistor (1222_1), and the 2_3 bit setting transistor (1222_3) It is maintained in a closed state, and (ii) the 1_2 bit setting transistor (1212_2), the 1_4 bit setting transistor (1212_4), the 2_2 bit setting transistor (1222_2), and the 2_4 bit setting transistor (1222_4). Allows connected switches to remain open.

For reference, in FIG. 5, (i) the compensation voltage generator 1200 includes two transistor groups 1210 and 1220, and (ii) the first non-compensation transistor 1213 is located at the bottom of the first compensation transistor 1211. is connected in series, the second correction transistor 1221 is connected in series to the bottom of the first non-compensation transistor 1213, and the second non-compensation transistor 1223 is connected in series to the bottom of the second correction transistor 1221. and (iii) the number of bit setting transistors corresponding to each of the first correction transistor 1211 and the second correction transistor 1221 is shown as 4, but this is only an example to aid understanding, and the present invention This is not limited to this.

For example, (i) the compensation voltage generator may include a single transistor group or a group of three or more transistors, (ii) the connection order of the compensation transistor and the non-compensation transistor may be different (e.g., at the bottom of the first non-compensation transistor) A first correction transistor may be connected in series, a second correction transistor may be connected in series to the bottom of the first correction transistor, and a second non-compensation transistor may be connected in series to the bottom of the second correction transistor), (iii ) The number of bit setting transistors corresponding to each of the first correction transistor and the second correction transistor may be a value other than four.

In addition, for convenience of explanation, the ON/OFF state of the switch connected to the bit setting transistor of the first transistor group and the ON/OFF state of the switch connected to the bit setting transistor of the second transistor group were described as being the same. The present invention is not limited to this.

For example, the ON/OFF state of the switch connected to the bit setting transistor of the first transistor group may be different from the ON/OFF state of the switch connected to the bit setting transistor of the second transistor group.

For example, to correspond to the optimal temperature coefficient of the compensation voltage, (i) the switch connected to the 1_1 bit setting transistor 1212_1 and the 1_4 bit setting transistor 1212_3 of the first transistor group is maintained in a closed state, (ii) the switch connected to the 1_2 bit setting transistor 1212_2 and the 1_3 bit setting transistor 1212_4 of the first transistor group remains open, and (iii) the 2_1 bit setting transistor of the second transistor group The switch connected to the transistor 1222_1 and the 2_2 bit setting transistor 1222_3 is kept open, and (iv) the 2_3 bit setting transistor 1222_2 and the 2_4 bit setting transistor ( The switch connected to 1222_4) can be kept closed. Of course, unlike this example, it can be assumed that the ON/OFF state of the bit setting transistor of the first transistor group and the ON/OFF state of the bit setting transistor of the second transistor group do not have the same or opposite relationship. You will be able to.

Meanwhile, below, the process by which the optimal temperature coefficient of the compensation voltage is determined will be explained using a specific formula.

Referring again to FIG. 3, the equation for the voltage V _S at the second input terminal of the amplifier 1300 is as follows.

V _S = V _S0 * [1 + (α _BGR + α _COMP (N)) * △T]

For reference, V _S0 is the voltage at the second input terminal at room temperature (e.g., 27 degrees Celsius), α _BGR is the temperature coefficient of the constant voltage source, and α _COMP (N) is the temperature coefficient of the compensation voltage of the compensation voltage generator. , △T is the result of subtracting room temperature from the current temperature.

And, when the resistance includes a switched capacitor and a crystal oscillator, the formula for the resistance R _avg,SWCAP is as follows.

R _{avg ,SWCAP = 1} / _[ f

_For reference, f _XO is _the clock frequency of the crystal oscillator, _α

And, due to the characteristics of a transconductor (eg, NMOS transistor), the formula for the current I ₀ output by the temperature compensation current source is as follows.

I ₀ = V _S / R _avg,SWCAP

₌ V _{S0 *} f _{XO *} C ₀ * (1 ₊ α

_{And ,} ( ₁ + _α The formula for I ₀ can be approximated as follows.

V_S0 * f_XO * C₀ * (1 + α_XO + α_C + α_BGR + α_COMP(N)) * △TI ₀

_{V S0 *} f _{XO *} C _{0 *} (1 + α

_{Here , if α _ _ _ _} It is possible to output a reference current that is independent of .

At this time, α _BGR and α _COMP (N) will be described in more detail with reference to the compensation voltage generator 1200 shown in FIG. 5. (i) first non-compensation use of the first transistor group 1210 When the transistor 1213 is called an M1 transistor, and (ii) the first correction transistor 1211 and the 1_1 bit setting transistors 1212_1 to 1_4 bit setting transistors 1212_4 are called M2 transistors, the M1 transistor and The drain currents I _M1 and I _M2 flowing through each of the M2 transistors can be expressed as follows.

I _M1 = μ * C _OX * (W/L) * V _T ² * exp((V _COMP - V _mid - V _TH1 )/mV _T )

I _M2 = μ * C _OX * N * (W/L) * V _T ² * exp((V _BGR - V _COMP - V _TH2 )/mV _T )

For reference, μ is the charge mobility, C _OX is the gate-oxide capacitance, W is the channel width of the transistor (e.g. MOSFET), L is the channel length of the transistor (e.g. MOSFET), and N is the ratio of the channel widths of M1 and M2. , V _T is the thermal voltage (kT/q), m is the subthreshold slope factor, V _TH1 and V _TH2 are the threshold voltages of M1 and M2, and V _BGR is the constant voltage source (e.g., bandgap reference). is the output voltage.

At this time, since M1 and M2 are connected in series, I _M1 = I _M2 is established, and the compensation voltage V _COMP can be expressed as follows.

V _COMP = (1/2) * [(3/2) * V _BGR + mV _T * lnN + △V _TH ]

For reference, △V _TH is the value obtained by subtracting V _TH2 from V _TH1 .

At this time, the effect of temperature change on V _BGR , V _T , and △V _TH is expressed in the equation as follows.

V _BGR = V _BGR0 * (1 + α _BGR * △T)

V _T = (k/q) * (T ₀ + △T)

△V _TH = △V _TH0 * (1 + α _△VTH * △T)

For reference, V _BGR0 is the output voltage of the voltage source (e.g., bandgap reference) at room temperature, △V _TH0 is the difference between the threshold voltages of M1 and M2 at room temperature, and α _BGR is the thermometer of the voltage source (e.g., bandgap reference). Number, α _△VTH is the temperature coefficient of the threshold voltage difference between M1 and M2, k is the Boltzmann constant, and q is the charge of the electron.

Substituting the above values into the equation for compensation voltage V _COMP is as follows.

V _COMP = (1/2) * [(3/2) * V _BGR0 + (mkT ₀ /q) * lnN + △V _TH ] + (1/2) * [(3/2) * V _BGR0 *α _BGR + (mk/q) * lnN + △V _TH *α _△VTH ] * △T

At this time, when (1/2) * [(3/2) * V _BGR0 + (mkT ₀ /q) * lnN + △V _TH ] is replaced with V _COMP0 (N), the above equation can be organized as follows. .

V _COMP = V _COMP0 (N) * [1 + (1/2) * ((3/2) * V _BGR0 *α _BGR + (mk/q) * lnN + △V _TH *α _△VTH ) / (( 1/2) * ((3/2) * V _BGR0 + (mkT ₀ /q) * lnN + △V _TH )) * △T]

At this time, [(1/2) * [(3/2) * V _BGR0 *α _BGR + (mk/q) * lnN + △V _TH *α _△VTH ]] / [(1/2) * [(3 /2) * V _BGR0 + (mkT ₀ /q) * lnN + △V _TH ), among the terms in the denominator, the (3/2) * V _BGR0 term that is unrelated to the N value is dominant, and among the terms in the numerator, Since the (mk/q) * lnN term related to the N value is dominant, the temperature coefficient of the compensation voltage can increase as the N value increases.

For reference, [(1/2) * [(3/2) * V _BGR0 *α _BGR + (mk/q) * lnN + △V _TH *α _△VTH ]] / [(1/2) * [( 3/2) When * V _BGR0 + (mkT ₀ /q) * lnN + △V _TH ) is replaced with the temperature coefficient of compensation voltage α _COMP (N), the equation for compensation voltage V _COMP can be organized as follows.

V _COMP = V _COMP0 (N) * (1 + α _COMP (N) * △T)

For example, when the temperature coefficient of the compensation voltage V _COMP at the compensation voltage output terminal is α _COMP , the temperature coefficient of the switched capacitor is α _{C , and} the temperature coefficient of the oscillator is α By changing the ON or OFF state of the switch connected to each of the 1_1th bit setting transistor to the 1_m bit setting transistor, and the n_1 to n_1 bit setting transistor to the n_m bit setting transistor, the second current mirror is changed by the bit controller. As a result of monitoring the difference between the 2_1 radiation current generated at the first temperature and the 2_2 radiation current generated at the second temperature, the compensation voltage The temperature coefficient is given in the formula below:

α _COMP + α _C + α _XO = 0

The temperature coefficient of the compensation voltage when satisfying is determined as the optimal temperature coefficient and the corresponding set value of the bit controller (i.e., the specific ON/OFF state of the switch connected to each bit setting transistor inside the compensation voltage generator) can be maintained. there is.

For example, referring back to the temperature compensated current source 1000 of FIG. 3, when the temperature coefficient of the resistor 1500 including the switched capacitor and the oscillator is 20ppm/°C, the total resistance value of the resistors R _{avg and SWCAP} is formula below

R _{avg ,SWCAP = 1} / _[ f

Therefore, α _C + α _XO = -20ppm/°C.

In addition, when the control value of 4 bits is 0000, the temperature coefficient of the compensation voltage of the compensation voltage generator 1200 is 16ppm/°C, and the LSB (least significant bit) of the bit controller is 1ppm/°C, the control value When set to 0100, the temperature coefficient of the compensation voltage of the compensation voltage generator increases by 4ppm/°C to 20ppm/°C, so the formula α _COMP + α _C + _α A robust reference current can be generated.

As another example, unlike the above case where the value of the predetermined voltage supplied from the constant voltage source does not change according to the temperature change, when the value of the predetermined voltage supplied from the constant voltage source changes according to the temperature change, the constant voltage source When the temperature coefficient is α _BGR , as the 2_1st radiation current and the 2_2nd radiation current approach within the preset threshold range, the temperature coefficient of the compensation voltage is expressed in the formula below:

α _COMP + α _C + α _XO + α _BGR = 0

The temperature coefficient of the compensation voltage when satisfying is determined as the optimal temperature coefficient and the corresponding set value of the bit controller (i.e., the specific ON/OFF state of the switch connected to each bit setting transistor inside the compensation voltage generator) can be maintained. there is.

So far, the temperature compensation current source 1000 according to an embodiment of the present invention has been described, and below, a current measurement system including the temperature compensation current source 1000 according to an embodiment of the present invention will be described.

Figure 6 schematically shows a current measurement system including a temperature compensated current source 1000 according to another embodiment of the present invention.

Referring to FIG. 6, a current measurement system including a temperature compensated current source 1000 according to another embodiment of the present invention, (i) between the power supply unit 3000 and the load 5000 or between the power supply unit 3000 and charging (i) ) a shunt resistor 2000 located between the unit 4000, (ii) a temperature compensation current source 1000 that supplies a reference current to the shunt resistor 2000, and (iii) a power current and temperature compensation current source supplied from the power supply unit ( As the reference current supplied from (1000) passes through the shunt resistor (2000), it may include a voltage measuring unit (not shown) that measures a voltage value D _OUT corresponding to the voltage that drops in the shunt resistor (2000).

For reference, in FIG. 6, the charging unit 4000 and the load 5000 are shown together, but the current measurement system including the temperature compensation current source 1000 according to another embodiment of the present invention is limited to this configuration. It doesn't work. In addition, in FIG. 6, the power supply unit 3000 is shown as a battery pack for convenience, but the present invention is not limited thereto.

For example, if the power unit 3000 is not a battery pack, a separate charging unit 4000 for charging the power unit 3000 may not be necessary, and therefore the charging unit 4000 is not included in the current measurement target circuit. Instead, only the load (5000) can be included.

As another example, if the power supply unit 3000 is a battery pack, a separate charging unit 4000 may be needed to charge the power supply unit 3000, so the current measurement target circuit may include only the charging unit 4000 or the charging unit (4000) and load (5000) may be included.

For reference, as previously explained, the shunt resistance R _S (2000) has a very small value (e.g., a value of 5 mΩ or less), so the voltage across the shunt resistor can be amplified using an amplifier as shown in FIG. 6. It may be possible. However, in order to measure the current using a current measurement system including the temperature compensated current source 1000 according to another embodiment of the present invention, voltage amplification through an amplifier is not necessarily necessary, and the voltage measured across the shunt resistor is It is also possible to measure the current flowing through the shunt resistor 2000.

Meanwhile, the formula below regarding the measured current I^ in FIG. 2

I^ = (V _I / V _R ) * I _R

By applying , the formula for the measured current I _BAT ^ in FIG. 6 is summarized as follows.

I _BAT ^ = (D _BAT (T) / D _R (T)) * I _R

For reference, (i) D _BAT (T) means that as the power current from the power supply unit (e.g., battery pack) 3000 passes through the shunt resistor 2000, the voltage that drops in the shunt resistor 2000 drives the amplifier. After being amplified through, it is a voltage measured by the voltage measurement unit through the ADC, and (ii) D _R (T) is the reference current from the temperature compensation current source 1000 according to an embodiment of the present invention through the shunt resistance (2000). ), the voltage that drops in the shunt resistor (2000) is amplified by an amplifier and then passed through an ADC and measured by a voltage measuring unit, (iii) I _R is the voltage from the temperature compensated current source (1000). This is the value of the reference current.

And, D _BAT (T) and D _R (T) can each be calculated according to the formulas below.

D _BAT (T) = I _BAT * R _S (T) * A(T) * (2 ^N / V _ref (T))

D _R (T) = I _R * R _S (T) * A(T) * (2 ^N / V _ref (T))

Therefore, the formula for the current I _BAT ^ to be measured can be organized as follows.

I _BAT ^ = [D _BAT (T) / D _R (T)] * I _R

= [(I _BAT * R _S (T) * A(T) * (2 ^N / V _ref (T))) / (I _R * R _S (T) * A(T) * (2 ^N / V _ref (T)))] * I _R

= I _BAT

Referring to the above formula, it can be confirmed that even if the value of resistance, etc. changes depending on the temperature, regardless of this, the value of the measured current I _BAT ^ is accurately calculated as the value of the actual current in the circuit I _BAT .

Meanwhile, referring back to the formula for the current I _BAT ^ to be measured, it is as follows.

I _BAT ^ = [D _BAT (T) / D _R (T)] * I _R

In the above formula, actual measured values are substituted for D _BAT (T) and D _R (T), and the reference current value of the temperature compensation current source 1000 is substituted for _IR .

However, according to a current measurement system including a temperature compensated current source 1000 according to another embodiment of the present invention, the measured voltage value measured through the voltage measurement unit is the power supply corresponding to the power current I _BAT supplied from the power supply unit 3000. It is determined as the sum of the voltage value D _BAT (T) and the reference voltage value D _R (T) corresponding to the reference current I _R supplied from the temperature compensated current source.

Therefore, in order to estimate the measured current value I _BAT ^, it may be necessary to distinguish between the power supply voltage value D _BAT (T) and the reference voltage value D _R (T).

To this end, a current measurement system including a temperature compensated current source 1000 according to another embodiment of the present invention includes a time division multiplexing method, a frequency division multiplexing method, and a code division multiplexing method. Power supply voltage value D _BAT (T) and reference voltage value D _R corresponding to the voltage that drops across the shunt resistor as the multiplexing current, which multiplexes the power current and reference current using a specific multiplexing method, passes through the shunt resistor. (T) can be identified using a specific multiplexing method, and the power supply current can be identified by referring to the reference current value I _R , the power supply voltage value D _BAT (T), and the reference voltage value D _R (T).

Below, the frequency division multiplexing method will be described as an example among the multiplexing methods. However, this is only for convenience of understanding, and the present invention is not limited thereto. Those skilled in the art will use the time division multiplexing method or the code division multiplexing method. Even without additional explanation, it will be possible to clearly understand the method of identifying the power supply voltage value D _BAT (T) and the reference voltage value D _R (T) by the time division multiplexing method or the code division multiplexing method.

Referring to Figure 7a, (i) apply low-pass filtering (LPF1) to the measured voltage value D _OUT to obtain the power supply voltage value D _BAT , and (ii) the power supply voltage value D _BAT is subtracted from the measured voltage value D _OUT . After applying high-pass filtering (HPF) to the frequency f _IR of the reference current, apply low-pass filtering (LPF2) to the demodulated value to obtain the reference voltage value D _R (T), (iii) the power supply By dividing the voltage value D _BAT (T) by the reference voltage value D _R (T) and multiplying it by the reference current value I _R , it can be confirmed that the measured current value I _BAT ^ corresponding to the power supply current I _BAT is calculated.

And, referring to FIG. 7b, the spectrum of the power supply voltage value D _BAT (T) and the reference voltage value D _R (T) according to frequency division multiplexing can be confirmed.

For reference, by making the frequency band of the current measurement system sufficiently wide, it can be designed so that there is no change in the transfer function of the system for multiplexed signals.

Meanwhile, in order to use the frequency division multiplexing method as described above, a frequency modulation process may be conducted first, which will be described with reference to FIGS. 8A to 8C.

FIGS. 8A to 8C schematically show the first current mirror 1600 shown in FIG. 3.

Referring to FIG. 8A, in order to modulate the first radiation current I _R corresponding to the reference current output from the temperature compensation current source 1000 according to an embodiment of the present invention, a switch is turned on/off according to each operation cycle. You can check P1 and P2.

Specifically, referring to FIG. 8b, the transistor on the left (e.g., PMOS transistor) generates a voltage corresponding to I ₀ at the X node. As P1 closes and P2 opens according to the switch operation cycle, the X node Since a voltage of is applied to the gate of the right transistor (e.g., PMOS transistor), if the sizes of the left and right transistors are the same, I ₀ is copied to the right transistor (i.e., I _R = I _O ).

Meanwhile, referring to Figure 8c, as P1 opens and P2 closes according to the switch operation cycle, the voltage of the is turned off so that I ₀ is not copied (i.e., I _R = 0).

Accordingly, by variously setting the operation cycles of the switches P1 and P2, the frequency of the first radiation current can be variously modulated.

In the above, the present invention has been described with specific details such as specific components and limited embodiments and drawings, but this is only provided to facilitate a more general understanding of the present invention, and the present invention is not limited to the above embodiments. , a person skilled in the art to which the present invention pertains can make various modifications and variations from this description.

Therefore, the spirit of the present invention should not be limited to the above-described embodiments, and the scope of the patent claims described below as well as all modifications equivalent to or equivalent to the scope of the claims fall within the scope of the spirit of the present invention. They will say they do it.

Claims (11)

In a temperature compensated current source,
A constant voltage source that supplies a predetermined voltage;
a compensation voltage generator connected to the constant voltage output terminal of the constant voltage source and generating a compensation voltage that is a temperature compensated voltage;
an amplifier whose first input terminal is connected to the compensation voltage output terminal of the compensation voltage generator;
A transconductor connected to the output terminal of the amplifier; and
a resistor connected to the transconductor and a second input terminal of the amplifier;
Including,
A temperature compensated current source that outputs a reference current that is resistant to temperature changes through the transconductor. According to paragraph 1,
a first current mirror connected to the transconductor, which is an output current mirror that copies the reference current to generate a first radiation current;
A temperature compensated current source further comprising: According to paragraph 2,
A current mirror for measurement that copies the reference current and generates a second radiation current while changing the temperature coefficient of the compensation voltage at the compensation voltage output terminal by a bit controller for varying the transistor included in the compensation voltage generator. As a result of monitoring the difference between the 2_1 radiated current generated at the first temperature and the 2_2 radiated current generated at the second temperature through the second current mirror, the second current mirror connected to the transconductor, the When the 2_1 radiation current and the 2_2 radiation current are close to within a preset threshold range, the temperature coefficient of the compensation voltage is determined as the optimal temperature coefficient and the corresponding set value of the bit controller is maintained, and the first current A temperature compensated current source characterized in that the first radiation current is generated through a mirror. According to paragraph 1,
The compensation voltage generator includes a first transistor group to an n-th transistor group, wherein the first transistor group to the n-th transistor group are connected in series, and n is an integer of 1 or more,
Each of the first transistor group to the n-th transistor group is connected in parallel to (i) each of the first to the n-th correction transistors, and (ii) each of the first to the n-th correction transistors, but each is ON. 1_1st bit setting transistor to 1_m bit setting transistor, n_1th bit setting transistor to n_m bit setting transistor, respectively, connected through respective switches for controlling /OFF, and (iii) the first correction A temperature compensation current source comprising a first non-compensation transistor to an n-th non-compensation transistor connected in series to each of the n-th compensation transistors. According to paragraph 4,
a first current mirror connected to the transconductor, which is an output current mirror that copies the reference current to generate a first radiation current;
Additionally including,
Each of the 1_1 bit setting transistor to the 1_m bit setting transistor, the n_1 bit setting transistor to the n_m bit setting transistor is operated by a bit controller for varying the transistor included in the compensation voltage generator. A second current mirror, which is a current mirror for measurement that copies the reference current and generates a second radiation current while changing the ON or OFF state of the connected switch - the second current mirror is connected to the transconductor - As a result of monitoring the difference between the 2_1 radiant current generated at a temperature of 1 and the 2_2 radiated current generated at a second temperature, the compensation when the 2_1 radiated current and the 2_2 radiated current are close to within a preset threshold range. A temperature compensated current source characterized in that the temperature coefficient of the voltage is determined as the optimal temperature coefficient and the corresponding setting value of the bit controller is maintained, and the first radiation current is generated through the first current mirror. According to paragraph 1,
A temperature compensated current source wherein the resistor includes a switched capacitor and an oscillator. According to clause 6,
The compensation voltage generator includes a first transistor group to an n-th transistor group, wherein the first transistor group to the n-th transistor group are connected in series, and n is an integer of 1 or more,
Each of the first transistor group to the n-th transistor group is connected in parallel to (i) each of the first to the n-th correction transistors, and (ii) each of the first to the n-th correction transistors, but each is ON. 1_1st bit setting transistor to 1_m bit setting transistor, n_1th bit setting transistor to n_m bit setting transistor, respectively, connected through respective switches for controlling /OFF, and (iii) the first correction It includes a first non-compensation transistor to an n-th non-compensation transistor connected in series to each of the n-th correction transistors,
When the temperature coefficient of the compensation voltage at the compensation voltage output terminal is α _COMP , the temperature coefficient of the switched capacitor is α _{C, and} the temperature coefficient of the oscillator is _α By changing the ON or OFF state of the switch connected to each of the 1_1 bit setting transistor to the 1_m bit setting transistor, the n_1 bit setting transistor to the n_m bit setting transistor, by the bit controller, A 2_1 radiated current and a second 2_1 radiated current generated at a first temperature through a 2nd current mirror, which is a measuring current mirror that copies the reference current and generates a 2nd radiated current - the 2nd current mirror is connected to the transconductor As a result of monitoring the difference between the 2_2 radiated current generated at temperature, the 2_1 radiated current and the 2_2 radiated current approach within a preset threshold range, and the temperature coefficient of the compensation voltage is expressed by the formula below:
α _COMP + α _C + α _XO = 0
A temperature compensated current source characterized in that, when satisfying, the temperature coefficient of the compensation voltage is determined as the optimal temperature coefficient and the corresponding set value of the bit controller is maintained. In clause 7,
When the value of the predetermined voltage supplied from the constant voltage source changes according to a temperature change, when the temperature coefficient of the constant voltage source is α _BGR , the 2_1 radiation current and the 2_2 radiation current are at the preset threshold. As it approaches within the range, the temperature coefficient of the compensation voltage is calculated using the formula below:
α _COMP + α _C + α _XO + α _BGR = 0
A temperature compensated current source characterized in that, when satisfying, the temperature coefficient of the compensation voltage is determined as the optimal temperature coefficient and the corresponding set value of the bit controller is maintained. According to paragraph 1,
The constant voltage source is a temperature compensated current source comprising any one of a bandgap reference, a charged capacitor, and a Zener diode. A current measurement system comprising the temperature compensated current source of claim 1, comprising:
A shunt resistor located between the power supply unit and the load or between the power supply unit and the charging unit, wherein the charging unit supplies power to the power supply unit;
the temperature compensated current source supplying the reference current to the shunt resistor; and
a voltage measuring unit that measures a measurement voltage value corresponding to a voltage that drops in the shunt resistor as the power current supplied from the power unit and the reference current supplied from the temperature compensation current source pass through the shunt resistor;
Including,
A current measurement system characterized in that the power current supplied from the power supply unit is identified by referring to the value of the reference current and the measured voltage value measured through the voltage measurement unit. According to clause 10,
The measured voltage value measured through the voltage measurement unit is determined as the sum of the power supply voltage value corresponding to the power current supplied from the power supply unit and the reference voltage value corresponding to the reference current supplied from the temperature compensation current source,
Multiplexing current - The multiplexing current is generated by using a specific multiplexing method among the time division multiplexing method, frequency division multiplexing method, and code division multiplexing method, and the power current and the reference current. is a multiplexed current - the power voltage value and the reference voltage value corresponding to the voltage that drops in the shunt resistor as it passes through the shunt resistor are identified using the specific multiplexing method, and the value of the reference current is, A current measurement system characterized in that the power current is identified by referring to the power supply voltage value and the reference voltage value.