KR20190129186A - Apparatus for controlling inverter - Google Patents

Abstract

Disclosed is an apparatus for controlling an inverter. According to the present invention, the control apparatus comprises: a command voltage generation unit for outputting a command voltage from a command frequency based on a voltage/frequency (V/f) operation, and outputting a pulse width modulation (PWM) voltage corresponding to the command voltage to the inverter; and a slip frequency determination unit for determining a slip frequency by using phase current of a motor driving the inverter and the command voltage. The slip frequency determination unit includes: a first converter converting the phase current of the motor into dq-axis current of a stationary coordinate system; a second converter converting the command voltage into a dq-axis voltage of the stationary coordinate system; a first estimation unit estimating a rotor magnetic flux canceling phase distortion caused by a filter from the dq-axis current, the dq-axis voltage, and a rotor command magnetic flux; a second estimation unit estimating a phase angle of the rotor magnetic flux from the rotor magnetic flux; a third converter converting the phase current of the motor into torque current and magnetic flux current of a rotary coordinate system using the phase angle; and a third estimation unit outputting an estimated slip frequency using the torque current, the magnetic flux current and a rotor time constant.

Description

Inverter Controller {APPARATUS FOR CONTROLLING INVERTER}

The present invention relates to an inverter control device.

In general, an inverter is a power converter that converts an AC power into a commercial AC power, converts it into a DC power, and then converts the AC power into an AC power suitable for the motor, and supplies the motor to the AC power. It is widely used in systems requiring variable speed operation.

Such inverters are based on power semiconductors, and various topologies are possible depending on the application field, and the magnitude and number of output voltages, the voltage synthesis scheme, and the like vary depending on the configuration. As an industrial inverter, a three-phase half bridge inverter is mainly used. The three phase half bridge inverter is a structure in which three single phase half bridge inverters are connected in parallel, and each half bridge is a basic circuit constituting an inverter called a pole, an arm, or a leg.

Induction motors, which are widely used in the industry, can be controlled by voltage / frequency (V / f) operation, so they are mainly used in fields such as fans, pumps and blowers that do not require fast dynamic characteristics in the operating range below the rated speed. .

However, since the slip frequency is generated according to the application in which the load varies, constant speed operation is impossible. In particular, in areas where constant speed operation is required, such as conveyors, the slip frequency must be compensated properly so that the actual operating speed matches the command speed. That is, in voltage / frequency operation, an inverter control is required to improve the speed error due to the slip frequency generation, thereby enabling a constant speed operation regardless of the load.

1 is a configuration diagram showing a general inverter control system.

The inverter controller 300 includes a command voltage generator 310 and a sleep frequency determiner 320 and outputs a three-phase PWM voltage to the inverter 100. The inverter 100 provides a three-phase output voltage to the motor 200 by the three-phase PWM voltage.

The command voltage generator 310 receives the command frequency ω _ref and generates a command voltage of the inverter 100 corresponding to the command frequency ω _ref based on the voltage / frequency (V / f) operation. At this time, the command voltage generator 310 generates a command voltage such that the ratio of the output voltage and the frequency is constant. The slip frequency determiner 320 determines a slip frequency corresponding to the speed error, and the inverter controller 300 reduces the speed error by adding the slip frequency to the command frequency.

2 is a detailed configuration diagram of the command voltage generator 310.

The voltage determining unit 311 determines the magnitude of the output voltage V _{v / f} from the operating frequency ω _{V / f} , and determines the phase θ _{v / f} of the output voltage through the integrator 312 and the trigonometric function application unit 313. By the multiplier 314, the command voltages V _{as_ref} , V _{bs_ref} , and V _{cs_ref} which are three-phase AC sine waves are output, and are synthesized by the PWM output unit 315 into three-phase PWM voltages corresponding to the command voltages. .

3 is an exemplary diagram for explaining a frequency-voltage relationship. The output voltage increases in proportion to the inverter output frequency, and the voltage determination unit 311 determines the magnitude of the output voltage V _{v / f} from the operating frequency ω _{V / f} according to the relationship. When the inverter 100 is initially started, the operating frequency ω _{V / f} of the inverter starts from 0, and thus outputs a small voltage, and outputs a voltage having a magnitude proportional to the frequency as the frequency increases. When the output frequency reaches the target frequency ω _ref , the frequency does not increase any more and it runs at constant speed.

4 is a detailed configuration diagram of the inverter 100 of FIG. 1.

The inverter unit 120 outputs an AC output voltage of three phases by using a DC voltage provided from the DC voltage providing unit 110 and supplies power to the motor 200 which is a three phase load. The output voltage of the three phases is determined according to the on / off state of the three phase switch of the inverter unit 120.

Two switches are connected in series with each phase leg, and each phase operates independently of each other to generate an output voltage. The output voltage of each phase is controlled to have a phase difference of 120 degrees from each other.

DC voltage providing unit 110 is composed of a capacitor or a battery, and maintains a constant voltage. The switch of the inverter unit 120 is a device that converts a DC voltage into an AC voltage.

The inverter controller 300 determines the switching state of the inverter unit 120 so that the motor 200 rotates at the same speed as the command frequency.

5 is a detailed configuration diagram of the slip frequency determiner 320 of FIG. 1.

The first coordinate conversion unit 321 converts the three-phase abc axis currents I _as , I _bs , I _cs to the stationary coordinate system dq axis currents I _dss , I _qss , and the second coordinate conversion unit 322 is the stationary coordinate system dq axis The currents I _dss and I _qss are _converted into rotary coordinate currents I _dse and I _qse . This is expressed as the following equation.

The product of the magnitude V _{V / f} of the inverter output voltage and the effective current I _qse is determined by the multiplier 323, and the output power determiner 324 determines the output power in consideration of the number of poles P. The calculation unit 325 determines the output torque T _load by dividing the calculated output power by the operating frequency ω _{V / f} , and applies the ratio of the rated slip frequency and the rated torque (326) through low band filtering (327). Determine the slip frequency.

At this time, the phase angle used to determine the effective current I _qse is the command phase angle θ _{v / f} with respect to the operating frequency.

The voltage / frequency control described above is a motor driving method that is widely used in the industry, and has the advantage of speed control and easy implementation. However, under high load operating conditions, the motor rotates differently from the speed input by the user due to an increase in the slip frequency, thereby lowering the speed accuracy.

To compensate for this, the inverter controller 300 properly compensates the slip frequency to increase the operating frequency of the inverter. As described above, the conventional slip frequency compensation calculates the output power and torque of the inverter and estimates the slip frequency through the slip frequency and torque ratio.

However, the torque is calculated by approximating the operating frequency of the inverter 100 and the rotational frequency of the actual motor 200 in the output torque calculation. In the low speed operation region, the operating frequency of the inverter 100 and the rotational frequency of the motor 200 are calculated. Since the error is relatively large and the influence of the loss of the motor 200 is large, accurate output power, torque and slip frequency calculation is difficult.

The technical problem to be solved by the present invention, by directly estimating the rotor flux in the low-speed operation region, by calculating the flux current and torque current from it to estimate the slip frequency through it, to accurately control the speed of the motor, It is to provide an inverter control device.

In order to solve the above technical problem, the inverter control apparatus of an embodiment of the present invention, based on the voltage / frequency (V / f) operation, outputs a command voltage from the command frequency, the pulse corresponding to the command voltage A command voltage generator for outputting a width modulation (PWM) voltage to an inverter; And a slip frequency determiner configured to determine a slip frequency by using a phase current of the motor driving the inverter and the command voltage. The slip frequency determiner includes: a first frequency that converts the phase current of the motor into a dq axis current of a stationary coordinate system; A conversion unit; A second converter converting the command voltage into a dq-axis voltage of a stationary coordinate system; A first estimation unit for estimating the rotor magnetic flux canceling the phase distortion by the filter from the dq axis current, the dq axis voltage, and the rotor command flux; A second estimation unit for estimating the phase angle of the rotor magnetic flux from the rotor magnetic flux; A third converting unit converting the phase current of the motor into a torque component current and a magnetic flux component current of a rotary coordinate system using the phase angle; And a third estimation unit outputting an estimated slip frequency by using the torque component current, the magnetic flux component current, and the rotor time constant.

In one embodiment of the present invention, the first estimation unit may estimate the rotor magnetic flux by the following equation.

이며, λ_{dqrs_VM}은 유도전동기 회전자 자속으로

이고, λ_{dqrs_ref}는 지령자속으로

,

일 수 있다. 이때, λ_rated는 정격자속이고, θ_est는 상기 위상각일 수 있다. 또, α는 상기 고역통과필터에 의해 왜곡된 위상일 수 있다.Where λ _{dqrs_est} is the estimated rotor _flux and F (s) is the highpass filter

Λ _{dqrs_VM} is the magnetic flux of the induction motor rotor

Λ _{dqrs_ref} is the command

,

Can be. In this case, λ _rated may be a rated magnetic flux and θ _est may be the phase angle. In addition, α may be a phase distorted by the high pass filter.

In one embodiment of the present invention, the second estimation unit, the fourth conversion unit for converting the stationary coordinate system rotor flux into a rotational coordinate system rotor flux; A proportional integral controller that outputs a frequency of the rotor flux by controlling the q-axis component of the rotor coordinate rotor to be zero; And an integrator for outputting a phase angle by integrating the frequency of the rotor magnetic flux.

In one embodiment of the present invention, the second estimation unit may further include a low pass filter for low-passing the estimated slip frequency and outputting a compensation slip frequency.

In one embodiment of the present invention, the integrator may integrate the sum of the frequency of the rotor flux and the compensation slip frequency.

In an embodiment of the present disclosure, the fourth transform unit may use the phase angle for coordinate transformation.

In an embodiment of the present invention, the third estimator may estimate the estimated slip frequency by the following equation.

In this case, I _torque may be torque current, I _flux may be flux current, and T _r may be a rotor time constant.

In one embodiment of the present invention, the command voltage generator may output a command voltage from an operating frequency corresponding to the sum of the command frequency and the compensation slip frequency.

As described above, the present invention has an effect of estimating the rotor flux and the phase angle, compensating the slip frequency based on the estimated phase angle of the rotor flux, and controlling the inverter to operate at a constant speed regardless of the load. have.

According to the present invention, the slip frequency is compensated precisely even in a region where the error between the operating frequency of the inverter and the rotational frequency of the transmission period is relatively large in the low speed operation region of the inverter, thereby controlling the drive to operate at a constant speed regardless of the load. .

1 is a configuration diagram showing a general inverter control system.
2 is a detailed configuration diagram of the command voltage generation unit.
3 is an exemplary diagram for explaining a frequency-voltage relationship.
4 is a detailed configuration diagram of the inverter of FIG. 1.
5 is a detailed configuration diagram of the slip frequency determining unit of FIG. 1.
6 is a schematic configuration diagram illustrating an inverter system according to an embodiment of the present invention.
7 is a configuration diagram of a slip frequency determination unit in an embodiment of the present invention.
FIG. 8 is a detailed circuit diagram of each component of FIG. 7.

In order to fully understand the constitution and effects of the present invention, preferred embodiments of the present invention will be described with reference to the accompanying drawings. However, the present invention is not limited to the embodiments disclosed below, but may be embodied in various forms and various changes may be made. However, the description of the present embodiment is provided to make the disclosure of the present invention complete, and to fully inform the person of ordinary skill in the art the scope of the present invention. In the accompanying drawings, for convenience of description, the size of the components is larger than the actual drawings, and the ratio of each component may be exaggerated or reduced.

Terms such as 'first' and 'second' may be used to describe various components, but the components should not be limited by the above terms. The term may only be used to distinguish one component from another. For example, a 'first component' may be referred to as a 'second component' without departing from the scope of the present invention, and similarly, a 'second component' may also be referred to as a 'first component'. Can be. In addition, singular forms also include plural forms unless the context clearly indicates otherwise. Unless otherwise defined, terms used in the embodiments of the present invention may be interpreted as meanings commonly known to those skilled in the art.

Hereinafter, an inverter control apparatus according to an embodiment of the present invention will be described with reference to FIGS. 6 and 7.

6 is a schematic configuration diagram illustrating an inverter system according to an embodiment of the present invention.

As shown in the figure, the system of one embodiment of the present invention comprises an inverter controller 3 for controlling the inverter 1, and an electric motor 2 driven by an AC voltage output from the inverter 1. Can be.

The inverter controller 3 may include a command voltage generator 10 and a sleep frequency determiner 20. Unlike the system of FIG. 1, the inverter controller 3 of the embodiment of the present invention may directly estimate the rotor magnetic flux and calculate the phase angle by using the stator voltage and current of the electric motor 2.

The command voltage generation unit 10 receives a frequency corresponding to the sum of the command frequency ω _ref and the compensation slip frequency ω _{slip_comp} as an operating frequency, and an inverter corresponding to the operating frequency based on the voltage / frequency (V / f) operation ( The command voltage of 1) can be generated. In this case, the command voltage generator 10 may generate the command voltage such that the ratio of the output voltage and the frequency is constant.

The detailed configuration of the command voltage generator 10 is as described with reference to FIG. 2.

That is, the magnitude V _{v / f} of the output voltage may be determined according to the frequency-voltage relationship of FIG. 3 from the operating frequency, and the phase θ _{v / f} of the output voltage may be determined by applying the integral and trigonometric functions. by it can be synthesized in a three-phase alternating-current sine wave the reference voltage V _{as_ref, bs_ref} V, 3-phase PWM voltage and outputs a V _{cs_ref,} those from which the reference voltage V _{abc_PWM.}

The slip frequency determiner 20 may determine the slip frequency by using the phase current of the motor 2 and the command voltage _{Vabc_ref} output from the command voltage generator 10. Although not shown, a current sensor may be used to measure the phase current of the electric motor 2.

The slip frequency determining unit 20 of the present invention estimates the rotor flux λ _{dqr_est} and the phase angle θ _est from the command voltage V _{abc_ref} and the motor current I _abcn , and compensates the slip frequency from the relationship between the current and the slip frequency.

7 is a configuration diagram of the slip frequency determination unit 20 in one embodiment of the present invention, Figure 8 is a detailed circuit configuration of each configuration of FIG.

As shown in the figure, the slip frequency determining unit 20 according to an embodiment of the present invention, the first coordinate conversion unit 21, the second coordinate conversion unit 22, the third coordinate conversion unit 23, The magnetic flux estimator 24, the rotor magnetic flux angle estimator 25, and the slip frequency output unit 26 may be included.

The first coordinate conversion unit 21 may convert the three-phase abc axis stator current into the stationary coordinate system dq axis current. The second coordinate conversion unit 22 may convert the three-phase abc axis stator voltage into the stationary coordinate system dq axis voltage. The coordinate transformed voltage V _dqss and current I _dqss are input to the rotor flux estimator 24, from which the stationary coordinate system rotor flux λ _{dqrs_est} can be estimated.

The rotor flux can be estimated by the following equation.

Where λ _rated is the rated flux and θ _est is the phase angle of the rotor flux.

In Equation 2, K _p and K _i are proportional integral controller gains, and λ _{dqrs_ref} is the command flux as in Equation 3. In Equation 2, the first term is the product of the second order high pass filter and the induction motor rotor flux λ _{dqrs_VM} , and the second term is the product of the second order low pass filter and the command flux. In one embodiment of the present invention, the cutoff frequency of each filter may be selected as the gain of the proportional integral controller.

가 비례적분 제어기의 전달함수이다. 본 발명이 속하는 기술분야에서 비례적분 제어기의 구성은 이미 널리 알려진 바와 같으므로, 그 상세한 구성의 설명은 생략하기로 한다. Proportional integral controller is a control method that uses the integral control in parallel with the proportional control integrating the error signal to produce a control signal,

Is the transfer function of the proportional integral controller. Since the configuration of the proportional integral controller is well known in the art to which the present invention pertains, a description of the detailed configuration will be omitted.

In equation (2), lambda _{dqrs_VM} is an induction motor rotor magnetic flux and is represented by the following equation.

)의 발산으로 인해 자속의 추정이 어렵다. 이를 위해, 본 발명의 일실시예의 회전자 자속추정부(24)는, 수학식 2와 같이 고역통과필터를 사용하여 회전자 자속을 추정하고, 회전자 지령자속을 저역통과필터를 통해 보완함으로써 개선할 수 있다.The rotor flux equation of Equation 4 above is calculated from the voltage equation of the induction motor, and it is easy to estimate the rotor at high speed, while the integrator is caused by the offset of voltage or current information in the low speed region.

), The flux is difficult to estimate. To this end, the rotor flux estimator 24 according to the embodiment of the present invention improves by estimating the rotor flux using a high pass filter as shown in Equation 2 and supplementing the rotor command flux through the low pass filter. can do.

However, accurate phase angle estimation becomes difficult because phase distortion occurs in the estimated flux by the use of the filter. Since phase angle information affects system performance, accurate phase angle information of the estimated flux is essential.

The cause of the phase distortion can be confirmed by the following equation.

Equation 5 is a reconstruction of Equation 2, where F (s) represents a second-order high pass filter. As can be seen from Equation 5, if there is an error between the magnetic flux and the command flux estimated through the voltage equation, the influence of F (s) is large, it can also cause phase distortion. Therefore, in order to cancel the phase distortion caused by F (s), it is possible to compensate the phase through the following equation.

In this case, ω _e is a synchronization frequency, and in the case of FIG. 7, ω _{e_est} . And α is the phase distorted by the filter.

Equation 6 is a phase compensation applied to Equation 5, and corresponds to the output of the rotor flux estimation unit 24 according to an embodiment of the present invention. As can be seen from Equation 6, it can be seen that the phase is compensated in the opposite direction by the distorted phase.

T (θ) of the rotor flux estimation unit 24 converts the stationary coordinate system variable into the rotational coordinate system variable, and can be defined by the following equation.

Here, X _ds and X _qs are static coordinate system variables, X _de and X _qe are rotation coordinate system variables, and θ represents a phase.

The third coordinate conversion unit 23 may convert the stationary coordinate system variable into a rotational coordinate system variable. In this case, the phase angle of the rotor flux estimated by the rotor flux angle estimator 25 may be used for coordinate transformation, and the torque current I _torque and the flux current I _flux may be output instead of the active current.

The slip frequency output unit 26 may output a slip frequency through the torque component current, the magnetic flux component current, and the rotor time constant T _r as follows.

The slip frequency (estimated slip frequency) output as described above may be output as the compensated slip frequency ω _{slip_comp} through the low pass filter LPF of the rotor flux angle estimator 25. The compensation slip frequency corresponds to a speed error, and the inverter controller 3 according to an embodiment of the present invention determines the operation frequency by adding the compensation slip frequency to the command frequency, thereby allowing constant speed control regardless of the load.

In addition, the rotor magnetic flux angle estimator 25 can estimate the rotor magnetic flux angle θ _est from the rotor magnetic flux λ _{dqr_est} estimated by the rotor magnetic flux estimation unit 24.

The rotor flux angle estimator 25 may be configured in the form of a phase locked loop (PLL). The PLL is a system for controlling an output signal by using a phase difference between an input signal and a signal fed back from an output signal. The PLL adjusts the frequency of the output signal in accordance with the input signal.

The rotor magnetic flux angle estimator 25 may convert the stationary coordinate system rotor magnetic flux into the rotary coordinate system rotor magnetic flux (T (θ _est )).

)를 사용하여, 회전자 자속각 기준의 dq축 동기좌표계에서 회전자 자속의 q축 성분 λ_{qre_est}이 0이 되도록 제어할 수 있으며, 비례적분 제어기의 출력과 보상 슬립주파수를 합산하여 회전자 자속의 주파수 ω_{e_est}를 결정하고, 이를 적분하여 회전자 자속의 위상각 θ_est를 추정할 수 있을 것이다. In addition, the rotor flux angle estimator 25 has a proportional integral controller (

), The q-axis component λ _{qre_est} of the rotor flux can be controlled to be 0 in the dq-axis synchronous coordinate system based on the rotor flux angle, and the output of the proportional integral controller and the compensation slip frequency are summed to determine the rotor flux. The frequency ω _{e_est} can be determined and integrated to estimate the phase angle θ _est of the rotor flux.

According to an embodiment of the present invention, the rotor flux and the phase angle may be estimated, and the slip frequency may be compensated based on the estimated phase angle of the rotor magnetic flux, thereby controlling to operate at a constant speed regardless of the load.

According to one embodiment of the present invention, even in a region where the error of the operating frequency of the inverter 1 and the rotational frequency between the motor 2 is relatively large in the low speed operation region of the inverter, the slip frequency is accurately compensated for, regardless of the load. It can be controlled to drive at a constant speed.

Although embodiments according to the present invention have been described above, these are merely exemplary, and it will be understood by those skilled in the art that various modifications and equivalent embodiments of the present invention are possible therefrom. Therefore, the true technical protection scope of the present invention will be defined by the following claims.

1: inverter 2: electric motor
3: inverter controller 10: command voltage generator
20: slip frequency determination unit

Claims (8)

A command voltage generator for outputting a command voltage from a command frequency and outputting a pulse width modulation (PWM) voltage corresponding to the command voltage to an inverter based on a voltage / frequency (V / f) operation; And
A slip frequency determiner configured to determine a slip frequency by using a phase current of the electric motor driving the inverter and the command voltage, wherein the slip frequency determiner includes:
A first converter converting a phase current of the motor into a dq-axis current of a stationary coordinate system;
A second converter converting the command voltage into a dq-axis voltage of a stationary coordinate system;
A first estimation unit for estimating the rotor magnetic flux canceling the phase distortion by the filter from the dq axis current, the dq axis voltage, and the rotor command flux;
A second estimation unit for estimating the phase angle of the rotor magnetic flux from the rotor magnetic flux;
A third converting unit converting the phase current of the motor into a torque component current and a magnetic flux component current of a rotary coordinate system using the phase angle; And
And a third estimating unit configured to output an estimated slip frequency by using the torque component current, the magnetic flux component current, and the rotor time constant.
The inverter control apparatus of claim 1, wherein the first estimation unit estimates the rotor magnetic flux by the following equation.

Where λ _{dqrs_est} is the estimated rotor _flux and F (s) is the highpass filter

Λ _{dqrs_VM} is the magnetic flux of the induction motor rotor

Λ _{dqrs_ref} is the command

,

being. Where λ _rated is the rated flux and θ _est is the phase angle. Is the phase distorted by the high pass filter)
The method of claim 1, wherein the second estimation unit,
A fourth converting unit converting the static coordinate system rotor magnetic flux into the rotary coordinate system rotor flux;
A proportional integral controller that outputs a frequency of the rotor flux by controlling the q-axis component of the rotor coordinate rotor to be zero; And
Inverter control device including an integrator for outputting a phase angle by integrating the frequency of the rotor magnetic flux.
The method of claim 3, wherein the second estimation unit,
And a low pass filter configured to low pass the estimated slip frequency and output a compensated slip frequency.
The method of claim 4, wherein the integrator,
And a sum of the frequency of the rotor magnetic flux and the compensation slip frequency.
The method of claim 3, wherein the fourth conversion unit,
Inverter control device using the phase angle for coordinate transformation.
The inverter control device according to claim 1, wherein the third estimating unit estimates an estimated slip frequency by the following equation.

(I _torque is torque current, I _flux is flux current and T _r is rotor time constant.)
The method of claim 4, wherein the command voltage generation unit,
And a command voltage output from an operating frequency corresponding to the sum of the command frequency and the compensation slip frequency.

Images