KR20150112597A - Resonance coupled wireless power transfer method and system - Google Patents

Abstract

The present invention relates to a resonance-coupled wireless power transmitting method and a system thereof. According to an embodiment of the present invention, the wireless power transmitting method to match impedance by considering a distance of a load from a source side of the wireless power transmitting system includes a step (a) of storing a system parameter about the wireless power transmitting system - the system parameter includes at least one among a coil parameter, a coupling coefficient, and reflecting power by distance -; a step (b) of determining where to place the load in an over-coupled region or an under-coupled region by using the peak of the reflecting power reflected from the load side; a step (c) of primarily estimating a distance between the source and the load by using the system parameter and the determined region; a step (d) of determining a capacitor value of the source side according to the primarily estimated distance; and a step (e) of exploring for optimal impedance while changing the capacitor value, starting from the determined capacitor value.

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention [0001] The present invention relates to a resonance-

The present invention relates to a resonant-coupled wireless power transmission method and system, and more particularly, to a method and system capable of efficiently performing impedance matching.

As the number of wireless devices requiring charging increases, various researches on wireless power transmission have been conducted recently.

The wireless power transmission has an inductive coupling method and a resonant coupling method.

Inductively coupled and resonant coupled systems use a nonradiative attenuated alternating current signal present around the coil at the reactive near-field.

Most of the energy in the near field is present as magnetic energy because the loop type coil is mainly used, and the energy transfer is also mediated through the magnetic signal or the magnetic field. Therefore, both of these methods have very good permeability to an object other than a magnetic body, and can be utilized in the ground or water.

Inductive coupling can transmit from a few milliwatts to several tens of kilowatts in transmission power, and the transmission distance depends on the transmission capacity, but it is several centimeters below several hundred watts. The frequency band used is several hundreds kHz or less.

In the inductive coupling system, the transmission / reception section is designed to resonate in order to increase the transmission efficiency. Generally, the quality factor of the transmission / reception coil is as low as several tens.

Inductive coupling is applied to various applications. However, since the inductive coupling method uses a resonance coil having a low quality factor, it is very important to increase the transmission efficiency or to reduce the transmission distance in order to increase the coupling coefficient of the transmission coil and the transmission coil so that the center of the transmission coil coincides .

As described above, the inductive coupling method has a very small degree of freedom in the position between the transmitting and receiving coils.

Recently, the wireless power consortium (WPC) has announced the WPC group standard for interoperability for the commercialization of the wireless charging products of low power wireless devices based on the inductive coupling technology. The market is growing very rapidly, and the market size is growing rapidly.

The resonance coupling method has a very high transmission efficiency due to the resonance phenomenon of the resonance coil even if the distance between the transmission and reception resonance coils is large or the mutual inductance is small due to the displacement of the transmission and reception coils. . The resonant coupling technique can transmit transmit power from a few watts to less than a kilowatt of a few tens of centimeters to a few meters. Compared to the conventional inductive coupling method, the transmission distance can be greatly improved, and the degree of freedom in arranging the transmitting and receiving coils is very high, and a technique capable of charging multiple devices has been actively studied. In addition, since power transmission occurs only between devices having the same resonance frequency, unlike the microwave method, even if the transmission distance increases, heterogeneous electronic devices and human bodies existing between transmission and reception resonators are not affected. In recent years, studies have been made on power transmission systems of several kilowatts or more in order to apply resonance coupling technology to wireless charging of electric vehicles.

However, in the resonance coupling method, impedance adjustment is required according to the distance between the source and the load (wireless device). According to the related art, there is a problem in that it takes considerable time to estimate the distance or to estimate the distance.

This time requirement is a significant constraint on the freedom to move the location of the wireless device in real time.

SUMMARY OF THE INVENTION The present invention has been made in view of the above problems, and it is an object of the present invention to provide a resonant-coupled wireless power transmission method and system capable of effectively performing impedance matching.

According to a preferred embodiment of the present invention, there is provided a method of matching an impedance in consideration of a distance of a load at a source side of a wireless power transmission system, comprising the steps of: (a) Storing a parameter, the system parameter including at least one of a coil parameter, a coupling coefficient, and a reflected power according to distance; (b) determining whether the load is located in an over-coupled region or an under-coupled region using a peak of the reflected power reflected from the load side; (c) estimating a distance between the source and the load using the determined region and the system parameter; (d) determining a source-side capacitor value according to the primarily estimated distance; And (e) searching the optimum impedance while changing the capacitor value with the determined capacitor value as a starting point.

Wherein the source comprises a source loop and a first resonator disposed adjacent to the source loop, the load comprising a second resonator and a load loop adjacent to the second resonator, the coupling factor comprising a first resonator and a second resonator, Lt; RTI ID = 0.0 > resonant < / RTI >

In the step (b), when there are two peaks in the reflected power, it is determined that the load is located in the over-coupling region, and when there is one peak, the load is located in the under- .

Wherein the source side comprises a first capacitor bank comprising a plurality of first switches and a second capacitor bank comprising a plurality of second switches, wherein the plurality of first switches and the plurality of The second switch may be controlled to determine a first capacitor value of the first capacitor bank and a second capacitor value of the second capacitor bank.

The step (e) may search for the optimal impedance using a hill-climbing algorithm.

Wherein the step (e) comprises: generating a matrix having the first capacitor value as a row and the second capacitor value as a column, and generating a matrix having a first capacitor value and a second capacitor value determined in the step (d) The impedance can be searched.

Wherein the step (e) searches for a candidate point corresponding to a local optimum impedance at at least some of the eight points adjacent to the current landing point, and when the impedance at the candidate point does not change, the impedance at the candidate point Can be determined as the optimum impedance.

Wherein the step (e) searches for a candidate point corresponding to a local optimal impedance at at least some of the eight points adjacent to the current landing point, and when the impedance at the candidate point changes, 1 < / RTI > and the second capacitor value as starting points.

According to another aspect of the present invention, there is provided a computer-readable recording medium on which a program for performing the above-described method is recorded.

According to another aspect of the present invention, there is provided a wireless power transmission system including a source-side source loop and a first resonator, a second resonator on a load side, and a load loop, comprising: a first capacitor bank including a plurality of first switches; A tunable matching network comprising a second capacitor bank comprising a plurality of second switches; A directional coupler for causing a signal of a predetermined frequency to be radiated to the load side through the tunable matching network and receiving reflected power from the load side; A digital control unit for generating a control signal for impedance adjustment according to the reflected power; And a switch driver for performing on / off of the first switch and the second switch in accordance with the control signal, wherein the digital control unit uses the peak of the reflected power so that the load is over- coupled region and an under-coupled region, and searching for an optimal impedance using the determined region and the previously stored system parameters.

According to the present invention, impedance matching is efficiently performed through fast distance estimation between a source and a load, and high efficiency wireless power transmission can be performed even if the position of the wireless device moves freely in real time.

의 측정 결과를 나타낸 도면.
도 5는 d=15cm에서 측정된

을 도시한 도면.
도 6은 거리의 함수로서 입력 임피던스를 나타낸 도면.
도 7은 제1 및 제2 공진기의 코일 형상을 도시한 도면.
도 8a는 거리에 따른 주파수 스플리팅 및 주파수 스플리팅 현상이 일어나는 조건에서 거리에 따른

를 나타낸 도면.
도 9a는 미스얼라인먼트(mis-alignment) ρ가 없다고 가정할 때, d의 함수로서 측정된 커플링 계수와 계산된 커플링 계수를 비교한 도면이고, 도 9b는 d=15 및 30cm에서 ρ의 함수로서 k_{23 , Maxwell}와 k_{23 , means}를 비교한 도면.
도 10은 본 발명의 일 실시예에 따른 무선 전력 전송 시스템의 블록도.
도 11은 본 발명의 일 실시예에 따른 조정 가능한 인덕터 및 캐패시터 뱅크를 이용하여 구성된 임피던스 매칭 네트워크를 도시한 도면.
도 12는 d=15,40, 80cm에서 모의

를 도시한 도면.
도 13 내지 15는 본 발명의 일 실시예에 따른 임피던스 매칭을 위한 탐색 알고리즘을 도시한 도면.1 is a flowchart of a wireless power transmission process according to a preferred embodiment of the present invention;
2 illustrates a resonant coupled wireless power transmission system;
3 shows an equivalent circuit of the wireless power transmission system of Fig. 2; Fig.
Fig. 4 is a graph of the performance of a wireless power transmission system as a function of distance d for different k ₁₂ (= k ₃₄ )

Fig.
Figure 5 shows the measured values at d = 15 cm

Fig.
6 shows input impedance as a function of distance;
7 is a view showing coil shapes of first and second resonators;
FIG. 8A is a graph showing a relationship between a distance and a frequency in a condition where frequency splitting and frequency splitting phenomenon occur

Fig.
Fig. 9A is a view comparing the calculated coupling coefficient with the calculated coupling coefficient as a function of d, assuming that there is no mis-alignment rho, and Fig. K _{23 , Maxwell} and k _{23 , means} .
10 is a block diagram of a wireless power transmission system in accordance with an embodiment of the present invention.
Figure 11 illustrates an impedance matching network configured using adjustable inductors and capacitor banks in accordance with one embodiment of the present invention.
Fig. 12 is a graph showing the results of simulation at d = 15, 40,

Fig.
Figures 13 to 15 illustrate search algorithms for impedance matching according to an embodiment of the present invention.

While the invention is susceptible to various modifications and alternative forms, specific embodiments thereof are shown by way of example in the drawings and will herein be described in detail. It should be understood, however, that the invention is not intended to be limited to the particular embodiments, but includes all modifications, equivalents, and alternatives falling within the spirit and scope of the invention. Like reference numerals are used for like elements in describing each drawing.

Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the accompanying drawings. In order to facilitate a thorough understanding of the present invention, the same reference numerals are used for the same means regardless of the number of the drawings.

1 is a flowchart of a wireless power transmission process according to a preferred embodiment of the present invention.

The wireless power transmission process shown in FIG. 1 may be performed in a computing device that controls the source side in a wireless power transmission system including a source and a load (a wireless device to be charged).

Referring to FIG. 1, the wireless power transmission process according to the present invention may include system calibration (step 100), coarse search (step 102), and fine search (step 104).

The steps 100 to 104 are a process of estimating the distance between the source and the load and matching the impedance corresponding thereto.

Generally, in the impedance matching process, the impedance search for all the points can guarantee the optimum impedance matching, but it takes too much time. This time requirement is a significant constraint on the freedom to move the location of the wireless device in real time.

To this end, the present invention proposes an efficient impedance matching process by fast distance estimation of a source and a load.

In step 100, the parameters of the wireless power transmission system are determined. Here, the system parameters may include coil parameters, coupling parameters, and reflected power over distance.

In addition, step 100 may be a process of calculating and measuring system parameters.

These system parameters will be described below again.

According to one embodiment of the present invention, the coupling coefficient can be calculated through a predetermined equation, and the reflected power can be measured while changing the distance between the source and the load from 15 cm to 100 cm in 5 cm increments.

The system parameters determined in step 100 are stored in advance in the form of a table.

Step 102 is a process of roughly estimating the distance between the source and the load using the system parameter determined in step 100 and the signal reflected by the load after applying the signal at the source.

According to an embodiment of the present invention, in step 102, a reflection signal is used to determine whether the load is in an over-coupled or under-coupled region.

The resonant coupling scheme is paired with the transmission coil on the source side and the reception coil on the load side. Step 102 is a process for determining whether the reception coil is located in the over-coupling region or the under-coupling region.

More specifically, in step 102 coupling coefficients according to the distance of the source and the load can be obtained, and for this purpose, according to one embodiment of the present invention, when the source and the load are located close to each other, the frequency splitting splitting occurs.

By frequency splitting, two peaks are detected when the load side is located in the over-coupling region, otherwise one peak is detected. According to an embodiment of the present invention, it is determined whether the receiving coil on the load side is located in the over-coupling region or the under-coupling region through peak detection.

After the approximate distance is searched in step 102, a fine search process is performed.

Step 104 is a process of searching for the optimum impedance of the source and the load. According to an embodiment of the present invention, an optimal impedance matching process may be performed through the hill-climbing algorithm of the present invention .

According to one embodiment of the present invention, the source side includes a plurality of switches and capacitors, and the capacitor value can be adjusted by operating the switch according to the distance between the source and the load. If the distance between the source and the load is estimated through the precise search in step 104, the switch is operated to adjust the capacitor value, thereby performing optimum impedance matching.

Hereinafter, a wireless power transmission system and an impedance matching process according to an embodiment of the present invention will be described in detail with reference to the drawings.

FIG. 2 is a diagram showing a resonance-coupled wireless power transmission system, and FIG. 3 is a diagram showing an equivalent circuit of the wireless power transmission system of FIG.

2 to 3, the wireless power transmission system includes a source loop 200 on the source side and a first resonator 202 and a second resonator 204 on the receive side and a load loop 206 .

2 to 3, the signal source is labeled V _S , the source and load resistances are labeled R _S and R _L , respectively, and center-to-center misalignment between the two resonators 202 and 204 is ρ.

The source loop 200 and the load loop 206 are inductively coupled to the first resonator 202 and the second resonator 204 and the step-up (step-up) ) To form a transformer.

Due to the large turn ratio, R _S and R _L are transformed into a large effective resistance together with the first and second resonators 202 and 204, which produces a high quality factor required in a high efficiency radio power transmission system.

Efficiency in such a system is affected by various parameter changes, and the main parameters are 1) the distance between the coils, 2) center-to-center alignment, 3) the load change, and 4) the ambient medium.

In particular, 1), 2) and 3) will be mainly discussed below.

As shown in FIG. 3, the parameters of each resonator coil are lumped elements R _i L _i C _i (i = 1 to 4, 1 is a source loop, 2 is a first resonator, 3 is a second resonator , And 4 is a load loop).

The coupling coefficients between adjacent coils are k ₁₂ , k ₂₃ , k ₃₄ , where the cross coupling coefficient (k ₁₃ = k ₁₄ = k ₂₄ = 0) is ignored.

By applying Basic circuit theory, the following equation 1 is obtained.

When R _S and R _L are connected to the source loop 200 and the load loop 206, the quality factor of the loops is affected by these resistors. For the sake of simplicity, it is assumed that all the coils resonate at the same frequency ω ₀ = 2πf ₀ .

The impedance value (passive and reciprocal) is expressed by the following equation (2).

Where Q _i is the quality factor of coil-i. Equation 1 is solved to obtain the current I ₁ in the source loop 200 and the current I ₄ in the load loop 206.

An accurate representation of these currents is obtained by applying Equation (2) to Equations (3) and (4) below as shown in Equations (5) to (6).

또는 전송 계수)를 이용하여 평가된다. 2 to 3 can be analyzed using the two-port network theory, the transmission capability of the system shown in FIGS.

Or transmission coefficient).

이 전체 시스템의 효율을 정확하게 나타내는 것은 아닐지라도, 공진기의 손실, 이의 주파수 의존성은 벡터 네트워크에 의해 쉽게 추정될 수 있다. Although

Although this does not accurately represent the efficiency of the overall system, the loss of the resonator, its frequency dependence, can be easily estimated by the vector network.

가 수학식 6을 이용하여 수학식 7과 같이 얻어진다. At the resonant frequency

(7) using Equation (6).

의 사용은 네트워크가 양 포트에서 매칭되는 경우 전력 전송 효율의 계산이 가능하게 한다. Also,

Allows calculation of the power transfer efficiency when the network is matched at both ports.

을 이용하여 계산될 수 있다. When R _S and R _L are equal to the reference impedance (50 OMEGA), the power transfer efficiency is

. ≪ / RTI >

은 코일의 형상 및 위치에 의해 결정된다. Since the coupling factor k is a function of d and p

Is determined by the shape and position of the coil.

이 k₂₃(제1 공진기 및 제2 공진기의 커플링 계수)의 함수이기 때문에 최적 효율에 도달하기 위한 최적 커플링 계수 k₂₃, _opt가 존재한다.

Since there is a function of k ₂₃ (the coupling coefficient of the first resonator and the second resonator), there is an optimum coupling coefficient k ₂₃ , _opt for reaching the optimum efficiency.

At a given distance d, the combination of resonator parameters to reach the optimum efficiency is specified. To find such a condition, the derivative of Equation (7) is taken for k ₂₃ and set to 0 to derive the following result.

Equation 8 specifies one optimal coupling coefficient or distance d for a given quality factor.

When d is changed, at least one of k ₁₂ and / or k ₃₄ can be adjusted to satisfy equation (8).

의 측정 결과를 나타낸 것이다. Fig. 4 is a graph of the performance of a wireless power transmission system as a function of distance d for different k ₁₂ (= k ₃₄ )

. Fig.

를 증가시키기 위해, 더 작은 k₁₂가 요구되는 것을 확인할 수 있다. As d increases,

, It can be seen that a smaller k ₁₂ is required.

The values of the coupling coefficients k ₁₂ and / or k ₃₄ can be adjusted by changing the distance between the resonator and the loop.

However, a change in k ₁₂ and / or k ₃₄ due to mechanical tuning is inefficient.

According to one embodiment of the present invention, the input impedance Z _IN of the wireless power transmission system is investigated to avoid mechanical tuning.

The following expression for the input impedance can be obtained by using Equation (5).

및

를 이용하면 수학식 10을 얻을 수 있다. Depending on the low resistance and radiation loss in the loops 200, 206,

And

(10) can be obtained.

When k ₂₃ (or d) changes, equation (10) shows that the input impedance (Z _IN ) also changes nonlinearly correspondingly.

Further, the equation (11) is obtained from the condition for the input impedance matching to the R _S.

Equations (8) and (11) are similar to each other.

및

이면, 수학식 8과 11은 거의 동일해진다. If all the coils have a very high quality factor, i. E.

And

, The equations (8) and (11) become almost equal.

This equivalence is satisfied when using a high quality factor and is valid in a well-designed resonant coupled wireless power transmission system.

In one embodiment of the present invention, an electrical tuning approach is applied, not mechanical tuning, assuming that all coils have a very high quality factor.

It is necessary to simplify the system shown in Fig. 2 for the electrical tuning of the coupling coefficients, and for this, it is assumed that the cross coupling (k ₁₃ , k ₁₄ and k ₂₄ ) is ignored at a sufficiently long distance.

을 도시한 것으로서, 도 5에 도시된 바와 같이, 가까운 거리에서 교차 커플링 계수는 증가한다. Figure 5 shows the measured values at d = 15 cm

As shown in FIG. 5, the cross coupling coefficient increases at a close distance.

피크 주파수를 시프트시키는 것을 확인할 수 있으며, 교차 커플링의 강도는 주파수 시프트의 양으로 결정할 수 있다. Referring to FIG. 5, if the cross coupling is low and two

It can be confirmed that the peak frequency is shifted, and the strength of the cross coupling can be determined by the amount of the frequency shift.

In a system that includes cross coupling, the input impedance can be expressed as:

Figure 6 shows the input impedance as a function of distance.

To obtain the simulation data, the measured coil parameters are used as shown in Table 1 below using Equation (12).

Inductance
(μH) Resistance
(Ω) Resonance Freq.
(MHz) Q (unloaded)
@ 6.78MHz Source loop 1.33 0.5 6.75 113 Resonator-2 40.1 5.5 6.76 311 Resonator-3 39.5 5.2 6.78 324 Load loop 1.35 0.4 6.73 144

As shown in FIG. 6, in a wide range of distance, the measured data and the simulated data show good agreement with each other.

It is also confirmed that the input impedance Z _IN changes rapidly with distance.

Therefore, k ₂₃ satisfied at a predetermined distance (or input impedance) _Match is not applicable at other distances.

The results indicate that input impedance matching is required to ensure high efficiency over a wide range of distances.

Coupling coefficients and quality factors are important factors in determining power transfer efficiency. The inductance L _loop of the source and load loops 200 and 206 is obtained using the following equation.

The resistance R _loop of a single loop coil is the sum of the ohmic resistance and the radiation resistance expressed by the following equation.

는 각주파수이다. Where r ₁ is the radius of the loop, a ₁ is the radius of the wire cross section, μ ₀ is the permeability of free space, σ is the conductivity of copper, v _c is the speed of light

Is the angular frequency.

Fig. 7 is a view showing the coil shapes of the first and second resonators. Fig.

The inductance L _spiral of a spirally wound coil can be calculated using the Wheeler formula.

Where b is the depth of winding, r _m is the average radius, N is the number of turns, and all units are inches.

In this embodiment, the coil radius of the source and load loops is 20 cm, and the external variable capacitors in the source and load loops are carefully tuned using a network analyzer.

According to one embodiment of the present invention, self-capacitance is used which exists in the shape of the first and second resonators 202 and 204.

The inner radius r _m of the resonator shown in Fig. 7 is 25 cm, b = 5 cm, and N = 6.

All coils are made of copper wire with a diameter of 2a ₁ = 0.3 cm.

The source loop 200 and the load loop 206 are centered on the first resonator 202 and the second resonator 204 (k ₁₂ = k ₃₄ = 0.27) and ρ = 0.

Table 1 shows the parameters of the coil using an Agilent 8714ES network analyzer.

The inductance calculated using equations (13) and (15) is consistent with the measured data in the 4.5% range.

The unloaded Q values estimated using equation (14) are 1130 and 4880 for the loop coil and internal resonator, respectively.

By using the Neumann formula, the coupling coefficient (k _{23 , Neumann} , radius r ₂ , r ₃ and distance d) between the first resonator 202 and the second resonator 204 can be obtained by the following equation .

Here, N ₂ and N ₃ are the number of turns, and L ₂ and L ₃ are the inductances calculated using Equation (15).

In equation (16), the approximation r ₂ , r _{3 <} d is applied.

A more accurate representation of the coupling coefficient is obtained using the filament method (so-called Maxwell method).

The mutual inductance M of the two parallel single coils is expressed as: < EMI ID = 17.0 >

Where J ₀ and J ₁ are the 0th and 1st order Bessel functions.

Equation 17 does not include the wire radius a, and it is assumed that a / r ₂ and a / r ₃ are sufficiently small.

Using equation (17), the coupling coefficient k _{23 , Maxwell} of the two plural-wound coils (first resonator and second resonator) is obtained as follows.

이고,

이다. here,

ego,

to be.

To extract the coupling coefficient k _{23 , means} by measurement, we use the frequency splitting that occurs at the near distance as follows.

Where f _H and f _L are the higher (odd mode) and lower (even mode) split frequencies of the coupled resonator, respectively.

At long distances, the coupling weakens and frequency splitting disappears.

이 이용되며, 측정된

에서 도 2의 시스템 모델을 조정함에 의해 k_{23 , means}가 도출된다. In this case, the measured values in the frequency range of 4 to 10 MHz

Is used,

K _{23 , means} is derived by adjusting the system model of Fig.

8A is a diagram illustrating frequency splitting along a distance.

In Figure 8A, the range of frequency splitting is 5.92 to 8.13 MHz.

를 나타낸 도면이고, f_L=5.92, f_H=8.13이고, f₀=6.5MHz를 이용하여 측정된 거리에 따른

이다. FIG. 8B is a graph showing the relationship between the distance under the condition where the frequency splitting phenomenon occurs

And f _L = 5.92, f _H = 8.13, and f ₀ = 6.5 MHz.

to be.

는 f₀와 f_H보다 f_L에서 측정될 때 더 크다. Referring to Figure 8, at a distance of less than 40 cm, frequency splitting occurs,

Is larger when measured at f _L than f ₀ and f _H.

9A compares the calculated coupling coefficient with the measured coupling coefficient as a function of d, assuming there is no misalignment.

K _{23 , Neumann} calculated using equation (16) is larger than the actual value, while k _{23 , Maxwell} calculated using equation (18) corresponds to the measured data.

For the simulation, the parameters in Table 1 were used in the system of FIG.

FIG. 9B is a diagram comparing k _{23 , Maxwell} and k _{23 , means} as a function of ρ at d = 15 and 30 cm.

By the approximation in Equation 17, the calculated value is somewhat greater than the measured value.

Hereinafter, a configuration of a wireless power transmission system according to the present invention will be described in detail with reference to the drawings.

10 is a block diagram of a wireless power transmission system in accordance with an embodiment of the present invention.

A wireless power transmission system according to an embodiment of the present invention may include a transmission unit 1000, a reception unit 1050, and coil structures 200 to 206. [

The transmission unit 1000 includes an oscillator 1002, a power amplifier (PA) 1004, a directional coupler 1006, a tunable matching network 1008, a digital control unit 1014, Gt; 1024 < / RTI >

The oscillator 1002 generates a signal of a predetermined frequency. In this embodiment, a single frequency of 6.78 MHz can be used.

The power amplifier 1004 amplifies the signal output from the oscillator 1002 and outputs the amplified signal to the directional coupler 1006.

The signal input to the directional coupler 1006 is radiated through the tunable matching network 1008 and the directional coupler 1006 outputs the reflected power from the receiving side to the power detector 1010.

The voltage divider 1012 outputs a voltage level corresponding to the reflected power to the ADC 1016 included in the digital control unit 1014. [

The supply voltage for the power amplifier 1004 is adjusted by control of a digital programmable potentiometer 1018 and a DC-DC converter 1020 based on the information from the ADC 1016. [

The information received from the envelope detector 1022 is used to change the transmission voltage level to maintain a constant rectified voltage V _REC at the receiver 1050.

The digital control unit 1014 monitors the reflected power through the directional coupler 1006 and transmits the control signal to the switch driver 1024 for the tunable matching network 1008. [

In accordance with the control signal, the switch driver 1024 adjusts the capacitor value of the tunable matching network.

To facilitate the closed loop control operation, the receiver 1050 returns information about the power captured by the load shift keying modulator (LSK).

The receiver may include a full-bridge rectifier 1052, a regulator 1054, an LSK modulator 1056, a 10-bit ADC 1058, and a microcontroller 1060.

11 is a diagram illustrating an impedance matching network configured using an adjustable inductor and a capacitor bank.

To implement an adjustable inductor, a fixed inductor and a variable capacitor are placed in parallel. For accurate impedance matching, first identify parasitics in the board and switch.

These measurements are for L _par inductors in the L _fix series combination in the board, L _par = 60nH, and in each relay switch (open state) the parasitic capacitance is C _par = 15pF.

When describing these parasitic components, the equivalent inductance (L _eq ) is as follows.

은 병렬 LC 회로의 공진 주파수이고, L_eq의 동작 주파수(w)

이다. here,

Is the resonant frequency of the parallel LC circuit, and L _eq is the operating frequency (w)

to be.

In order to fine-tune the impedance, a small L _fix is preferred. However, a small L _fix causes narrow inductance in the tuning range.

In the design according to the present embodiment, the distance varies between 15 and 100 cm, and the magnitude of the reflection coefficient varies between 0.24 and 0.91.

In consideration of the trade-off between the attainable impedance range and the minimum impedance step, L _fix was set to 440nH.

And, when C _var1 is changed in 10 pF increments within the range of 10 to 1000 pF, L _eq varies in the range of 505 to 5398 nH.

Using a set of capacitor banks (10, 20, 40, 50, 100, 200, 400 and 500 pF), the capacitor values C _var1 and C _var2 vary from 10 to 1320 pF in 10 pF increments.

These capacitors are surface mount types with a rating of 100V, and the switches are implemented using electrical control relays that can be operated with switch power up to 60W (125VA).

Their set and release times are 3ms and 2ms with 5V operating voltage.

To control the capacitor value, 20 switches are used with 16 switches (SW _S1 to SW _S8 and SW _P1 to SW _P8 ) used for C _var1 and C _var2 .

The four switches, SW ₁ to SW ₄ , provide flexibility in implementing a matching network.

When SW ₂ = SW ₃ = 1 and SW ₁ = SW ₄ = 0, the matching network is bypassed, and when SW ₁ = SW ₃ = 1 and SW ₂ = SW ₄ = (Series-L / shunt-C configuration).

When Z _IN is changed by a distance or alignment change, the tunable matching network searches for C _var1 and C _var2 to allow the impedance to reach R _S.

In the serial-L / short-C arrangement, _Leq and C _var2 are obtained by the following equations.

The value required for C _var2 can be found using Equation (21) or (23) above.

의 특성을 살펴본다. For proper search algorithm selection for impedance matching, it is possible to use C _var1 and C _var2

.

In Fig. 11, C _var1 and C _var2 are changed to select the combination of the capacitors C _sn and C _pn (n = 1 to 8).

The combination of these capacitor values matches the impedance matching point.

이다. Fig. 12 is a graph showing the results of simulation at d = 15, 40,

to be.

It can be seen from FIG. 12 that one peak appears for each d when the tunable matching network sweeps through all of the capacitor values.

가 나타나는 것을 확인할 수 있다. Up to 80 cm wide range, one peak

Is displayed.

Investigation of all impedance matching points can guarantee optimal impedance matching, but it takes too much time.

This is because the relay switch is very slow and takes more than 8 minutes if the impedance seek point is 132x312.

Thus, according to the present invention, an efficient impedance matching method is proposed.

13 to 15 are diagrams illustrating a search algorithm for impedance matching according to an embodiment of the present invention.

Figs. 13 to 15 show the respective steps of Fig. 1 in detail.

13 is a system calibration process according to an embodiment of the present invention in which R _i , L _i , C _i , R _S , and R _L are measured (step 1300).

Further, the coupling coefficients k and d23 along the distance are calculated through Equation (18), and the calculation results are stored in the form of a table (Step 1302).

Further, a DC voltage (voltage according to the reflected power) with respect to the distance is measured (steps 1304 and 1306), and the voltage at the strong region and the voltage at the far region are measured respectively.

Next, the reflected power is measured (step 1308) while changing the distance from 15 to 100 cm in 5 cm increments in the matching network of the bypass arrangement (SW2 = SW3 = 1 (ON) and the other switch is OFF).

All of the system parameters are calculated and measured through steps 1300 to 1308, wherein the more accurate the system parameters, the better the starting point for the coarse search.

FIG. 14 is a detailed view illustrating a coarse search process according to an embodiment of the present invention.

Referring to FIG. 14, information on the measured reflected power in step 1308 is prestored within a range of 5.5 to 8.5 MHz (step 1400). Frequency scanning is performed in the range of 5.5 to 8.5 MHz in the coarse search process.

More specifically, frequency scanning is performed after setting f = f0 (step 1402), and voltage corresponding to the current reflected power is read during frequency scanning (step 1404).

Thereafter, the voltage corresponding to the reflected power is stored (step 1406), and it is determined whether there are two peaks in the reflected power (step 1408).

If two peaks are present, it is determined that the load is in an over-coupled region (Strong Region) and the measured data is referenced at Step 1304 (Step 1410).

Otherwise, if there is one peak, it is determined that the load is in an under-coupled region (weak region), and reference is made to the measured data in Step 1306 (Step 1412).

The distance between the source and the load is estimated through the above-described step 1410 or 1412 (step 1414).

Referring to the estimated distance and the table at step 1302 (step 1416), the coupling coefficient is estimated (step 1418).

Next, the input impedance Z _IN according to the estimated coupling coefficient is calculated (step 1420).

Step 1420 is a process of calculating the Z-parameter of Equation (12) in combination with the system parameters in Table 1.

Next, it is judged whether R _in is smaller than R _S (step 1422). If it is small, S1 and S3 are turned OFF, S2 and S4 are turned ON (step 1424), and in the opposite case, S1 and S3 are turned ON, S2 and S4 are turned OFF (step 1426).

Then, the value in the L-network is calculated (step 1428).

Step 1428 is a process of calculating input impedance using equation (12) and calculating C _var1 and C _var2 using equations (21) to (25).

FIG. 15 is a flowchart of a fine search process according to an embodiment of the present invention.

In the fine search, a process of evaluating a neighboring point in the direction of minimizing the reflected power is performed until the peak point is found, and the already visited point is removed to increase the search speed.

Also, the combination of C _var1 and C _var2 is matched to the s row and p column of the square matrix Z, where s and p vary in the range of 1 to 80, and C _var1 and C _var2 vary in the range of 10 to 1320 pF.

)을 설정한다(단계 1500).Referring to FIG. 15, using C _var1 and C _var2 calculated in the step 1428, a start matching point (

(Step 1500).

Next, the current position z is added to the list L (step 1502).

Thereafter, eight adjacent points A (z) are determined based on the current position (step 1504).

Here, A (z) is as follows.

A (z) and the list L are looked up to find an unvisited adjacent point R (z) (step 1506).

Next, the optimum point, z _opt , of the eight adjacent points of R (z) is found using the voltage (reflected power) detected at the directional coupler (step 1508).

It is determined whether z _opt searched in step 1508 is changed (step 1510), and if it remains unchanged, the fine search process is terminated.

Step 1508 searches for a candidate point at at least some of the eight points adjacent to the current landing point, and when the impedance at the candidate point does not change, the impedance at the candidate point is determined as the optimum impedance.

There may be eight adjacent points around the current landing point, where at least three points may be previously visited, in which case impedance measurements may be performed at five of the adjacent points. have.

If z _opt is changed, the already examined point R (z) is added to the list L (step 1512), the local optimal point z _opt is assigned as a new starting point (step 1514), and the process returns to step 1504.

Embodiments of the present invention may be implemented in the form of program instructions that can be executed on various computer means and recorded on a computer readable medium. The computer-readable medium may include program instructions, data files, data structures, and the like, alone or in combination. The program instructions recorded on the medium may be those specially designed and configured for the present invention or may be available to those skilled in the art of computer software. Examples of computer-readable media include magnetic media such as hard disks, floppy disks and magnetic tape; optical media such as CD-ROMs and DVDs; magnetic media such as floppy disks; Examples of program instructions, such as magneto-optical and ROM, RAM, flash memory and the like, can be executed by a computer using an interpreter or the like, as well as machine code, Includes a high-level language code. The hardware devices described above may be configured to operate as at least one software module to perform operations of one embodiment of the present invention, and vice versa.

It will be apparent to those skilled in the relevant art that various modifications, additions and substitutions are possible, without departing from the spirit and scope of the invention as defined by the appended claims. The appended claims are to be considered as falling within the scope of the following claims.

Claims (15)

A method of matching an impedance in consideration of a distance of a load at a source side of a wireless power transmission system,
(a) storing system parameters for the wireless power transmission system, the system parameters including at least one of a coil parameter, a coupling coefficient, and a reflected power according to distance;
(b) determining whether the load is located in an over-coupled region or an under-coupled region using a peak of the reflected power reflected from the load side;
(c) estimating a distance between the source and the load using the determined region and the system parameter;
(d) determining a source-side capacitor value according to the primarily estimated distance; And
(e) searching the optimum impedance while changing the capacitor value with the determined capacitor value as a starting point. The method according to claim 1,
The source comprising a source loop and a first resonator disposed adjacent the source loop, the load comprising a second resonator and a load loop adjacent to the second resonator,
Wherein the coupling coefficient is a coupling coefficient between the first resonator and the second resonator. The method according to claim 1,
In the step (b), when there are two peaks in the reflected power, it is determined that the load is located in the over-coupling region, and when there is one peak, the load is located in the under- The impedance matching method of the wireless power transmission system. The method according to claim 1,
Wherein the source side comprises a first capacitor bank comprising a plurality of first switches and a second capacitor bank comprising a plurality of second switches, wherein the plurality of first switches and the plurality of And controlling the second switch to determine a first capacitor value of the first capacitor bank and a second capacitor value of the second capacitor bank. The method according to claim 1,
Wherein the step (e) searches for the optimal impedance using a hill-climbing algorithm. 5. The method of claim 4,
Wherein the step (e) comprises: generating a matrix having the first capacitor value as a row and the second capacitor value as a column, and generating a matrix having a first capacitor value and a second capacitor value determined in the step (d) Method of impedance matching in a wireless power transmission system seeking an impedance. The method according to claim 6,
Wherein the step (e) searches for a candidate point corresponding to a local optimum impedance at at least some of the eight points adjacent to the current landing point, and when the impedance at the candidate point does not change, the impedance at the candidate point To an optimum impedance. The method according to claim 6,
Wherein the step (e) searches for a candidate point corresponding to a local optimal impedance at at least some of the eight points adjacent to the current landing point, and when the impedance at the candidate point changes, 1 < / RTI > and a second capacitor value as starting points. The method of claim < RTI ID = 0.0 > 1, < / RTI > A computer-readable recording medium on which a program for performing the method according to claim 1 is recorded. A radio power transmission system comprising a source-side source loop and a first resonator and a load-side second resonator and a load loop,
A tunable matching network comprising a first capacitor bank comprising a plurality of first switches and a second capacitor bank comprising a plurality of second switches;
A directional coupler for causing a signal of a predetermined frequency to be radiated to the load side through the tunable matching network and receiving reflected power from the load side;
A digital control unit for generating a control signal for impedance adjustment according to the reflected power; And
And a switch driver for performing on / off of the first switch and the second switch according to the control signal,
The digital control unit determines whether the load is located in an over-coupled region or an under-coupled region using the peak of the reflected power, And searching for an optimal impedance using the pre-stored system parameters.
11. The method of claim 10,
Wherein the digital control unit primarily estimates a distance between the source and the load using the determined region and the system parameter, and estimates a first capacitor value at a source side and a second capacitor value at a source side according to the primarily estimated distance, And searches the optimum impedance while changing the capacitor value with the determined first and second capacitor values as starting points. 11. The method of claim 10,
Wherein the system parameter comprises at least one of a coil parameter, a coupling coefficient, and a reflected power according to distance. 13. The method of claim 12,
The source comprising a source loop and a first resonator disposed adjacent the source loop, the load comprising a second resonator and a load loop adjacent to the second resonator,
Wherein the coupling coefficient is a coupling coefficient between the first resonator and the second resonator. 11. The method of claim 10,
The digital control unit determines that the load is located in the over-coupling region when there are two peaks in the reflected power, and when the peak exists, the load is located in the under-coupling region Wireless power transmission system. Wherein the digital control unit searches for the optimal impedance using a hill-climbing algorithm.

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