KR20150006772A - Power supply - Google Patents

Abstract

An embodiment of the present invention relates to regulation of a line current, which includes generating a feedback voltage by comparing a reference voltage with a line sensing voltage, and controlling a switching operation of a power switch using the feedback voltage. The reference voltage may be a voltage with a constant level, a voltage varying according to an output current, or a voltage following a sine wave for power factor correction. Another embodiment of the present invention relates to sensing an output current, which includes sensing the output current using a feedback voltage corresponding to a voltage between both terminals of an inductor connected to a power switch, a peak of current flowing through the power switch, and a switching cycle of the power switch. According to the embodiments, a line current may be regulated in a desired waveform. Also, information regarding an output current may be obtained without an additional resistor or subsidiary coil.

Description

Power supply {POWER SUPPLY}

Description of embodiments of the invention relates to a power supply, particularly to line current regulation and output current sensing.

The reference value must be set to regulate the line current. Conventionally, line current regulation is performed under the open loop condition, which makes it difficult to set the reference value.

To regulate the output current, information about the output current is needed. A resistor or auxiliary winding is used to acquire the information.

Through the embodiments, the line current can be regulated in a desired form. In addition, information on the output current can be obtained without a separate resistor or auxiliary winding.

The line current regulation according to any one of the embodiments compares a reference voltage with a line sense voltage to generate a feedback voltage and uses the feedback voltage to control the switching operation of the power switch. The reference voltage is a constant level voltage, a voltage varying depending on an output current, or a voltage corresponding to a sine wave for power factor compensation.

According to another embodiment of the present invention, an output current is sensed by using a feedback voltage corresponding to a voltage across an inductor connected to a power switch, a peak of a current flowing through the power switch, and a switching period of the power switch .

A power supply apparatus according to an embodiment includes a power switch for controlling power transmission, a sense resistor through which a line current flows, and a comparator for generating a feedback voltage by comparing a reference voltage and a line sense voltage generated in the sense resistor, The duty of the power switch is determined using the feedback voltage.

The power supply further includes a duty determiner for generating a gate voltage in accordance with the feedback voltage, and the power switch switches according to the gate voltage.

Wherein the comparator includes a first stage to which the reference voltage is input and a second stage to which the line sense voltage is input, amplifies a voltage obtained by subtracting the input of the first stage from an input of the second stage, Voltage can be generated.

Wherein the power supply device further comprises a sawtooth generator for generating a sawtooth wave having a predetermined period and a PWM comparator for outputting a result of comparing the feedback voltage and the sawtooth wave, Control the duty.

Wherein the power supply further comprises an SR latch including a first stage for inputting an oscillator signal and a second stage for receiving an output of the PWM comparator, The switch is turned on and the power switch is turned off according to the input of the second stage.

The power supply further includes a PWM comparator for comparing the feedback voltage with a voltage corresponding to the current flowing in the power switch and controls the duty of the power switch in accordance with the output of the PWM comparator.

Wherein the power supply further comprises an SR latch including a first stage for inputting an oscillator signal and a second stage for receiving an output of the PWM comparator, The switch is turned on and the power switch is turned off according to the input of the second stage.

The commercial reference voltage may be at a constant level.

The power supply may further include an output comparator that generates the reference voltage according to a difference between a sense voltage corresponding to an output current of the power supply device and a predetermined output reference voltage.

The power supply may further include a clamping circuit for controlling a maximum value of the output reference voltage. The clamping circuit includes a diode including a cathode connected to a clamping voltage and an anode connected to an output terminal of the output comparator, and the maximum value of the reference voltage is controlled by the clamping voltage.

The power supply further includes a PWM comparator for comparing the feedback voltage with a voltage corresponding to a current flowing in the power switch, and the duty of the power switch can be controlled according to the output of the PWM comparator.

Wherein the power supply device further comprises a sawtooth generator for generating a sawtooth wave having a predetermined period and a PWM comparator for outputting a result of comparing the feedback voltage and the sawtooth wave, The duty can be controlled.

The commercial reference voltage may be constant when the phase angle of the input voltage is equal to or larger than a predetermined angle and may vary according to the phase angle of the input voltage when the phase angle of the input voltage is smaller than the predetermined angle.

Wherein the power supply further comprises an output comparator for generating a first feedback voltage according to a difference between a sense voltage corresponding to an output current of the power supply device and a predetermined output reference voltage, The AC input of the supply generates the reference voltage based on the rectified input voltage. The power supply may further include a multiplier that multiplies the first feedback voltage by a voltage corresponding to the input voltage to generate the reference voltage. The power supply device further includes a phase inverter that inverts the polarity of the input detection voltage that detects the input voltage to generate a negative input detection voltage, and the negative input detection voltage is a voltage corresponding to the input voltage.

 The line current flows to a rectifying circuit that rectifies the AC input from the ground to generate an input voltage, and the sense resistor is connected between the ground and the rectifying circuit, and the line sensing voltage may be a negative voltage.

The power supply device according to another embodiment includes a power switch including one end connected to an input voltage, an inductor connected to the other end of the power switch, and a feedback circuit for detecting a peak of the inductor current, feedback corresponding to a voltage across the inductor, And a sensing controller that senses an output current using a switching period of the power switch.

Wherein the sensing controller senses a peak of the inductor current by using a current flowing in the power switch at a time point when the power switch is turned off and detects a discharge period after the power switch is turned off using the feedback voltage And the output current can be calculated using the peak of the sensed inductor current, the discharge period, and the switching period of the power switch.

The sensing controller may calculate the output current based on a result of dividing the peak of the sensed inductor current by the discharge period and dividing the switching period of the power switch.

By controlling the line current to a desired waveform, the line current can be regulated or the power factor can be improved. In addition, the output current can be sensed without separate resistors and secondary windings.

1 is a diagram illustrating a power supply apparatus according to an embodiment of the present invention.
2 is a view illustrating a power supply apparatus according to another embodiment of the present invention.
3 is a view illustrating a power supply apparatus according to another embodiment of the present invention.
4 is a view illustrating a power supply apparatus according to another embodiment of the present invention.
5 is a waveform diagram illustrating an input voltage, a line current, a line sense voltage, a reference voltage, and a feedback voltage according to another embodiment of the present invention.
6 is a view illustrating a power supply apparatus according to another embodiment of the present invention.
7 is a diagram illustrating an input voltage, a line current, a first feedback voltage, and a reference voltage according to another embodiment of the present invention.
8 is a view illustrating a power supply apparatus according to another embodiment of the present invention.
9 is a waveform diagram showing an output reference voltage according to a phase angle.
10 is a view illustrating a power supply apparatus according to another embodiment of the present invention.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings so that those skilled in the art can easily carry out the present invention. The present invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. In order to clearly illustrate the present invention, parts not related to the description are omitted, and similar parts are denoted by like reference characters throughout the specification.

Throughout the specification, when a part is referred to as being "connected" to another part, it includes not only "directly connected" but also "electrically connected" with another part in between . Also, when an element is referred to as "comprising ", it means that it can include other elements as well, without departing from the other elements unless specifically stated otherwise.

A line current regulation method according to an embodiment of the present invention detects a line current and controls a duty according to a sensed line current. For example, as the line current decreases, the duty increases, and when the line current increases, the duty decreases. The line current can be regulated by such a duty control.

Specifically, a feedback voltage is generated based on the sensed line current, and the duty is controlled according to the feedback voltage. A voltage corresponding to the sensed line current may be compared with a predetermined reference voltage to generate a feedback voltage. The reference voltage may be a constant value or may vary depending on the output current of the secondary side.

Hereinafter, a power supply apparatus to which a line current regulation method according to an embodiment of the present invention is applied will be described with reference to the drawings.

1 is a diagram illustrating a power supply apparatus according to an embodiment of the present invention.

1, the power supply device 1 includes a dimmer 10, a rectifier circuit 20, a line sense resistor RL, a transformer 30, a duty determiner 40, (D1), an output capacitor (COUT), an inductor (L), a power switch (M), and a filter capacitor (CF).

The transformer 30, the rectifier diode D1, and the power switch M constitute a flyback converter. Although the power supply apparatus according to the embodiment of the present invention is implemented as a flyback converter, the present invention is not limited thereto.

The dimmer 10 sifters the AC input AC according to the dimmer angle.

The rectifying circuit 20 rectifies the set AC input AC. The inductor L is connected to the rectifying circuit 20 so that the line current IL flows through the inductor L. [ The filter capacitor CF is connected between one end of the primary winding CO1 and the primary ground to filter the switching current. The input voltage VIN is the voltage supplied to the flyback converter and is connected to the filter capacitor CF.

 The line current IL flows from the primary side ground to the rectifying circuit 20 through the line sense resistor RL. The line current IL is the sum of the current supplied to the flyback converter and the current flowing in the filter capacitor CF.

The input voltage VIN is transmitted to one end of the primary winding CO1 and the drain of the power switch M is connected to the other end of the primary winding CO1. A sense resistor (RS) is connected between the source of the power switch M and the ground. A gate voltage (VG) is input to the gate of the power switch (M). The power switch M controls the electric power transmitted from the input end to the output end. In Fig. 1, the input terminal is the primary side, the output terminal may be the secondary side, and the primary side and the secondary side are insulated.

The duty determiner 40 receives the feedback voltage VFB and generates a gate voltage VG that determines the duty of the power switch M. [ For example, as the feedback voltage VFB increases, the duty determiner 40 generates a gate voltage VG that increases the duty.

Since the power switch M is an n-channel transistor, the level at which the power switch M is turned on is a high level, and the level at which the power switch M is turned off is a low level.

The comparator 50 generates the feedback voltage VFB according to the result of comparing the line sense voltage VL with the reference voltage VR. The comparator 50 includes a non-inversion terminal (+) to which the line sense voltage VL is input and an inversion terminal (-) to which the reference voltage VR is input. The comparator 50 can generate the feedback voltage VFB according to the difference between the input of the non-inverting terminal (+) and the input of the inverting terminal (-). For example, the comparator 50 amplifies the voltage obtained by subtracting the reference voltage VR, which is the input of the inverting terminal (-), from the line sense voltage VL, which is the input of the non-inverting terminal (+ Can be generated.

Since the line current IL flowing through the line sense resistor RL flows from the primary side ground, the line sense voltage VL is a negative voltage. The reference voltage (VR) can also be set to a negative voltage.

The transformer 30 includes a primary winding CO1 and a secondary winding CO2. An anode electrode of the rectifier diode D1 is connected to one end of the secondary winding CO 2. When the rectifying diode D1 is turned on, a current flowing in the secondary winding CO2 is transmitted to the output capacitor COUT and the load.

1 shows an LED string 60 in which a plurality of LEDs are connected in series to a load. However, the embodiment of the present invention is not limited thereto.

When the power switch M is turned on, the line current IL flows in the primary winding CO1 and energy is stored in the primary winding CO1. During this period, the rectifier diode D1 is in a non-conducting state. When the power switch M is turned off and the rectifier diode D1 is turned on, the energy stored in the primary winding CO1 is transferred to the secondary winding CO2 and the current flowing in the secondary winding CO2 is rectified And flows through the diode D1.

As the line current IL increases, the line sense voltage VL, which is a negative voltage, decreases. In the embodiment of the present invention, the decrease in the negative voltage means that the amplitude of the voltage increases. As the decreasing line sense voltage VL becomes smaller than the reference voltage VR, the feedback voltage VFB decreases. Then the duty is reduced.

Conversely, when the line current IL decreases, the line sense voltage VL which is a negative voltage increases. In the embodiment of the present invention, the negative voltage means that the amplitude of the voltage decreases. The larger the line sensing voltage VL is larger than the reference voltage VR, the more the feedback voltage VFB increases. Then the duty increases.

In this manner, a regulation operation in which the line current is kept constant is performed. The duty control method can be variously modified.

For example, the power switch can be turned off according to the result of comparing the feedback voltage with the sawtooth wave.

2 is a view illustrating a power supply apparatus according to another embodiment of the present invention. In the power supply shown in Fig. 2, the line current is regulated in the voltage mode.

The same reference numerals are used for the same components as those of the power supply device 2 of the other embodiment of the present invention, and the description of the same components is omitted.

2, the power supply 2 includes a saw-tooth generator 51, a PWM comparator 52, and an SR latch 41.

The comparator 50 generates the feedback voltage VFB according to the result of comparing the line sense voltage VL with the reference voltage VR.

The sawtooth wave generator 51 generates a sawtooth wave VSAW having a predetermined period. The period of the sawtooth wave (VSAW) can be determined according to the switching period of the power switch. For example, the turn-on time of the power switch M and the start time of the rising of the sawtooth wave VSAW may be synchronized.

The PWM comparator 52 generates the OFF control signal OFFS1 according to the result of comparing the sawtooth wave VSAW with the feedback voltage VFB. The PWM comparator 52 includes a non-inverting terminal (+) to which the sawtooth wave VSAW is input and an inverting terminal (-) to which the feedback voltage VFB is input.

The PWM comparator 52 outputs a high level when the input of the non-inverting terminal (+) is above the input of the inverting terminal (-), and outputs a low level when the input is opposite. For example, the PWM comparator 52 generates a high-level off control signal OFFS1 when the rising sawtooth wave VSAW reaches the feedback voltage VFB.

The SR latch 41 includes a reset stage R to which an oscillator signal VOSC is input and a reset stage R to which an off control signal OFFS12 is input. When the input of the set S is at a high level, And generates the gate voltage VG of the low level when the input of the reset terminal R is at the high level. The SR latch 41 outputs the gate voltage VG through the output terminal Q. [

The SR latch 41 controls the turning on point of the power switch M in accordance with the oscillator signal VOSC and the turning point of the power switch M in accordance with the off control signal OFFS1.

For example, at the time when the oscillator signal VOSC rises to the high level, the SR latch 41 generates the gate voltage VG of the high level, and the OFF control signal OFFS1 rises to the high level The SR latch 41 generates the gate voltage VG of the low level. At the time when the next oscillator signal VOSC rises to the high level, the gate voltage VG again rises to the high level.

As described above, the feedback voltage VFB is determined in accordance with the line current IL, and the duty of the power switch M is determined in accordance with the feedback voltage VFB and the sawtooth wave VSAW, do.

As another variant for controlling the duty, the power switch can be turned off according to the result of comparing the drain current flowing through the power switch with the feedback voltage.

3 is a view illustrating a power supply apparatus according to another embodiment of the present invention. In the power supply shown in Fig. 3, the line current is regulated in the current mode.

The same reference numerals are used for the same components as those of the preceding embodiment among the components of the power supply device 3 of the other embodiment of the present invention, and description of the same components is omitted.

As shown in FIG. 3, the power supply 3 includes a PWM comparator 53 and an SR latch 42.

The comparator 50 generates the feedback voltage VFB according to the result of comparing the line sense voltage VL with the reference voltage VR.

The PWM comparator 53 generates the OFF control signal OFFS2 according to the result of comparing the sense voltage VCS with the feedback voltage VFB. The PWM comparator 53 includes a non-inversion terminal (+) to which the sense voltage VCS is input and an inversion terminal (-) to which the feedback voltage VFB is input. The sense voltage VCS is a voltage generated in accordance with the drain current ID flowing through the resistor RS when the power switch M is turned on.

The PWM comparator 53 outputs a high level when the input of the non-inverting terminal (+) is an input of the inverting terminal (-) or more, and outputs a low level when the inverting terminal For example, the PWM comparator 53 generates a high level off control signal OFFS2 when the rising sense voltage VCS reaches the feedback voltage VFB.

The SR latch 42 includes a reset stage R to which an oscillator signal VOSC is input and a reset stage R to which an off control signal OFFS2 is input. When the input of the set S is at a high level, And generates the gate voltage VG of the low level when the input of the reset terminal R is at the high level. The SR latch 42 outputs the gate voltage VG through the output terminal Q. [

The SR latch 42 controls the turning on point of the power switch M in accordance with the oscillator signal VOSC and the turning point of the power switch M in accordance with the off control signal OFFS2.

For example, at the time when the oscillator signal VOSC rises to the high level, the SR latch 42 generates the high level gate voltage VG, and the off control signal OFFS2 rises to the high level The SR latch 42 generates a low level gate voltage VG. At the time when the next oscillator signal VOSC rises to the high level, the gate voltage VG again rises to the high level.

As described above, the feedback voltage VFB is determined according to the line current IL, and the duty of the power switch M is determined according to the feedback voltage VFB and the sense voltage VCS, .

A line current regulation method according to another embodiment of the present invention determines a reference voltage according to an output current and determines a feedback voltage according to a result of comparing a line sensing voltage with a reference voltage.

In the previous embodiment, the reference voltage VR has a constant level, but in this embodiment, the reference voltage varies depending on the result of sensing the output current.

4 is a view illustrating a power supply apparatus according to another embodiment of the present invention. In the power supply shown in Fig. 4, the line current is regulated in the current mode.

The same reference numerals are used for the same components as those of the preceding embodiment among the components of the power supply device 4 of the other embodiment of the present invention, and explanations of the same components are omitted.

As shown in FIG. 4, the power supply 4 includes an output comparator 55, a line comparator 54, a PWM comparator 56, a clamping circuit 57, and an SR latch 43.

The output comparator 55 generates the reference voltage VR2 for generating the feedback voltage VFB using the LED sensing voltage VLED and the output reference voltage VR1 corresponding to the LED current ILED. The LED sensing voltage (VLED) may be received from the secondary side through an optocoupler, or may be sensed using an auxiliary winding on the primary side.

Specifically, the voltage directly sensing the LED current ILED flowing through the LED column 60 is transmitted to the primary side through an optocoupler (not shown), so that the LED sensing voltage VLED can be generated. Or the LED sense voltage VLED corresponding to the LED current ILED may be generated using the voltage across the auxiliary winding that is insulated from the secondary winding of the transformer 30 in the primary side.

The LED current ILED is an example of the output current of the power supply, and the embodiment is not limited thereto.

The output of the output comparator 55, that is, the reference voltage VR2, is determined such that the LED sense voltage VLED is regulated to the output reference voltage VR1. For example, the output comparator 55 generates an output corresponding to a result obtained by subtracting the LED sense voltage VLED from the output reference voltage VR1 input to the non-inverting terminal (+).

The output reference voltage VR1 is a reference voltage for determining the amount of current of the LED current ILED. The output reference voltage VR1 is constant when the phase angle of the AC input AC passing through the dimmer 10 is large and varies depending on the phase angle when the phase angle is smaller than the predetermined angle.

9 is a waveform diagram showing an output reference voltage according to a phase angle.

As shown in FIG. 9, the output reference voltage VR1 is constant when the phase angle is larger than the PAT, and the output reference voltage VR1 may be changed in proportion to the phase angle when the phase angle is smaller than the PAT.

When the LED sensing voltage VLED becomes smaller than the output reference voltage VR1, the reference voltage VR2 decreases. Then, the switching operation is controlled so that the output power rises. For example, the duty of the power switch M increases.

Therefore, as the reference voltage VR2 of the line sense voltage VL decreases, more line current IL flows, the LED current ILED increases, and the LED sense voltage VLED becomes higher than the output reference voltage VR1. . In this way, the LED current ILED is regulated.

The maximum value of the reference voltage VR2, which is a negative voltage, is controlled by the clamping circuit 57.

The clamping circuit 57 includes a diode 58 which includes a cathode connected to the clamping voltage VRM and an output connected to the output of the output comparator 55, i.e., the reference voltage VR2.

When the reference voltage VR2 increases and becomes greater than the clamping voltage VRM, the diode 58 becomes conductive, and the reference voltage VR2 becomes the clamping voltage VRM.

The line comparator 54 generates the feedback voltage VFB1 according to the result of comparing the line sense voltage VL with the reference voltage VR2.

The PWM comparator 56 generates the OFF control signal OFFS3 according to the result of comparing the sense voltage VCS with the feedback voltage VFB1. The PWM comparator 56 includes a non-inversion terminal (+) to which the sense voltage VCS is input and an inversion terminal (-) to which the feedback voltage VFB1 is input. The sense voltage VCS is a voltage generated in accordance with the drain current ID flowing through the resistor RS when the power switch M is turned on.

The PWM comparator 56 outputs a high level when the input of the non-inverting terminal (+) is an input of the inverting terminal (-) or more, and outputs a low level when the input is negative. For example, the PWM comparator 56 generates a high level off control signal OFFS3 when the rising sense voltage VCS reaches the feedback voltage VFB1.

The SR latch 43 includes a reset stage R to which an oscillator signal VOSC is input and a reset stage R to which an off control signal OFFS3 is input. When the input of the set S is at a high level, And generates the gate voltage VG of the low level when the input of the reset terminal R is at the high level. The SR latch 43 outputs the gate voltage VG through the output terminal Q. [

The SR latch 43 controls the turning on point of the power switch M in accordance with the oscillator signal VOSC and the turning point of the power switch M in accordance with the off control signal OFFS3.

For example, at the time when the oscillator signal VOSC rises to the high level, the SR latch 43 generates the gate voltage VG of the high level, and the off control signal OFFS3 rises to the high level Level gate voltage VG.

Hereinafter, operation of the power supply apparatus according to another embodiment of the present invention will be described with reference to FIG.

5 is a waveform diagram illustrating an input voltage, a line current, a line sense voltage, a reference voltage, and a feedback voltage according to another embodiment of the present invention.

As shown in Fig. 5, the input voltage VIN when the dimmer angle of the dimmer gradually decreases is shown. A peak of the line current IL is generated at a time point T1 at which the input voltage VIN is generated. The line sense voltage VL is controlled to be a negative voltage so as to be equal to the reference voltage VR2. When the input voltage VIN becomes zero voltage at the time point T2, the line current IL is not generated and the line sense voltage VL becomes zero voltage.

During the period T1-T2 during which the input voltage VIN is generated, the feedback voltage VFB1 is generated in accordance with the line sense voltage VL and the reference voltage VR2. Since the reference voltage VR2 is generated in accordance with the LED sense voltage VLED, the feedback voltage VFB1 varies according to the output current so that the output current regulation is performed in the closed loop CLOSE-LOOP.

For example, in FIG. 5, the peak of the input voltage VIN is 100V at the time point T12 of the period T1-T2, and the feedback voltage VFB1 at this time is 1V. Since the input voltage VIN decreases from 100 V to 99 V after the time T12 and the feedback voltage VFB1 is maintained at 1 V, the input voltage VIN decreases at the same duty ratio, The line sense voltage VL which is the voltage increases (the absolute value of the line sense voltage decreases).

Then, the line sense voltage VL rises slightly higher than the reference voltage VR2, and the feedback voltage VFB1, which is the output of the line comparator 54, slightly rises (for example, 1.1 V). Since the feedback voltage VFB1 slightly increases, the duty increases and the line current IL increases in a situation where the input voltage VIN is 99 V, and the line sense voltage VL, which is a negative voltage, And then approaches the reference voltage VR2. In this manner, the feedback voltage VFB1 is generated.

In Fig. 5, the variation of each of the line current IL and the feedback voltage VFB1 described above is so small that it is not shown.

The feedback voltage VFB1 is generated in accordance with the reference voltage VR2 which is a negative voltage while the line sense voltage VL is at zero voltage. For example, (-), the feedback voltage VFB1 can be a positive voltage and a constant level.

Next, a peak of the line current IL occurs at a time T3 when the input voltage VIN is generated. The line sense voltage VL is controlled to be a negative voltage so as to be equal to the reference voltage VR2. When the input voltage VIN becomes zero voltage at the time T4, the line current IL does not occur and the line sense voltage VL becomes zero voltage. During the period T3-T4 during which the input voltage VIN is generated, the feedback voltage VFB is generated in accordance with the line sense voltage VL and the reference voltage VR2.

The phase angle of the input voltage VIN of the period T3-T4 is reduced compared to the input voltage VIN of the period T1-T2. For example, it is assumed that the output reference voltage VR1 changes according to the relationship between the output reference voltage VR1 and the phase angle shown in Fig.

Then, as the phase angle decreases, the output reference voltage VR1 decreases and the LED current ILED also decreases, so that the LED sense voltage VLED also decreases. Then, in accordance with the feedback loop described in FIG. 4, the reference voltage VR2 changes so that the LED sense voltage VLED decreases in accordance with the output reference voltage VR1.

In addition, the feedback voltage VFB1 is determined such that the line sense voltage VL changes in accordance with the fluctuated reference voltage VR2. 5, the reference voltage VR2 of the period T3-T4 increases (the absolute value of the reference voltage decreases) as compared with the period T1-T2, and the line sense voltage VL becomes higher than the reference voltage VR2 The feedback voltage VFB1 is determined so as to be equal to the voltage VR2.

The feedback voltage VFB1 in the period T3-T4 is changed in overall shape together with the phase compared with the feedback voltage VFB1 in the period T1-T2. When the feedback voltage VFB1 in the period T1-T2 is superimposed on the feedback voltage VFB1 in the period T3-T4 as a dotted line, the feedback voltage VFB1 in the period T1-T2 shown by the dotted line in Fig. The feedback voltage VFB1 of the period T3-T4 is smaller than that of the feedback voltage VFB1.

The phase angle of the input voltage VIN of the period T5-T6 is reduced compared to the input voltage VIN of the period T3-T4. 5, the reference voltage VR2 of the period T5-T6 increases (the absolute value of the reference voltage decreases) and the line sense voltage VL becomes higher than the reference voltage VR2 The feedback voltage VFB1 is determined to be the same as the voltage VFB1.

The feedback voltage VFB1 in the period T5-T6 is also compared with the feedback voltage VFB1 in the period T3-T4 to change the overall shape together with the phase. When the feedback voltage VFB1 in the period T3-T4 is superimposed on the feedback voltage VFB1 in the period T5-T6 as a dotted line, the feedback voltage VFB1 in the period T3-T4 shown by the dotted line in Fig. The feedback voltage VFB1 of the period T5-T6 is smaller than that of the feedback voltage VFB1.

The increased reference voltage VR2 is clamped to the clamping voltage VRM so that the reference voltage VR2 is maintained at the clamping voltage VRM even if the phase angle of the input voltage VIN further decreases.

In the prior period, when the phase angle of the input voltage VIN decreases and occurs during the period T5-T6, the reference voltage VR2 reaches the clamping voltage VRM. Thereafter, the phase angle further decreases and the input voltage VIN is generated during the period T7-T8. As shown in Fig. 5, the reference voltage VR2 during the period T7-T8 is maintained at the same clamping voltage VRM as the reference voltage VR2 during the period T5-T6.

The feedback voltage VFB1 is determined according to the clamping voltage VRM and the line sense voltage VL even if the phase angle further decreases after the period T7-T8.

The feedback voltage VFB1 in the period T7-T8 in which the phase angle of the input voltage VIN further decreases after the period T5-T6 has the waveform shown in Fig. As shown in Fig. 5, the feedback voltage VFB1 is determined such that the line sense voltage VL becomes equal to the reference voltage VR2. A feedback voltage VFB1 during the period T7-T8 corresponding to the phase angle of the input voltage VIN in the waveform of the feedback voltage VFB1 during the period T5-T6 is generated. In FIG. 5, when the feedback voltage VFB1 in the period T5-T6 is superimposed on the feedback voltage VFB1 in the period T7-T8 as a dotted line, the magnitude is the same and the phase is different.

Since the line current IL during the period T7-T8 is also controlled in accordance with the clamping voltage VRM, the magnitude of the line current IL during the period T7-T8 is equal to the magnitude of the line current IL during the period T5- same.

Therefore, when the reference voltage VR2 is clamped to the clamping voltage VRM, the LED sensing voltage VLED is larger than the output reference voltage VR1, but the reference voltage VR2 can not be further increased, May not be regulated to be equal to the output reference voltage VR1. Then, it is controlled by an open-loop, not a close-loop.

4, the switching operation of the power switch M is controlled by comparing the feedback voltage VFB1 with the sensing voltage VCS. However, the embodiment is not limited thereto. As shown in Fig. 10, the off control signal OFFS4 may be generated by comparing the sawtooth wave VSAW instead of the sense voltage VCS.

10 is a view illustrating a power supply apparatus according to another embodiment of the present invention.

As shown in Fig. 10, the power supply 7 includes a PWM comparator 59. The PWM comparator 59 includes a noninverting terminal (+) to which the sawtooth wave VSAW is inputted and an inverting terminal (-) to which the feedback voltage VFB1 is inputted, and the input of the noninverting terminal (+ Level OFF control signal OFFS4, and in the opposite case, generates a low-level OFF control signal OFFS4.

The SR latch 47 controls the gate voltage VG for turning on the power switch M by the oscillator signal OSC input to the set S and supplies the off control signal OFFS4 to turn the power switch M off. The sawtooth wave (VSAW) and the oscillator signal (VOSC) have been described above, and a detailed description thereof will be omitted.

As described above, the power supply device according to another embodiment of the present invention controls not only the input current regulation but also the output current regulation, so that the switching operation is controlled.

The power factor improves as the line current coincides with the phase of the input voltage. In another embodiment of the present invention, the reference voltage may follow a sinusoidal wave rather than a constant level.

6 is a view illustrating a power supply apparatus according to another embodiment of the present invention.

As shown in Fig. 6, the power supply 6 detects the input voltage VIN and generates a reference voltage VR4 which is a sinusoidal wave that follows the input voltage VIN.

The same reference numerals are used for the same configurations described in the description of the embodiments described above, and a detailed description thereof will be omitted.

6, the power supply 5 includes two resistors RIN1 and RIN2, an output comparator 61, a multiplier 62, a polarity inverter 63, a line comparator 64, And a determiner 45.

The two resistors (RIN1, RIN2) are connected in series between the input voltage (VIN) and the primary ground. The input voltage VIN is divided by the two resistors RIN1 and RIN2 to generate the input detection voltage VIND.

The output comparator 61 generates the first feedback voltage VFB2 using the LED sense voltage VLED and the output reference voltage VR4 corresponding to the LED current ILED. The LED sensing voltage VLED may be received from the secondary side through an optocoupler or may be sensed using an auxiliary winding on the primary side, and a detailed description thereof is given in the previous embodiment with reference to FIG.

The output comparator 61 outputs the difference between the output reference voltage VR4 and the LED sense voltage VLED. For example, the output comparator 55 generates an output corresponding to a result obtained by subtracting the LED sense voltage VLED from the output reference voltage VR4 input to the non-inverting terminal (+). The output of the output comparator 61 becomes the first feedback voltage VFB2. The output reference voltage VR4 can be set to a voltage higher than the LED sense voltage VLED. Then, the first feedback voltage VFB2, which is the output of the output comparator 61, can be generated with a positive voltage that varies in the opposite direction to the direction of change of the output current ILED.

For example, when the output current ILED decreases, the LED sense voltage VLED decreases and the difference between the output reference voltages VR4 increases. Then, the first feedback voltage VFB2 increases. Conversely, when the output current ILED increases, the LED sense voltage VLED increases and the difference between the output reference voltages VR4 decreases. Then, the first feedback voltage VFB2 decreases.

When the peak of the input voltage VIN increases, the output current ILED may increase. Conversely, when the peak of the input voltage VIN decreases, the output current ILED may decrease.

The phase inverter 63 inverts the polarity of the input detection voltage VIND to generate an input detection voltage (hereinafter referred to as a negative input detection voltage VINDN) that is a negative voltage.

The multiplier 62 multiplies the negative input detection voltage VINDN by the first feedback voltage VFB2 to generate the reference voltage VR5.

The line comparator 64 generates the second feedback voltage VFB3 according to the result of comparing the line sense voltage VL with the reference voltage VR5.

The duty determiner 45 generates the gate voltage VG that controls the switching operation of the power switch M using the second feedback voltage VFB3. For example, the switching operation may be controlled according to a result of comparing the voltage sensed by the current flowing through the power switch M and the second feedback voltage VFB3, or the sawtooth wave and the second feedback voltage VFB3, The switching operation can be controlled according to the comparison result.

Hereinafter, reference voltage generation according to another embodiment of the present invention will be described with reference to FIG.

7 is a waveform diagram showing an input voltage, a line current, a first feedback voltage, and a reference voltage according to another embodiment of the present invention.

As shown in FIG. 7, since the line voltage is a negative voltage, the reference voltage is a negative voltage and follows a sinusoidal wave synchronized with the input voltage.

As shown in FIG. 7, when the peak of the input voltage VIN is at the V3 level, the reference voltage VR5 is generated as a sinusoidal wave having the lowest level V5. The reference voltage VR5 can replace the reference voltage VR and the reference voltage VR2 in the previous embodiment of the present invention. Then, as shown in Fig. 7, the line current IL is controlled to follow the sinusoidal wave having the peak I1, so that the power factor is improved. The first feedback voltage VFB2 is maintained at the level V7 corresponding to the output current ILED generated according to the input voltage VIN whose peak is at the V3 level.

The current flowing through the filter capacitor CF varies according to the input voltage VIN and the larger the input voltage VIN is, the greater the variation width of the current flowing through the filter capacitor may cause the line current IL to be distorted .

For example, although the current supplied to the flyback converter follows a sinusoidal wave, the line current is distorted by the current flowing through the filter capacitor, and the power factor is lowered.

However, in another embodiment of the present invention, the reference voltage VR5 is generated as a negative voltage sine wave according to the input voltage VIN to control the line current IL, The current IL can be controlled by a sinusoidal wave.

When the input voltage VIN rises and the peak is at the V4 level, the first feedback voltage VFB2 decreases to the level V8 with the increase of the output current ILED. Then, with the decrease of the level of the first feedback voltage VFB2 which is multiplied by the negative input detection voltage VINDN, the lowest level of the reference voltage VR5 increases to V6 (decreases its absolute value). According to the increased reference voltage VR5, the line current IL is maintained at a sinusoidal wave and its peak is reduced to I2.

The embodiments described so far are various modifications for controlling the line current to a desired waveform. The reference voltage is generated according to a desired waveform, and the line current is controlled according to the reference voltage to be regulated or the power factor is improved.

Hereinafter, a method of detecting an output current without a separate resistor for detecting an output current will be described in addition to the above embodiments.

8 is a view illustrating a power supply apparatus according to another embodiment of the present invention.

8, the power supply 6 includes a rectifier circuit 100, a line filter 110, an LED column 120, an LED current indirect sensing controller 200, A switch SW, an inductor L2, three resistors RSW, RF1 and RF2, a diode D2, and an output capacitor C3.

The rectifying circuit 100 rectifies the AC input AC and the rectified input is transmitted to the power switch SW through the line filter 110. [ The line filter 110 may filter the noise of the input rectified by the low-pass filter.

The line filter 110 includes two capacitors C1 and C2 connected between both ends of the rectifying circuit 100 and an inductor L1. An inductor L1 is connected between one end of each of the two capacitors C1 and C2 and the other end of each of the two capacitors C1 and C2 is connected to the ground.

The power switch SW includes a drain connected to an input (hereinafter referred to as an input voltage) transmitted through the rectifying circuit 100 and the input filter 110, a source connected to the inductor L2 and a gate connected to the gate voltage VG1, Lt; / RTI >

A resistor RSW for sensing the switch current is connected between the power switch SW and the inductor L2. The voltage of the contact to which the source of the power switch SW and the resistor RSW are connected is referred to as a switch current sensing voltage VS1.

The inductor L2 includes one end connected to the cathode of the power switch SW and the diode D2 and the other end connected to the ground.

A resistor RF1 and a resistor RF2 for generating a feedback voltage VFB1 are connected in series at both ends of the inductor L2.

Diode D2 is a free wheeling diode, which is connected between inductor L2 and LED string 120. The output capacitor C3 is connected in parallel to the LED string 120.

When the power switch SW is turned on, a current generated by the input voltage flows to the ground through the inductor L2. Then energy is stored in the inductor L2.

When the power switch SW is turned off, the diode D2 is turned on, and the current flowing through the inductor L2 is supplied to the output capacitor C3 and the LED string 120. At this time, a free wheeling current flows through the diode D2.

The reference potential GND of the LED current indirect sensing controller 200 is the voltage of the node N1 and the voltage of the node N1 is the one end voltage of the inductor L2 and the one end voltage of the inductor L2 is the voltage of the power switch (Vin) when the switch SW is turned on and -Vout when the power switch SW is turned off.

The feedback voltage VFB1 input to the LED current indirect sensing controller 200 is a voltage divided by the resistors RF1 and RF2 at both ends of the inductor L2 and is lower than the reference potential GND And a negative voltage at the LED current indirect sensing controller 200.

During the turn-on period of the power switch SW, since the voltage between the node N1 and the ground is Vin, the feedback voltage VFB1 is - (RF2 / (RF1 + RF2)) * Vin. During the turn-off period of the power switch SW, since the voltage between the node N1 and the ground is -Vout, the feedback voltage VFB1 is (RF2 / (RF1 + RF2)) * Vout.

The LED current indirect sensing controller 200 controls the switching operation of the power switch SW using the switch current sensing voltage VS1 and the feedback voltage VFB1.

In order to directly detect the output current ILED, in the prior art, a separate sense resistor is provided between one end of the diode D2 and the inductor L2, or an auxiliary winding insulated and coupled to the inductor L2 is used.

However, in the power supply apparatus according to another embodiment of the present invention, the output voltage can be sensed by feedback of the voltage across the inductor L2 without using a separate sense resistor or auxiliary winding.

The LED current indirect sensing detector 200 detects a current value of the inductor L2 after a turn-off of the power switch SW (hereinafter referred to as a discharge period) Tdis, a peak value of the inductor L2 current, The output current ILED can be calculated using the switching period TS of the switch SW.

For example, the peak value of the inductor L2 current corresponds to the peak value VSP of the switch current sense voltage VS1 since it is the current flowing in the power switch SW at the time point when the power switch SW is turned off. Since the switch current sensing voltage VS1 is input to the LED current indirect sensing sensor 200, the peak value VSP can be calculated.

The feedback voltage VFB1 rises to a positive voltage at the time point when the power switch SW is turned off and then remains above a predetermined level during the discharge period of the inductor L2 current. When the discharge period Tdis is terminated, the voltage across the inductor L2 begins to decrease by resonance. Since the feedback voltage VFB1 is input to the LED current indirect sensing controller 200, the LED current indirect sensing controller 200 can sense the discharge period Tdis using the feedback voltage VFB1. That is, the LED current indirect sensing controller 200 can set the period from the turn-off time of the input power switch SW to the time when the feedback voltage VFB1 becomes smaller than the predetermined threshold level to the discharge period Tdis.

The LED current indirect sensing controller 200 controls the switching operation. The switching period TS is set in the LED current indirect sensing controller 200.

Accordingly, the LED current indirect sensing controller 200 can calculate the output current ILED through Equation (1).

[Equation 1]

ILED = ((VSP * Tdis) / 2) / TS)

As such, another embodiment of the present invention is able to sense the output current without a separate resistor and auxiliary winding. If a resistor is used to sense the output current, there is a problem of power consumption in the resistor. Primary Side Regulation, which uses secondary windings to sense the output current, has the problem of rising costs due to secondary windings.

In another embodiment of the present invention, such a problem can be solved.

While the present invention has been particularly shown and described with reference to exemplary embodiments thereof, it is to be understood that the invention is not limited to the disclosed exemplary embodiments, It belongs to the scope of right.

The power supply (1, 2, 3, 4, 5, 6, 7)
The dimmer 10, the rectifier circuits 20 and 100,
Duty determinators 40 and 45,
A line sense resistor (RL), a transformer (30)
The rectifier diodes D1 and D2, the output capacitors COUT and C3,
Inductors L, 11, L2, power switches M, SW,
The filter capacitor (CF)
A sawtooth wave generator 51, a PWM comparator 52, 53, 56, 59,
SR latches 41, 42 and 43, output comparators 55 and 61,
Line comparators 54 and 64, a clamping circuit 57,
Line filter 110, LED current indirect sensing controller 200,
The resistors (RSW, RF1, RF2, RIN1, RIN2)
A multiplier 62, a polarity inverter 63,

Claims (21)

A power switch for controlling power delivery,
A sense resistor through which the line current flows, and
And a comparator that compares a reference voltage with a line sense voltage generated in the sense resistor to generate a feedback voltage,
And uses the feedback voltage to determine a duty of the power switch. The method according to claim 1,
Further comprising a duty determiner for generating a gate voltage in accordance with the feedback voltage,
And the power switch switches according to the gate voltage. The method according to claim 1,
The comparator comprising:
And a second stage in which the reference voltage is input and a second stage in which the line sense voltage is input, and the voltage obtained by subtracting the input of the first stage from the input of the second stage is amplified to generate the feedback voltage Power supply. The method of claim 3,
A sawtooth wave generator for generating a sawtooth wave having a predetermined period, and
Further comprising a PWM comparator for outputting a result of comparing the feedback voltage and the sawtooth wave,
And controls the duty of the power switch according to the output of the PWM comparator. 5. The method of claim 4,
Further comprising an SR latch including a first stage for inputting an oscillator signal and a second stage for receiving an output of the PWM comparator,
The SR latch turns on the power switch in accordance with the input of the first stage and turns off the power switch in accordance with the input of the second stage. The method of claim 3,
Further comprising a PWM comparator for comparing the feedback voltage with a voltage corresponding to a current flowing in the power switch,
And controls the duty of the power switch according to the output of the PWM comparator. The method according to claim 6,
Further comprising an SR latch including a first stage for inputting an oscillator signal and a second stage for receiving an output of the PWM comparator,
The SR latch turns on the power switch in accordance with the input of the first stage and turns off the power switch in accordance with the input of the second stage. 8. The method according to any one of claims 1 to 7,
The commercial power supply is a constant level reference voltage. The method according to claim 1,
And an output comparator for generating the reference voltage according to a difference between a sense voltage corresponding to an output current of the power supply device and a predetermined output reference voltage. 10. The method of claim 9,
And a clamping circuit for controlling a maximum value of the output reference voltage. 11. The method of claim 10,
The clamping circuit comprising:
A diode coupled to a cathode coupled to the clamping voltage and an anode coupled to the output of the output comparator,
Wherein the maximum value of the reference voltage is controlled by the clamping voltage. 12. The method of claim 11,
Further comprising a PWM comparator for comparing the feedback voltage with a voltage corresponding to a current flowing in the power switch,
And controls the duty of the power switch according to the output of the PWM comparator. 12. The method of claim 11,
A sawtooth wave generator for generating a sawtooth wave having a predetermined period, and
Further comprising a PWM comparator for outputting a result of comparing the feedback voltage and the sawtooth wave,
And controls the duty of the power switch according to the output of the PWM comparator. 14. The method according to any one of claims 9 to 13,
Wherein the commercial reference voltage is constant when the phase angle of the input voltage is equal to or larger than a predetermined angle and changes according to the phase angle of the input voltage when the phase angle of the input voltage is smaller than the predetermined angle. The method according to claim 1,
Further comprising an output comparator for generating a first feedback voltage according to a difference between a sense voltage corresponding to an output current of the power supply device and a predetermined output reference voltage,
Wherein the first feedback voltage and the AC input of the power supply generate the reference voltage based on the rectified input voltage. 16. The method of claim 15,
Further comprising a multiplier for multiplying the first feedback voltage by a voltage corresponding to the input voltage to generate the reference voltage. 17. The method of claim 16,
Further comprising a phase inverter that inverts the polarity of the input detection voltage that has detected the input voltage to generate a negative input detection voltage,
Wherein the negative input detection voltage is a voltage corresponding to the input voltage. The method according to claim 1,
The line-
Wherein the sensing resistor is connected between the ground and the rectifier circuit and the line sense voltage is a negative voltage. 1. A power supply apparatus for supplying an output current to a load,
A power switch including an end connected to an input voltage,
An inductor connected to the other end of the power switch, and
And a sensing controller for sensing the output current using a peak of the inductor current, a feedback voltage corresponding to a voltage across the inductor, and a switching period of the power switch. 20. The method of claim 19,
The sensing controller includes:
A peak of the inductor current is sensed by using a current flowing through the power switch at a time point when the power switch is turned off,
Detecting a discharge period after the turn-off point of the power switch using the feedback voltage,
And calculates the output current using a peak of the sensed inductor current, the discharging period, and a switching period of the power switch. 21. The method of claim 20,
The sensing controller includes:
Multiplying a peak of the sensed inductor current by the discharge period and calculating the output current based on a result of dividing the switching period of the power switch.

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