KR20090115729A - Automated noise measurement system - Google Patents

Abstract

A noise test measurement system configured to measure a noise component of a transmitted RF signal is described. The noise test measurement system may include an antenna, a low-noise amplifier, a local oscillator, a first coupler, a first variable phase-shifter, a first mixer, and a processor.

Description

Automated Noise Measurement System {AUTOMATED NOISE MEASUREMENT SYSTEM}

FIELD OF THE INVENTION The present invention relates to noise measurement, and more particularly to automated noise measurement systems used in wireless transmitters.

In general, the purpose of an electrical, electro-optical, electroacoustic communication system is to transmit an information-bearing signal (commonly known as a "baseband signal") over a communication channel separating the transmitter from the receiver. The term baseband signal (also known as baseband wave, baseband waveform, or "baseband") is used to designate a band of frequencies that represents a raw information-bearing signal delivered from an information source.

Efficient utilization of a communication channel typically requires shifting the frequency range of the baseband signal to another frequency range suitable for transmission over the communication channel, and then correspondingly shifting back to the original frequency range after reception. For example, a typical wireless system should typically be operated with signals with frequencies above 30 kHz, while baseband signals typically include frequencies in the radio frequency range (i.e., 20-20 kHz frequency range), and thus are part of the frequency-band shift. Forms should be used in wireless systems for satisfactory operation. A shift in the frequency range within a signal is defined as a process that changes some characteristic of a carrier signal (also known as a carrier wave, carrier waveform, or "carrier") in response to a modulated signal (also known as a modulated wave or modulated waveform). modulation "" process. The carrier signal is a signal having a continuous waveform (usually an electrical continuous waveform) whose characteristics can be modulated or applied by a second signal, which is typically an information-bearing signal (ie, a modulation signal). Typically, the carrier signal itself does not convey any information until it is changed to a predetermined form, such as having an amplitude, frequency or phase changed by the modulation signal. The changes induced by these modulated signals convey information through the resulting signal from the modulation process. In this situation, the baseband signal is referred to as a modulated signal and the result of the modulation process is referred to as a modulated signal (or modulated waveform). At the receiving end of the communication system, it is usually necessary for the modulated signal (i.e., the raw baseband signal, which is also a raw information-transfer signal) to be recovered to receive the information carried in the modulated signal. This is accomplished by using a process known as demodulation, which is the inverse of the modulation process.

Unfortunately, the noise of electrical, electro-optical and electroacoustic systems can disturb the amplitude and / or phase of the signal utilized and / or processed by these types of systems. In this situation, the term “noise” is used to designate unwanted signals that tend to disrupt the transmission and processing of the desired signals within electrical, electro-optical and electroacoustic systems. However, many of these types of systems are relatively insensitive to amplitude variations, so phase fluctuations (indicated as "phase-noise") are more problematic. For example, an oscillator is a device capable of generating an oscillation output signal having an oscillation frequency (eg, a sinusoidal signal commonly known as a "sine signal"). In general, oscillators typically include some kind of amplitude-limiting feature that attenuates the effects of any potential amplitude variations such that phase-noise is a major noise contributing factor to the final oscillator output signal.

Since noise, such as phase-noise, is an important factor in the construction and / or performance of electrical, electro-optical and / or electroacoustic systems, designers typically want to measure phase-noise for a given system. In the past, various approaches have been used to characterize the phase-noise of a given system. For example, an amplifier is characterized by first introducing an input signal of known frequency into the input of an amplifier and then measuring the amplified output signal resulting therefrom with a spectrum analyzer. In this case, the spectrum analyzer is a device capable of displaying the spectral waveform configuration of a predetermined electrical, acoustical or optical signal, including each amplitude magnitude and frequency for a given signal. Unfortunately, the measurement sensitivity of this approach is limited by the relatively poor sensitivity of the spectrum analyzer. Moreover, the measurement of phase-noise at a frequency value close to the carrier frequency of the signal which is the frequency of the carrier signal is difficult.

Unlike a spectrum analyzer, a phase locked discriminator system has a relatively good sensitivity, which usually allows measurements close to the carrier frequency. However, the configuration of the phase locked discriminator system is cumbersome and time consuming. In an attempt to alleviate this problem, an automated phase lock discriminator noise as described in US Pat. No. 6,793,372, the contents of which are incorporated herein by reference, to alleviate the cumbersome shortcomings in such a system. Test measurement systems (“APD systems”) have been developed. 1 shows a functional block diagram of an embodiment of an APD system 100 in which signal under test unit (“UUT”) 102 and spectrum analyzer 104 are in signal communication via signal paths 106, 108, 110. .

The APD system 100 includes a variable low-noise source 112, a variable phase shifter 114, a variable amplifier 116, a mixer 118, a variable low-noise matching amplifier 120, an analog-to-digital converter ( "ADC") 122 and controller 124. The UUT 102 is in signal communication with a variable low-noise source 112 and a variable low-noise amplifier 116 via signal paths 106 and 108, respectively. The mixer 118 is in signal communication with the variable phase shifter 114, the variable amplifier 116 and the variable low-noise matching amplifier 120 through each of the signal paths 126, 128, 130. The variable phase shifter 114 is in signal communication with the variable low-noise source 112 via a signal path 132. The ADC 122 is in signal communication with the variable low-noise matching amplifier 120 and the controller 124 via signal paths 134 and 136, respectively. The controller 124 passes through the signal paths 110, 138, 140, 142, and 144, respectively, to the spectrum analyzer 104, the variable low-noise source 112, the variable phase shifter 114, and the variable. In signal communication with an amplifier 116 and the variable low-noise matching amplifier 120.

In operation, the variable low-noise source 112 generates a UUT input signal 146 (low-noise carrier signal) for driving the UUT 102. The UUT 102 may be any device that wishes to make phase-noise test measurements, such as, for example, an amplifier, phase shifter, deplexer or other suitable device or system of devices. The UUT 102 receives the UUT input signal 146 via the signal path 106 and processes it to produce the UUT output signal 148. For example, if UUT 102 is an amplifier, UUT output signal 148 will be an amplified form of UUT input signal 146. The UUT output signal 148 is received via a signal path 108 and amplified by the variable amplifier 116 to generate a variable amplifier signal 150 that is passed through the signal path 128 to the mixer 118. . The variable low-noise source 112 generates a variable phase shifter input signal 152 that is passed through the signal path 132 to the variable phase shifter 14. The variable phase shifter input signal 152 is the same as the UUT input signal 146 and has the same frequency as the UUT input signal 146. The variable phase shifter 114 phase shifts the variable phase shifter input signal 152 by 90 degrees to produce a variable phase shifted signal 154 that is passed through the signal path 126 to the mixer 118. . In this form, the carrier signal of the variable amplifier signal 150 is removed from the mixer output signal 156 generated by the mixer 118 by the mixer 118. In order to keep the mixer output signal 156 within the appropriate dynamic range of the ADC 122, the mixer output signal 156 is passed through the signal path 130 to the variable low-noise matching amplifier 120 and processed by it. Thereby generating a variable low-noise matched output signal 158 that is passed through the signal path 134 to the ADC 122.

In order to remove the carrier signal of the variable amplifier signal 150, the variable phase shifted signal 154 must be in an orthogonal phase (ie, 90 degree shifted) with respect to the carrier signal. If a quadrature relationship between the variable amplifier signal 150 and the variable phase shifted signal 154 is not established, there will be a DC offset in the digital ADC output signal 160 from the ADC 122.

The controller 124 monitors the ADC output signal 160 and controls the variable phase shifter 114 using the variable phase shifter control signal 162 to control the variable phase shifter 114 with the variable amplifier signal 150. A quadrature phase relationship between the signals 154 is maintained. The variable phase shifter control signal 162 is transmitted to the variable phase shifter 114 via a signal path 140.

Removing the carrier signal from the variable amplifier signal 150 is such that the carrier signal (i.e., UUT input signal 146) and the phase shifted form of the carrier (i.e., the variable phase shifted signal 154) are combined with the mixer ( 118) depends on whether they have the same power upon reception. Thus, similar to the control of the variable phase shifter 114, the controller 124 responds to processing the ADC output signal 160 using the variable amplifier control signal 164 via the signal path 142. Controlling the variable amplifier 116 maintains the same power for the variable phase shifted signal 154 and the variable amplifier signal 150. These powers do not need to remain exactly the same, but may instead be within just enough range of each other to ensure linear operation of the mixer 118. Those skilled in the art will appreciate that the variable amplifier 116 may perform attenuation instead of only amplifying in response to the variable amplifier control signal 164. For example, if the UUT 102 is an amplifier, the variable amplifier 116 outputs the UUT output signal 148 to maintain both the variable phase shifted signal 154 and the variable amplifier signal 150 at a significant power quality. Should be attenuated. In addition, the controller 124 utilizes the variable low-noise matched amplifier control signal 166 to control the variable low-noise matched amplifier 120 through the signal path 144 to control the variable low-noise matched output signal ( 158 may be maintained within an appropriate dynamic range for the ADC 122.

In controlling the components for quadrature operation, the controller 124 removes the carrier signal from the ADC output signal 160 such that the ADC output signal 160 simply represents phase-noise. The phase-noise introduced by the variable low-noise source 112 is a variable amplifier 116 in which the UUT 102 is removed and the variable low-noise source 112 may simply occur via a delay line (not shown). This may be due to a correction operation that causes direct input to the furnace. Phase-noise generated in the ADC output signal 160 during calibration may be stored in a memory (not shown) associated with the controller 124. Thus, during testing of the UUT 102, the controller 124 (or spectrum analyzer 104 associated with the controller 124) may perform Fourier analysis of the ADC output signal 160 to determine phase-noise power. The measured phase-noise can then be adjusted by the phase-noise introduced by the variable low-noise source 112 to determine additional phase-noise supplied by the UUT 102.

The phase-noise measured at ADC output signal 160 depends on the frequency of UUT input signal 146. For example, UUT 102 may be very noisy at one frequency but less noisy at other frequencies. To measure phase-noise over a frequency range, the controller 124 utilizes the variable low-noise source command signal 168 via the signal path 138 to change the frequency of the UUT input signal 146 and obtain the phase obtained. Commands can be made to the variable low-noise source 112 to measure noise, change the frequency again, measure the phase-noise obtained after the frequency change, and the like. Advantageously such measurements are automatically and accurately performed without the manual intervention or tailing that may be required in conventional phase-noise test measurement systems.

Although APD system 100 has made significant advances in the art, unfortunately there are still some challenges as many factors are involved in properly biasing or driving a given component for optimal low-noise performance. For example, if the UUT 102 is an optical fiber link, the APD system 100 will typically require manual biasing for optimal performance so that the optical fiber link can be properly tested by the APD system 100 from the user. You won't be able to test your fiber link without your extensive manual participation.

For example, FIG. 2 shows a functional block diagram of a conventional fiber optic link 200. The optical fiber link 200 may include an input amplifier 202, a laser diode 204, an optical fiber channel 206, a photo detector 208 and an output amplifier 210. In this example, the laser diode 204 is in signal communication with the input amplifier 202 and the optical fiber channel 206 through signal paths 212 and 214, respectively. The photo detector 208 is in signal communication with the optical fiber channel 206 and the output amplifier 210 via signal paths 216 and 218, respectively.

In operation, the input amplifier 202 amplifies the electrical input signal Sin (t) 220 and generates a laser input signal 222 that drives the laser diode 204. Again, the laser diode 204 drives the input optical signal 224 into the optical fiber channel 206 (which may be optical fiber). After passing through the optical fiber channel 206, the output optical signal 226 is converted into an electrical signal 228 at the photo detector 208. The output amplifier 210 then amplifies the electrical signal 228 to provide an output signal Sout (t) 230. Unfortunately, many factors are involved in properly biasing the optical fiber link 200 for optimal performance in an APD system. For example, the matching of the input and output amplifiers 202 and 210 to the optical fiber link 200 and the laser diode 204 and the photodetector (including the proper biasing of the transistors in the input and output amplifiers 202 and 210). Properly biasing 208 is any factor that affects additional phase-noise into which optical fiber link 200 is introduced into output signal Sout (t) 230. However, the designer of the fiber optic link 200 does not have any intelligent way of setting these elements. Similar situations exist for the proper setting of variables in many other systems and devices.

In addition, although APD system 100 can be advantageously used to characterize the phase-noise performance of many different kinds of UUTs 102, the APD system 100 is unfortunately capable of delivering source signals, for example. There is a need for a non-wireless signal path to a variable low-noise source 112, such as a transmission line, which may include coaxial cables or coplanar waveguides. There are transmitters such as those used in cell phones, where access to source signals through these signal paths is very inconvenient or impossible.

Thus, there is a need for an automated system that can measure the phase-noise performance of a wireless transmitter through the reception and analysis of transmitted wireless signals.

A noise test measurement system configured to measure a noise component of a transmitted RF signal is disclosed. The noise test measurement system includes an antenna, a low-noise amplifier, a frequency source (which may be a local oscillator), a first coupler, a first variable phase shifter, a first mixer and a controller (which may be a processor). The antenna may be configured to receive the transmitted RF signal to provide a received RF signal and the low-noise amplifier to amplify the received RF signal to provide an amplified RF signal. The frequency source may be configured to provide a frequency reference signal (which may be a LO signal). The first coupler may be configured to combine a first version of the LO signal with the amplified RF signal to provide a combined RF signal, wherein the first variable phase shifter transmits a second version of the LO signal to a control signal. Responsive to phase shift with the quadrature LO signal. The first mixer may have an RF port configured to receive the combined RF signal and an LO port configured to receive the quadrature LO signal. The first mixer may be further configured to mix signals at its RF and LO ports to provide a first mixed output signal. The controller may be configured to analyze the digitized version of the first mixed output signal to generate the control signal and measure the noise component.

Other systems, methods, features and advantages of the present invention will become apparent to those skilled in the art upon reference to the following drawings and detailed description. All such additional systems, methods, features and advantages are intended to be included within the description, fall within the scope of the invention, and be protected by the appended claims.

The invention can be better understood with reference to the drawings below. The components in the figures are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. In the drawings, like reference numerals designate corresponding parts throughout the other views.

1 is a functional block diagram of an embodiment of a known automated phase locked discriminator measurement noise-test system (“APD system”) in signal communication with a unit under test (UUT) and a spectrum analyzer.

2 is a functional block diagram of a conventional optical fiber link.

3 is a functional block diagram of an embodiment of an automated phase-noise measurement system (“ANM system”) in accordance with the present invention.

4 is a functional block diagram of the UUT shown in FIG.

5 is a functional block diagram of another embodiment of an ANM system in accordance with the present invention.

6 is a functional block diagram of another embodiment of an ANM system in accordance with the present invention.

7 is a functional block diagram of an embodiment of the adjustable delay line shown in FIG.

8 is a functional block diagram of an embodiment of a direct down-conversion receiver.

9 is a functional block diagram of another embodiment of an ANM system in accordance with the present invention.

10 is a functional block diagram of another embodiment of an ANM system in accordance with the present invention.

11 is a flowchart illustrating an example of an operating method of the ANM system according to the present invention.

DETAILED DESCRIPTION In the following description of the preferred embodiments, reference is made to the accompanying drawings, which form a part hereof, and which illustrate certain embodiments in which the invention may be practiced. It is to be understood that other embodiments may be utilized and structurally modified without departing from the scope of the present invention.

Unit under test with at least one controllable variable

3 is a functional block diagram of an embodiment of an automated phase-noise measurement system (“ANM system”) 300 in accordance with the present invention. The ANM system 300 utilizes feedback to tune one or more controllable variables of the unit under test (UUT) 302 in response to phase-noise measurements to provide additional phase-noise provided by the UUT 302. Minimize. The UUT 302 is in signal communication with the ANM system 300 via signal paths 304, 306, 324, respectively.

The ANM system 300 includes a variable low-noise source 310, a variable phase shifter 312, a variable amplifier 314, a mixer 316, a variable low-noise matching amplifier 318, an analog-to-digital converter ( "ADC" 320 and controller 322. In this embodiment, the UUT 302 is in signal communication with the variable low-noise source 310, the variable amplifier 314 and the controller 322 via signal paths 304, 306, 324, respectively. Can be. The mixer 316 may be in signal communication with the variable phase shifter 312, the variable amplifier 314 and the variable low-noise matching amplifier 318 through signal paths 326, 328, 330, respectively. . In addition, the variable phase shifter 312 may be in signal communication with the variable low-noise source 310 via a signal path 332. The ADC 320 is in signal communication with both the variable low-noise matching amplifier 318 and the controller 322 via signal paths 334 and 336, respectively. The controller 322 is configured to drive the UUT 302, the variable low-noise source 310, the variable phase shifter 312, and the variable amplifier through signal paths 324, 338, 340, 342, and 344, respectively. 314, in signal communication with the variable low-noise matching amplifier 318. The controller 322 may be a controller device, a microcontroller, a processor, a microprocessor, an application specific integrated circuit (“ASIC”), a digital signal processor (“DSP”), or other similar programmable device.

The term "signal communication" refers to a signal (including both an information transmission signal and an information-non-transmission signal) and / or information that passes from one device to another and includes a wireless, wired, analog and / or digital signal path. Those skilled in the art will appreciate that it means any communication and / or electromagnetic, acoustic, digital or information transfer connection and / or coupling along a predetermined signal path between two devices. The "signal path" may be, for example, a physical path such as a conductor, electromagnetic transmission line and / or waveguide, and is a terminal, semiconductor or dielectric material or element attached and / or electromagnetically or mechanically coupled, or the like. It may be a similar physical connection or coupling.

In addition, the signal paths (in the case of electromagnetic propagation) are information-transfer through digital components through which communication information is passed from one device to another upon changing the digital format without necessarily passing through free space or direct electromagnetic connections. It may be a non-physical route, such as a route.

The UUT 302 may be any device for which a user desires a phase-noise test measurement, such as, for example, an amplifier, phase shifter, deplexer, fiber optic link, or other suitable device or system of devices. Unlike the UUT shown in FIG. 1, the UUT 302 of this embodiment includes at least one controllable variable that affects the phase-noise that the UUT 302 introduces into its output signal.

In operation, the variable low-noise source 310 generates a UUT input signal 346 (which may be a low-noise carrier signal) for driving the UUT 302. The UUT 302 may be any device for which the user desires a phase-noise test measurement, such as, for example, an amplifier, phase shifter, deplexer, fiber optic link, or other suitable device or system of devices. The UUT 302 receives the UUT input signal 346 via the signal path 304 and processes it to produce a UUT output signal 348. By way of example, if the UUT 302 is an amplifier, the UUT output signal 348 would be an amplified version of the UUT input signal 346. The UUT output signal 348 is received via the signal path 306 and amplified by the variable amplifier 314 to produce a variable amplifier signal 350 that is passed through the signal path 328 to the mixer 316. The variable low-noise source 310 also generates a variable phase shifter input signal 352 that is passed through the signal path 332 to the variable phase shifter 312. The variable phase shifter input signal 352 is the same as the UUT input signal 346 and has the same frequency as the UUT input signal 346. The variable phase shifter 312 generates a variable phase shifted signal 354 that is passed through the signal path 326 to the mixer 316 through a 90 degree phase shift of the variable phase shifter input signal 352. . In this form, the carrier signal of the variable amplifier signal 350 is removed from the mixer output signal 356 generated by the mixer 316. In order to maintain the mixer output signal 356 in the proper dynamic range of the ADC 320, the mixer output signal 356 is passed through a signal path 330 to a variable low-noise matching amplifier 318 whereby The processing produces a variable low-noise matched output signal 358 that is passed through the signal path 334 to the ADC 320.

To remove the carrier signal of the variable amplifier signal 350, the variable phase shifted signal 354 must be in quadrature phase (ie, 90 degree shifted) with respect to the carrier signal. If the quadrature phase relationship between the variable amplifier signal 350 and the variable phase shifted signal 354 is not established, there will be a DC offset in the digital ADC output signal 360 from the ADC 320.

Similar to the APD system shown in FIG. 1, the controller 322 monitors the ADC output signal 360 and controls the variable phase shifter 312 using the variable phase shifter control signal 362 to control the variable amplifier. An orthogonal phase relationship is maintained between the signal 350 and the variable phase shifted signal 354. The variable phase shifter control signal 362 is transmitted to the variable phase shifter 312 via a signal path 340.

Removing the carrier signal from the variable amplifier signal 350 is such that the carrier signal (ie UUT input signal 346) and the phase shifted version of the carrier (ie variable phase shifted signal 354) are combined with the mixer 316. Depends on whether they have the same power upon reception. Thus, similar to the control of the variable phase shifter 312, the controller 322 uses the variable amplifier control signal 364 via the signal path 342 in response to processing of the ADC output signal 360. By controlling 314, the same power is maintained for the variable phase shifted signal 354 and the variable amplifier signal 350. These powers do not need to remain exactly the same, but may only be within sufficient range of each other to ensure linear operation of the mixer 316. Again, those skilled in the art will appreciate that the variable amplifier 314 may perform attenuation in addition to amplifying in response to the variable amplifier control signal 364. For example, if the UUT 302 is an amplifier, the variable amplifier 314 may output the UUT output signal 348 to maintain both the variable phase shifted signal 354 and the variable amplifier signal 350 at a significant power quality. There may be times when you need to attenuate. The controller 322 utilizes the variable low-noise matching amplifier control signal 366 via the signal path 344 to control the variable low-noise matching amplifier 318 so as to control the variable low-noise matching amplifier 318. May be maintained within an appropriate dynamic range for the ADC 320.

In controlling the components for orthogonal operation, the controller 322 removes the carrier signal from the ADC output signal 360 such that the ADC output signal 360 merely represents phase-noise. The phase-noise introduced by the variable low-noise source 310 is a variable amplifier 314 in which the UUT 302 is removed and the variable low-noise source 310 can simply occur through a delay line (not shown). This may be due to a correction operation so that direct input to the furnace occurs. Phase-noise generated in the ADC output signal 360 during calibration may be stored in a memory (not shown) associated with the controller 322. Thus, during testing of the UUT 302, the controller 322 may perform Fourier analysis of the ADC output signal 360 to determine phase-noise power. The measured phase-noise can then be adjusted by the phase-noise introduced by the variable low-noise source 310 to determine additional phase-noise supplied by the UUT 302.

The phase-noise measured at ADC output signal 360 depends on the frequency of UUT input signal 346. For example, UUT 302 may be very noisy at one frequency but less noisy at other frequencies. To measure phase-noise over a frequency range, the controller 322 utilizes a variable low-noise source command signal 368 via the signal path 338 to change the frequency of the UUT input signal 346 and obtain a phase Commands can be made to the variable low-noise source 310 to measure noise, change the frequency again, measure the phase-noise obtained after the frequency change, and the like.

If the device has at least one controllable variable that affects the phase-noise that the UUT 302 introduces into its output signal, the controller 322 is sent to the UUT 302 via the signal path 324. The command signal 370 is used to control the value of at least one controllable variable.

For example, the UUT 302 may include the amplifier 400 shown in FIG. 4. The amplifier may include a set of variable attenuators shown as first variable attenuator 402 and second variable attenuator 404 for simplicity; However, the variable attenuator set may be two or more variable attenuators. In this embodiment, both the first variable attenuator 402 and the second variable attenuator 404 are variable to vary the amount of additional phase-noise generated in the amplifier output signal (ie, the UUT output signal 348). It can have attenuation which becomes. The first variable attenuator 402 serves to match the amplifier 400 to an input line that transmits the amplifier input signal (ie, the UUT input signal 346), while the second variable attenuator 404 is the amplifier 400. ) Is matched to the output line that transmits the amplifier output signal (ie, UUT output signal 348).

Referring again to FIG. 3, in this embodiment the controller 322 can control the variable attenuators 402, 404 as further described herein. As a result, phase-noise measurements rely on the control of at least one controllable variable in the UUT 302. For example, if the UUT 302 includes an amplifier 400, the phase-noise measured within the UUT 302 in FIG. 4 will depend on the settings of the variable attenuators 402, 404. The controller 322 uses the UUT command signal 370 to change these settings (and the corresponding attenuation provided by the variable attenuators 402, 404) and accordingly phase-noise in the digital ADC output signal 360. Can be observed.

For example, if phase-noise was increased upon command of increased attenuation, controller 322 may repeat the measurement but command a smaller amount of attenuation. It will be appreciated that the interaction and degrees of freedom of variables controlled within the UUT 302 will increase in complexity as the number of variables increases. However, controller 322 may be configured to determine optimal settings of these variables using techniques common to the wireless industry. The method of toggling control for maximum efficiency and noise-floor can be automated because the controller 322 can detect and store the optimized data set. The "best fit" method or any signal processing method that can make a "best data fit" decision can be utilized to determine the optimal settings for the controllable parameters.

As mentioned at the outset, the phase-noise measured at ADC output signal 360 depends on the frequency of UUT input signal 346. For example, UUT 302 may be very noisy at one frequency but less noisy at other frequencies. To measure phase-noise over a frequency range, the controller 322 utilizes a variable low-noise source command signal 368 to command a variable low-noise source 310 to change the frequency of the UUT input signal 346. Can be performed. In response to the change of the controllable variable in the UUT 302, the process of optimizing the phase-noise is then repeated, the resulting phase-noise is recognized, the frequency is changed again, and the like.

In general, the amount and type of variables that can be controlled by the controller 322 in the UUT 302 is virtually infinite. For example, if the UUT 302 includes an optical fiber link, such as the known optical fiber link 200 discussed in FIG. 2, the amplifiers 202, 210 may be configured with attenuators as discussed in connection with FIG. 4, respectively. Four variables can be provided, including the amount of attenuation induced by the attenuator of. In addition (or alternatively), the biasing of transistor (s) in amplifiers 202 and 210 may be made as a controllable variable. Similarly, the biasing of laser diode 204 and photo detector 208 may also be configured as controllable variables. After determining the optimal setting of the controllable variables, a system or device may be fabricated by the manufacturer that is not controllable but is configured as determined using the ANM system 300.

Combined UUT and Variable Low-Noise Source

Those skilled in the art have illustrated that the UUT 302 and the variable low-noise source 310 shown in FIG. 3 are separated from each other. It is to be understood that this is only a conceptual separation since it can be combined. In the example of an oscillator, the variable low-noise source is a UUT. As such, it may be unnecessary to drive this kind of source-UUT with separate low-noise sources, so there is no need for an externally generated UUT input signal sent to the UUT.

Referring now to FIG. 5, a functional block diagram of another embodiment of ANM system 500 is shown. In this embodiment, the ANM system 500 is in signal communication with the source-UUT 502 via signal paths 504, 506, 508, 510.

ANM system 500 includes variable phase shifter 512, delay line 514, variable amplifier 516, mixer 518, variable low-noise matching amplifier 520, ADC 522, and controller 524. It may include. In this embodiment, the variable amplifier 516 may be in signal communication with the delay line 514, mixer 518, and controller 524 through signal paths 526, 528, and 530, respectively. The mixer 518 may also be in signal communication with the variable phase shifter 512 and variable low-noise matching amplifier 520 through signal paths 532 and 534, respectively. The variable phase shifter 512 may also be in signal communication with the source-UUT 502 via the signal path 510. The ADC 522 may be in signal communication with both the variable low-noise matching amplifier 520 and the controller 524 through signal paths 536 and 538, respectively. The controller 524 is configured to provide a source-UUT 502, a variable phase shifter 512, a variable amplifier 516 and a variable low-noise matching amplifier through signal paths 504, 508, 540, 530, 542, respectively. 520 and in signal communication. Again, the controller 524 may be a controller device, a microcontroller, a processor, a microprocessor, an application specific integrated circuit, a DSP or other similar programmable device.

In an operation of analyzing the phase-noise of the source-UUT 502, the delay line 514 can generate a delayed output signal 544 of the source-UUT output signal 546. The source-UUT output signal 546 is generated by the source-UUT 502. Those skilled in the art will appreciate that if the source-UUT 502 is a perfect source capable of generating a sine output signal such as cos (wt), the phase difference between any of the times t1 and t2 only depends on the delay period between these times. Will know. However, there will be some phase-noise that affects this phase difference in real world sources. In general, the choice of delay period can be seen to affect the ability of the ANM system 500 to measure phase-noise at a small frequency offset relative to the frequency of the carrier signal as well as phase-noise measurements. As the delay provided by the delay line 514 is increased, the ability to measure phase-noise at small offsets from the carrier frequency improves with sensitivity. However, the delay cannot be arbitrarily increased because the attenuation through delay line 514 can be so severe that it can affect the measurement.

3 and 5 show the basic similarity between the characteristics of the source-UUT and the non-source UUT. Regardless of whether the UUT is a source-UUT or non-source UUT, variable amplifier 314 or 516, mixer 316 or 518, variable low-noise matching amplifier 318 or 520, ADC 320 or 522 and variable The control and operation of the phase shifter 312 or 512 is the same. Thus, similar to FIG. 3, the source-UUT 502 is a variable phase shifter input signal 548 to the variable phase shifter 512 similar to that discussed with respect to the variable low-noise source 310 of FIG. 3. ). The variable amplifier 516 receives the delayed output signal 544 and generates a variable amplifier signal 550 that is passed to the mixer 518, which combines the variable amplifier signal 550 with the variable phase shifter. Mixer output signal 554 is generated by mixing to variable phase shifted signal 552 (90 degree phase shifted version of variable phase shifter input signal 548) generated by 512. The controller 524 controls the variable amplifier 516 with a variable amplifier control signal 556 to maintain linear operation of the mixer 518. The controller 524 uses a variable phase shifter control signal 558 with a variable phase shifter 512 through a signal path 540, 542, respectively, and a variable low-noise matching amplifier control signal 559 with a variable low. Further control the noise matching amplifier 520. The controller 524 uses the first UUT command signal 560 through the signal path 508 similarly as described with respect to tuning the non-source UUT 302 of FIG. 3 to the source-UUT 502. Operate to tune at least one controllable variable in the. Similarly, the controller 524 controls the carrier frequency used by the source-UUT 502 using the second UUT command signal 562 via the signal path 504 as described in connection with FIG. 3. do. Again similar to the example of FIG. 3, controller 524 generates signal path 538 generated by ADC 522 in response to receiving variable low-noise matching amplifier output signal 566 over signal path 536. Controlling the devices and / or modules of the ANM system 500 in response to monitoring the ADC output signal 564 received via the control.

6 shows a functional block diagram of another embodiment of an ANM system 600 in signal communication with a source-UUT 602 via signal paths 604, 606, 608, 610. Since the amount of delay provided by the delay line greatly affects the characteristics of phase-noise from the source-UUT 602, the ANM system 600 (unlike the ANM system 500 shown in FIG. 5) It includes a selectable delay characteristic shown as an adjustable delay line 612. However, similar to ANM system 500 of FIG. 5, ANM system 600 also has variable phase shifter 614, variable amplifier 616, mixer 618, variable low-noise matching amplifier 620, ADC 622 and controller 624 may be included. In this embodiment, the variable amplifier 616 may be in signal communication with the adjustable delay line 612, mixer 618, and controller 624 through signal paths 626, 628, 644, respectively. The mixer 618 may also be in signal communication with the variable phase shifter 614 and the variable low-noise matching amplifier 620 via signal paths 632 and 6334, respectively. The variable phase shifter 614 is also in signal communication with the source-UUT 602 via the signal path 610. The ADC 622 is in signal communication with the variable low-noise matching amplifier 620 and the controller 624 via signal paths 636 and 638, respectively. The adjustable delay line 612 is also in signal communication with the source-UUT 602 and the controller 624 via signal paths 608, 640, respectively. The controller 624 includes a source-UUT 602, a variable phase shifter 614, a variable amplifier 616, a variable low-noise matching amplifier through signal paths 604, 642, 644, 646, 648, respectively. 620 and in signal communication with the ADC 622. Again, the controller 624 may be a controller device, a microcontroller, a processor, a microprocessor, an application specific integrated circuit, a DSP or other similar programmable device.

The operation of the ANM system 600 is similar to the operation of the ANM system 500 shown in FIG. 5 except for differences in the operation and control of the uncontrolled delay line 514 versus the selectable delay characteristic shown in FIG. Do. In FIG. 6, the selectable delay characteristic is provided by an adjustable delay line 612 that delays the source-UUT output signal 650 received over the signal path 608 and generated by the source-UUT 602. do. Depending on the desired sensitivity and frequency offset in the phase-noise measurement, the controller 624 uses the adjustable delay line control signal 652 transmitted over the signal path 640 to delay lines in the adjustable delay line 612. Select (not shown).

By way of example, the adjustable delay line 612 may include a plurality of delay lines that provide a corresponding plurality of delays from delay τ ₁ to delay τ _N as shown in FIG. 7. For the sake of simplicity, only the first delay line 700 with the delay τ ₁ , the second delay line 702 with the delay τ ₂ , and the Nth delay line 704 with the delay τ _N are 7 is shown from a plurality of delay lines included in the adjustable delay line 612. These delay lines 700, 702, 704 may be selected by the controller 324 by an adjustable delay line control signal 653 through the operation of a switch such as, for example, field effect transistors (“FETs”) (not shown). Can be.

6 again, in the operation example, regardless of which delay line is selected within the adjustable delay line 612, the variable amplifier 616 takes the signal path 626 from the adjustable delay line 612. Amplify the received delayed output signal 654 to produce a variable amplifier output signal 656 that is passed to the mixer 618 similar to the process described with respect to FIG. 5. Similarly, the variable phase shifter input signal 658 is generated by the source-UUT 602 and passed through the signal path 610 to the variable phase shifter 614. The variable phase shifter 614 receives the variable phase shifter input signal 658 and phase shifts 90 degrees to produce a variable phase shifted signal 660 that is also received by the mixer 618. In this regard, the mixer 618 is passed through the signal path 634 to the variable low-noise matching amplifier 620 by mixing the variable amplifier output signal 656 with the variable phase shifted signal 660. Generate an output signal 662.

In this embodiment, the source-UUT 602 may be driven by at least one controllable variable, such as, for example, transistor bias voltage, or by the source-UUT control signal 664 of the controller 624 transmitted over the signal path 604. Include other controllable variables. Thus, the ANM system 600 operates similarly to the description of the operation of the ANM system 500 of FIG. 5. In response to the phase-noise of the source-UUT 602 using the digital ADC output signal 666 (via the signal path 648) generated by the ADC 622, the controller 624 causes the second source- Drive UUT control signal 668 (via signal path 606) to tune at least one controllable variable in source-UUT 602 to optimize the phase-noise performance of ANM system 600.

While the invention has been described in terms of specific embodiments, it will be appreciated that the description is merely an application of the invention and should not be treated as a limitation. For example, in FIGS. 5 and 6, the order of the variable amplifier 514 or 612 and the delay line 514 or the adjustable delay line 612 is that the delay line (or adjustable delay line) is delayed in the variable amplifier output signal. It may be reversed to provide a version to the mixer 518 or 618.

Wireless Phase-Noise Measurement

In another embodiment, the ANM system may measure the noise performance of the wireless transmitter through the reception and analysis of the transmitted wireless signal. For carrying out this embodiment, the user applies the direct down-conversion technique described in US Pat. No. 6,745,020 (patent entitled “Direct Downconversion Receiver”, filed Aug. 29, 2002, incorporated herein by reference). May enable telemetry of the noise-floor of the wireless transmitter.

Referring to FIG. 8, a functional block diagram of an embodiment of a direct down-conversion receiver 800 (described in US Pat. No. 6,745,020) is shown. The direct down-conversion receiver 800 may include a signal source 802, a variable phase shifter 804, a mixer 806, a first coupler 808, a second coupler 810, and an antenna 812. Can be. In this embodiment, the first coupler 808 can be in signal communication with both the signal source 802 and the second coupler 810 via signal paths 814 and 816, respectively. The second coupler 810 may be in signal communication with both mixer 806 and antenna 812 via signal paths 818 and 820, respectively. The variable phase shifter 804 may be in signal communication with the first coupler 808 and mixer 806 through signal paths 822, 824, respectively.

The direct down-conversion receiver 800 can mitigate the DC offset component of the baseband signal output 826 from the mixer 806. The mixer 810 has its baseband signal output 826 having a frequency at which mixer input signal 828 (via signal path 818) to mixer 806 and mixer 806 via signal path 824. It may be a double-sideband or single-sideband mixer that represents the product of the reference signal 830 (such as a local oscillator (“LO”)).

In operation, the direct down-conversion receiver 800 receives an input radio frequency (“RF”) signal 832 with a given carrier frequency at antenna 812. In this regard, antenna 812 generates a received RF signal 834 that is passed through signal path 820 to second coupler 810. The second coupler 810 couples the received RF signal 834 to the mixer input signal 828 and then passes it to the mixer 806. Those skilled in the art will appreciate that the input RF signal 832 is received by the antenna 812 and then processed by a filter and a low-noise amplifier (not shown) before being coupled to the mixer 806 as is typically practiced in receiver configurations. Will know.

Signal source 802 may be a voltage controlled oscillator (“VCO”) or other similar generator of appropriate LO signal output. For example, the signal source 802 can generate a sinusoidal output signal 836. Those skilled in the art will appreciate that this kind of LO signal output will depend on the particular modulation present in the input RF signal 832. Before the mixer input signal 828 is sent to the mixer 806, the first coupler main output signal 838 and the received RF signal 834 are coupled through the second coupler 810. The first coupler main output signal 838 is generated by passing a sine output signal 836 through the coupler 808. The first coupler 808 also generates a variable phase shifter input signal 840 that is passed through the signal path 822 to the variable phase shifter 804. It will be appreciated that a signal splitter may also be used instead of the first coupler 808. The variable phase shifter 804 receives a frequency reference signal 830 through the LO input port 842 (ie, signal path 824) of the mixer 806 by phase shifting the sine output signal 836 by 90 degrees. To a quadrature phase local oscillator signal 806 (i.e., frequency reference signal 830). While the variable phase shifter 804 is shown as a functional block of discrete components in this embodiment, it will be appreciated that the variable phase shifter 804 may also be included in the mixer 806. In the example of the variable phase shifter 804 being part of the mixer 806, the mixer 806 is connected via the RF input port 844 (ie, the signal path 818) of the mixer 806 with respect to the frequency reference signal 830. Coupled to the input receiving port 828) to minimize the possibility of self-mixing and producing an undesirable DC offset component at the baseband signal output 826. Alternatively, the variable phase shifter 804 may be coupled to the RF input port 844 so as to minimize the radial or reactive coupling of the frequency reference signal 830 to the mixer 806. It may be placed as close as possible to the LO input port 842.

The configuration of the first and second couplers 808 and 810 may be a type of waveguide (e.g., stripline (e.g., stripline) stripline, coaxial cable or microstrip type waveguide). In other words, the type of waveguide implemented depends on the carrier frequency, signal power level, space concerns, and other known configuration choices.

Baseband signal output 826 in this embodiment includes a frequency offset item that is a sinusoidal curve of the frequency difference between the received RF signal 834 and the RF frequency for sine output signal 836. In addition, various dual frequency items are generated, which can be easily filtered out. In this way, baseband signal output 826 can be obtained without the problem of DC offset present in other direct down-conversion receivers.

Referring to FIG. 9, an ANM system 900 is provided that includes the application of the direct down-conversion receiver configuration described in connection with FIG. 8. The ANM system 900 includes a source LO 902, variable amplifier 904, variable phase shifter 906, mixer 908, variable low-noise matching amplifier 910, ADC 912, controller 914. ), Antenna (s) 916, low-noise amplifier (“LNA”) 918, coupler 920, and optional isolator 922. The variable amplifier 904 may be in signal communication with the mixer 908, the controller 914, and the coupler 920 via signal paths 924, 926, 928, respectively. The mixer 908 can be in signal communication with the variable phase shifter 906 and the variable low-noise matching amplifier 910 via signal paths 930 and 932. The variable low-noise matching amplifier 910 may be in signal communication with the ADC 912 and the controller 914 through signal paths 934 and 936, respectively. The controller 914 may be in signal communication with the variable phase shifter 906 and the ADC 912 via signal paths 938 and 940, respectively. The source LO 902 may be in signal communication with the variable phase shifter 906 and optional isolator 922 via signal paths 942 and 944, respectively. The coupler 920 may be in signal communication with the LNA 918 and optional isolator 922 via signal paths 946 and 948, respectively. The LNA 918 may be in signal communication with the antenna 916 via a signal path 930.

In an example of operation, a wireless signal 950 is received at the antenna 916 and processed through the LNA 918 to produce an RF received signal 952. In this embodiment, the source LO 902 is configured similar to that described with respect to FIG. 5. The source LO 902 generates two identical signals, referred to as the source LO output signal 954 and the variable phase shifter input signal 956. However, in this embodiment the source LO output signal 954 or variable phase shifter input signal 956 acts as a LO on the received RF signal 952. As such, the variable phase shifter input signal 956 is phase shifted via the variable phase shifter 906 and is received in the quadrature phase LO signal received at the LO port 962 of the mixer 908 via the signal path 930. (I.e., variable phase shifted output signal 960). The received RF signal 952 is coupled to the source LO output signal 954 using a coupler 920 to form a coupled coupler output signal 964. In this example, the source LO output signal 954 is processed by an optional isolator 922 to form an isolated source LO signal output signal 966. Without using the optional isolator 922, the isolated source LO output signal 966 is the same as the source LO output signal 954.

As described in connection with FIG. 5, the variable amplifier 904 has a variable amplified output signal 968 obtained that is applied to the RF port 970 of the mixer 908 so that the mixer 908 operates linearly. The amplified coupler output signal 964 can be amplified to have sufficient power compared to the power of the port 962. In this regard, depending on the configuration and linearity of the mixer 908, the variable amplifier 904 may not be necessary in some implementations. Mixer output signal 972 from mixer 908 on signal path 932 may be processed as mentioned at the outset with respect to FIG. 5. In this implementation, the controller 914 shifts the variable phase as previously mentioned using the control signal 974 through the signal path 936, the signal paths 926, 978, and the signal paths 938, 976, respectively. 906, control the variable amplifier 904 and variable low-noise matching amplifier 910. To prevent the received RF signal 952 from disturbing the quadrature LO signal (ie, the source LO output signal 954), the source LO 902 uses a selective isolator 922 to coupler 920. Can be separated from). However, if coupler 920 is sufficiently directional, optional isolator 922 may not be needed. By performing spectral analysis, such as FET, on the digitized noise signal 980 generated by the ADC 912 on the signal path 940, the controller 914 generates a wireless signal 950 (not shown). Noise floor can be characterized. In this example, the noise measurement is done entirely wireless without the need for any transmission line connecting the ANM system 900 to the wireless transmitter.

In other embodiments, delay lines (not shown) may be used similarly as described with respect to FIG. 5. This delay line may be inserted before and after the coupler 920. However, the quadrature phase discriminator function for the received RF signal 952 will be applied to the source LO output signal 954 and the variable phase shifter input signal 956 instead of the received RF signal 952. In this respect it is not achieved through the inclusion of such a delay line. However, there may be embodiments that are advantageous for performing the discrimination function on the source LO 902. In any case, the source LO 902 should have a lower noise-floor than the radio transmitter (not shown) used to transmit the radio signal 916 to make further noise measurements. An example low-noise source that can be configured as source LO 902 is described in US patent application Ser. No. 11 / 494,884, filed Jul. 21, 2006, the contents of which are incorporated herein by reference. However, it will be appreciated that any suitable low-noise source may be used to form the source LO 902. An initial correction may be made to determine the noise-floor of the source LO 902 so that additional noise resulting from coupling with the received RF signal 952 may be determined. Since the down-conversion configuration is formed instead of the quadrature phase discriminator with respect to the received RF signal 952, the wireless signal 950 will be represented as a frequency difference (offset from LO) within the mixer output signal 972. The transmitter noise spectrum will be between this signal and DC. As described in connection with FIG. 8, an appropriate signal filter (not shown) may be used to remove the dual frequency component. Unlike the discriminator described in FIG. 5, this spectrum of the digitized noise signal 980 will represent amplitude-noise and phase-noise.

As a result, FIG. 10 shows a functional block diagram of an embodiment of ANM system 1000 capable of measuring amplitude-noise and phase-noise of wireless transmitter 1002, respectively. The ANM system 1000 includes a source LO 1004, a variable amplifier 1006, a first variable phase shifter 1008, a second variable phase shifter 1010, a first mixer 1012, a controller 1014. , First coupler 1016, second coupler 1018, antenna 1020, first LNA 1022, amplitude and phase (“AM & PM”) modulator 1024, optional isolator 1026, first It may include a second mixer 1028, a splitter 1030, a second LNA 1032, a third LNA 1034, a first ADC 1036, and a second ADC 1038. The source LO 1004 has a first variable phase shifter 1008, a second variable phase shifter 1010, a controller 1014 and an optional isolator through each of the signal paths 1040, 1042, 1044. 1026). The splitter 1030 transmits a predetermined signal generated by the source LO 1004 (and transmitted via the signal path 1040) through each of the signal paths 1045 and 1046 to the first variable phase shifter 1008. And a signal splitter capable of dividing into two divided signals sent to the second variable phase shifter 1010. The variable amplifier 1006 may Shinto communicate with the first coupler 1016, the second coupler 1018, and the controller 1014 through signal paths 1048, 1050, and 1052, respectively. The second coupler 1018 can be in signal communication with the first mixer 1012 and the second mixer 1028 through signal paths 1053, 1054, respectively. The first variable phase shifter 1008 may be in signal communication with the first mixer 1012 and the controller 1014 via signal paths 1056 and 1058, respectively. Similarly, second variable phase shifter 1010 may be in signal communication with second mixer 1028 and controller 1014 via signal paths 1060 and 1062, respectively. First coupler 1016 may be in signal communication with optional isolator 1026 and AM & PM modulator 1024 via signal paths 1064 and 1066, respectively. The first LNA 1022 may be in signal communication with the antenna 1020 and the AM & PM modulator 1024 via signal paths 1067 and 1068, respectively. The wireless transmitter 1002 may be in signal communication with the antenna 1020 via a signal path 1069, which is a wireless signal path in this embodiment. The controller 1014 may also be in signal communication with the transmitter 1002 via a signal path 1070, which is also a wireless signal path.

In an example of operation, the wireless transmitter 1002 receives an antenna signal 1071 to antenna (s) 1020 that generates a received signal 1072 that is passed through the signal path 1067 to the LNA 1022. Send a wireless signal 1071. The LNA 1022 processes the received signal 1072 to generate a received RF signal 1073 that is transmitted to the AM & PM modulator 1024 over the signal path 1068. The AM & PM modulator 1024 generates the modulated received signal 1074 by receiving the received RF signal 1073 and applying AM modulation and / or phase PM techniques to the received RF signal 1073. This modulated received signal is passed through a signal path 1066 to the first coupler 1016.

The source LO 1004 generates a source LO output signal 1075 that is passed through the signal paths 1044 and 1064 through the optional isolator 1026 to the first coupler 1016. If an optional isolator 1026 is present, the optional isolator 1026 generates a separate source LO output signal 1076. If the optional isolator 1026 is not present, the isolated source LO output signal 1076 will be the same as the source LO output signal 1075. As illustrated again in FIG. 9, the purpose of the optional isolator 1026 is to ensure that the modulated receive signal 1074 does not disturb the source LO output signal 1075, so that the optional isolator 1026 is configured to provide a source LO. 1004 is separated from the first coupler 1016. However, if the first coupler 1016 is sufficiently oriented, the optional isolator 1026 may not be necessary.

The first coupler 1016 combines a separate source LO output signal (or source LO output signal 1075 in the absence of an optional isolator 1026) and the modulated received signal 1074 to form a signal path 1048. To generate a combined received signal 1077 that is passed through the variable amplifier 1006. The variable amplifier 1006 receives the combined received signal 1077 and amplifies it to produce an amplified received signal 1078 that is passed through the signal path 1050 to the second coupler 1018. The second coupler 1018 receives the amplified received signal 1078 and the first amplified received signal 1079 and the second mixer 1028 that pass through the first mixer 1012 (via the signal path 1053). ) Into a second amplified received signal 1080 that is passed (via signal path 1054).

The source LO 1004 generates a variable phase shifter input signal 1081 that is transmitted via signal path 1040 to splitter 1030 (or another suitable device capable of dividing the signal into two signals). The splitter 1030 receives the variable phase shifter input signal 1081 and divides it so that the first variable phase shifter input signal 1082 and the second variable phase shifter along the signal paths 1045 and 1046 respectively. Generate an input signal 1083. The first variable phase shifter input signal 1082 is processed in a first variable phase shifter 1008 to provide a quadrature phase LO signal 1084 to the first mixer 1012 in a manner similar to that described with respect to FIG. To provide. The second variable phase shifter input signal 1083 is also processed in the second variable phase shifter 1010 to provide an in-phase LO signal 1085 to the second mixer 1028. As discussed above, both mixers 1012 and 1028 receive a segmented version of the amplified received signal 1078.

Since the second mixer 1028 mixes the in-phase LO signal 1085 with the second amplified received signal 1080, the resulting second mixer output signal 1086 is similar to that described with respect to FIG. Figure 2 shows an in-phase ("I") component after processing the second mixer output signal 1086 through a second LNA 1032 and a first ADC 1036. The I component signal 1087 thus obtained may be passed to the controller 1014 via the signal path 1088. Similarly, first mixer 1012 mixes first amplified received signal 1079 and quadrature phase LO signal 1084 to produce first mixer output signal 1089. The first mixer output signal 1089, once processed by the third LNA 1034 and the second ADC 1038, is passed through the signal path 1091 to the controller 1014 by a digital quadrature phase (“Q”). Component signal 1090 is shown.

In this embodiment, the determination of whether amplitude-noise or phase-noise is measured with these I and Q component signals 1087, 1090 is the modulation utilized on the received RF signal 1073 at the AM & PM modulator 1024. It is controlled by the type of. By way of example, when the AM & PM modulator 1024 applies a predetermined AM modulation followed by a predetermined PM modulation to the received RF signal 1073, the ANM system 1000 uses the I and Q component signals 1090, 1087. The effect of these modulations on C) can be analyzed to determine additional amplitude-noise and amount of additional phase-noise at wireless transmitter 1002.

In addition to determining the noise-floor at the wireless transmitter 1002, the ANM system 1000 may tune the wireless transmitter 1002 using the techniques described in US patent application 11 / 134,546. For example, the wireless transmitter 1002 can be biased by a transistor (not shown) or other suitable tunable device that can be controlled by the controller 1014 using the wireless transmitter control signal 1091 via the signal path 1070. Can have settings. By observing the effect on the noise-spectrum (or its combined effect as described in connection with FIG. 9) through the analysis of the I and Q component signals 1090 and 1087, the controller 1014 is configured to transmit a radio transmitter ( 1002 may be tuned to adjust the radio transmitter control signal 1091 (which may be designated as a control variable (“CV”)) to lower the noise-floor of the radio transmitter 1002. The controller 1014 analyzes the digitized noise signal (s) and adjusts corresponding control variables (“CVs”) accordingly (the first variable phase shift, which is another CV through the signal path 1058). Control the first variable phase shifter 1008 through the self control signal 1092 to maintain its output in quadrature, and the second variable phase shifter control signal (another CV via the signal path 1062). Controlling the second variable phase shifter 1010 (via 1093) maintains its output in phase with the source LO output signal 1075.

For example, the first variable phase shifter 1008 is in orthogonal phase with the signal 1078 (basically an associated version of the source LO output signal 1075) amplified from the variable low-noise amplifier 1006. The first variable phase shifter control signal 1092 is controlled. However, if the carrier signal is not canceled due to an error in the first variable phase shifter control signal 1092, the carrier signal will be present as a DC offset in the resulting noise-spectrum. This DC offset is scanned by the first variable phase shifter 1008 at an angle of nearly orthogonal phase (eg, 80-100 degrees) such that one side of the quadrature has one polarity and the other side of the orthogonal phase has opposite polarity. Will change the sign as The effect of CV on measurable variables ("MV"), such as DC offset, can be used in various control algorithms. For example, a change in the sign of the DC offset can be used for zero-crossing search. In general, the CV that produces the zero-crossing MV may have a range divided into a plurality of intervals. The controller 1014 steps the CV through these intervals and observes the effect on the zero-crossing MV. For example, the zero-crossing MV may change the sign in relation to two values MV _o , MV ₁ corresponding to the values for CV ₀ and CV ₁ . Given this straddling of zero-crossing points, the optimal setting for CV ("CV _optimal ") is expressed as follows:

Equation (1)

In general, the CV _optimal , such as the first variable phase shifter control signal 1092, will change over time due to temperature changes and other effects. This change in time can be tracked using a convergence algorithm. For example, the straddling interval (corresponding to CV ₀ and CV ₁ ) may be reduced by a convergence factor such as two. Then, for example, using Equation (1) to calculate a new value for CV _optimal . The difference between successive measurements can be averaged to the difference obtained previously to provide a time-varying-corrective factor. The calculated CV _optimal can then be adjusted according to the time-varying-correction factor. Prior to updating the CV, a measurement of the MV for the tolerance factor may be performed.

The second variable phase shifter 1010 is controlled in a complementary fashion in that the output signal of the second mixer 1028 should have a maximum carrier power. The second variable phase shifter control signal 1093 (also another CV) that controls the second variable phase shifter 1010 is controlled by the controller 1014 to find the maximum value in the corresponding MV. For example, the output signal of the second mixer 1028 has a corresponding value of MV in five consecutive increments in the CV range of values CV ₀ to CV ₄ , where MV ₂ > MV ₁ > MV ₀ and MV ₂ > MV. It can be incremented over that range to form ₃ > MV ₄ . The value of CV that produces the maximum MV will be CV ₂ . If no maximum value is found over the effective range of the CV, the spacing between successive CVs may be too wide for a new search to be made at intervals reduced by one half. If the maximum is found, the spacing between CV ₀ and CV ₄ is sampled at twice the previous rate such that the spacing between successive CV points is half that used in conventional searches. If the maximum pattern is no longer discernible, the algorithm can assume that it is zoomed into noise at the maximum of the MV variable. The maximum value of MV for the end-of-life repetition of a CV whose pattern is not discernible gives the corresponding CV _optimal value. This CV _optimal value can be tracked as discussed in connection with zero-crossing MV control.

Further control is performed via the control signal 1094 along the signal path 1042 (if the frequency variation is desired), the variable amplifier 1006 (control signal 1095 along the signal path 1052). And LNAs 1032, 1034 (which may be variable LNAs). Empirical observations in the ANM system 1000 may show that a single CV may have a predominant effect on other CVs. For example, the first variable phase shifter control signal 1092 may be a CV prevailing over the zero-crossing point in the first mixer output signal 1089. Thus, the predominant CV in this example may be tuned first, followed by the less predominant CVs. However, it will be appreciated that this tuning methodology can be easily extended with parallel control of multiple CVs.

Referring to FIG. 11, a flow diagram illustrating an example of a method of operation of an ANM system utilizing feedback that tunes one or more controllable variables in a UUT is illustrated. An exemplary process 1100 begins at step 1102, and the variable low-noise source generates an RF signal and transmits the RF signal to a UUT and a variable phase shifter at step 1104. In step 1106, the UUT receives an RF signal and processes it to produce an output signal, which is then passed to a variable amplifier, and in step 1108 the variable amplifier amplifies the output signal to produce a variable amplified signal. In step 1110, the variable phase shifter receives the RF signal and phase shifts the RF signal by 90 degrees to generate a variable phase shifted signal.

In step 1112, the variable amplified signal and the variable phase shifted signal are then passed to the mixer 1112. In step 1114, the mixer generates a mixer output signal that is sent to a variable low-noise matching amplifier, and in step 1116 the variable low-noise matching amplifier is passed to an analog-to-digital converter (ADC). Generate the output signal. The ADC then generates a digital ADC output signal that is passed to the controller in step 1118. In step 1120, the controller monitors the digital ADC output signal and utilizes a variable phase shifter control signal, a variable amplifier control signal and a variable low-noise matching amplifier control signal, respectively, to vary the phase shifter, variable amplifier and variable low. Control the noise matching amplifier and command the variable low-noise source to change the frequency of the UUT input signal utilizing the variable low-noise source command signal.

The controller is also configured to utilize the UUT command signal to control the value of at least one controllable variable of the UUT. As described in connection with FIG. 4, an example of such controllable variable may be a variable attenuator included in the UUT. At a decision step 1122, the controller determines whether an optimal setting has been made for the at least one controllable variable, such as if there is a minimum phase-noise. Once the optimal setting has been obtained, the process 1100 ends at step 1124. Otherwise, the process returns to step 1104 and repeats the process.

Those skilled in the art will appreciate that one or more of the functions, modules, units, blocks, processes, sub-processes or process steps described above may be performed by hardware and / or software. When the process is performed by software, the software can reside in the software memory (not shown) of the above-described predetermined device. Software in software memory is the execution of logical functions (ie, "logic", which may be performed in digital form, such as digital circuits or source code, or in analog form, such as analog sources or analog sources, such as analogue electrical sound or video signals). An instruction execution system, such as a computer-based system, a processor-mounted system, or other system, which may include an ordered list of executable instructions for a computer, and may selectively retrieve instructions from an instruction execution system, apparatus, or apparatus and execute the instructions; It may optionally be embedded in any computer readable (or signal bearing) medium used by or in connection with the device or instrument. In this document, "computer-readable media" and / or "signal-bearing media" may include, store, communicate, propagate, or transport a program used by or in connection with an instruction execution system, apparatus, or apparatus. It is a predetermined means. The computer readable medium may be, but is not limited to, electronic, magnetic, optical, electromagnetic, infrared, or semiconductor systems, devices, instruments, or propagation media, optionally. Although not an exhaustive list of computer readable media, more specific examples may include: electrical connections (electronic) including one or more wires, portable computer diskettes (magnetic), RAM (electronic), read-only memory "ROM" (electronic), erasable programmable read-only memory (EPROM or flash memory) (electronic), optical fiber (optical), and portable computer disk read-only memory "CDROM" (optical). The computer readable medium may even be paper or other suitable medium in which the program is printed, compiled, interpreted or otherwise appropriately suited as needed, such as by optically scanning a paper or other medium when electronically capturing the program. After being processed as, it may be a medium stored in the computer memory.

While various embodiments of the invention have been described, those skilled in the art will recognize that many more embodiments and implementations are possible within the scope of the invention. Accordingly, the invention is not to be restricted except in light of the attached claims and their equivalents.

Claims (12)

A noise test measurement system adapted to measure the noise component of a transmitted RF signal: An antenna for receiving the transmitted RF signal and providing a received RF signal; A low-noise amplifier configured to amplify the received RF signal to provide an amplified RF signal; A local oscillator adapted to provide an LO signal; A first coupler configured to couple a first version of the LO signal to the amplified RF signal to provide a combined RF signal; A first variable phase shifter configured to phase shift the second version of the LO signal into a quadrature phase LO signal in response to a control signal; An RF port adapted to receive the combined RF signal and an LO port adapted to receive the quadrature LO signal, and further configured to mix signals received at the RF and LO ports to provide a first mixed output signal A first mixer; A processor configured to analyze the digitized version of the first mixed output signal to generate the control signal and measure the noise component Noise test measurement system comprising a. The noise test measurement system of claim 1, further comprising an analog-to-digital converter providing the digitized version. 3. The noise test measurement system of claim 2 further comprising a second variable amplifier configured to variably amplify the mixer output to provide a signal to the analog-to-digital converter. 4. The noise test measurement system of claim 3, wherein the processor is further configured to control the second variable amplifier to maintain its output within a desired dynamic range of the analog-to-digital converter. The method of claim 1, A second variable phase shifter configured to phase shift a third version of the LO signal in response to a second control signal to provide an in-phase LO signal; An RF port adapted to receive the combined RF signal and an LO port adapted to receive the quadrature LO signal, and further configured to mix signals received at the RF and LO ports to provide a second mixed output signal A second mixer; The processor is further configured to analyze the digitized version of the second mixed output signal to generate the control signal and to measure the noise component. 6. The noise test measurement system of claim 5 wherein the processor is adapted to determine amplitude noise measurements and phase noise measurements from the noise components. 2. The noise test of claim 1, wherein the wireless transmitter providing the transmitted RF signal includes tunable parameters, and wherein the processor is configured to tune the tunable parameters to reduce the measured noise component. Measuring system. 8. The noise test measurement system of claim 7, wherein the tunable parameter is a bias setting of a transistor. As a method of measuring the noise floor of a wireless transmitter: Transmitting an RF signal from the wireless transmitter to provide a transmitted RF signal; Receiving the transmitted RF signal and providing a received RF signal; Combining the received RF signal with a predetermined version of the LO signal to provide a combined RF signal; Mixing the combined RF signal with a quadrature version of the LO signal to provide a mixer output signal; Analyzing the spectrum of the mixer output signal to determine the noise floor Noise floor measurement method comprising a. 10. The method of claim 9, further comprising adjusting the power of the combined RF signal to ensure that the mixing is performed linearly. 10. The method of claim 9, wherein the spectrum analysis step is: Digitizing the mixer output signal to form a digitized signal; Fast Fourier transforming the digitized signal. 12. The method of claim 11, further comprising controlling the phase shift based on the spectral analysis.

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