JP2010517016A - Automatic noise measurement system - Google Patents

Abstract

A noise test measurement system configured to measure a noise component of a transmitted RF signal is described. The noise test measurement system may include an antenna, a low noise amplifier, a local oscillator, a first combiner, a first variable phase shifter, a first mixer, and a processor. The system may further comprise an analog-to-digital converter and a second variable amplifier configured to provide a signal to the analog-to-digital converter. The processor may be further configured to control the second variable amplifier to maintain its output within the desired dynamic range of the analog to digital converter.

Description

（関連出願の相互参照）
本願は、２００５年５月２０日に出願された「Ｉｎｔｅｌｌｉｇｅｎｔ Ｌｏｗ Ｎｏｉｓｅ Ｄｅｓｉｇｎ」という名称の米国特許出願第１１／１３４，５４６号の一部継続出願（ＣＩＰ）である。本願は、また、２００６年７月２８日に出願された「Ｌｏｗ−Ｎｏｉｓｅ Ｓｏｕｒｃｅ」という名称の米国特許出願第１１／４９４，８８４号のＣＩＰであり、この米国特許出願第１１／４９４，８８４号自体は、米国特許出願第１１／１３４，５４６号のＣＩＰであり、これらの出願の両方が本明細書により、それら全体が参照により援用される。本願は、また、２００７年１月１７日に出願された「Ｗｉｒｅｌｅｓｓ Ｎｏｉｓｅ Ｍｅａｓｕｒｅｍｅｎｔ」という名称の米国特許出願第６０／８８５，２１８号に対する優先権を主張し、この出願は本明細書により、その全体が参照により援用される。

(Cross-reference of related applications)
This application is a continuation-in-part (CIP) of US patent application Ser. No. 11 / 134,546, filed May 20, 2005, entitled “Intelligent Low Noise Design”. This application is also the CIP of US patent application Ser. No. 11 / 494,884 entitled “Low-Noise Source” filed on Jul. 28, 2006, which is hereby incorporated by reference. As such, it is the CIP of US patent application Ser. No. 11 / 134,546, both of which are hereby incorporated by reference in their entirety. This application also claims priority to US Patent Application No. 60 / 885,218 entitled “Wireless Noise Measurement” filed on January 17, 2007, which is hereby incorporated by reference in its entirety. Is incorporated by reference.

(Background of the Invention)
(1. Field of the Invention)
The present invention relates to noise measurement, and more particularly to an automatic noise measurement system for a wireless transmitter.

(2. Related technology)
In general, the purpose of electrical, electro-optic, and electroacoustic communication systems is to transmit information bearing signals (commonly known as “baseband signals”) over a communication channel that separates the transmitter from the receiver. Baseband signals (also known as baseband waves, baseband waveforms, or simply “basebands”) are used to specify the frequency band that represents the original information carrier signal when delivered by the information source.

  Efficient use of a communication channel typically shifts the frequency range of the baseband signal to another frequency range suitable for transmission over the communication channel and then returns to the original frequency range after reception. A corresponding transition is necessary. As an example, a typical wireless system generally must operate with signals having a frequency of 30 kHz or higher, while baseband signals are typically in the audible frequency range (ie, the 20-20 kHz frequency range). In order for a wireless system to operate satisfactorily, including some frequencies, some form of frequency band transition must be used. The transition of a frequency range in a signal is a process that changes some characteristics of a carrier signal (also known as a carrier wave, carrier wavelength, or simply “carrier”) according to a modulated signal (also known as a modulated wave or modulated waveform). Can be achieved by using a process of “modulation” defined as The carrier signal can be modulated or applied in characteristics (usually electrical), typically by a second signal that is an information carrier signal (ie, a modulated signal). It is a signal having a continuous waveform. In general, the carrier signal itself does not convey any information until it is changed in several ways, such as its amplitude, its frequency, or its phase is changed by a modulation signal. The changes induced by these modulation signals convey information via signals resulting from the modulation process. In this situation, the baseband signal is referred to as the modulated signal and the result of the modulation process is referred to as the modulated signal (or modulated wave). At the receiving end in a communication system, it is usually necessary to restore the modulated signal (ie, the original baseband signal, which is also the original information carrier signal) so as to receive the information carried by the modulated signal. This is accomplished by using a process known as demodulation as opposed to the modulation process.

  Unfortunately, noise in electrical, electro-optic, and electro-acoustic systems can interfere with both the amplitude and / or phase of the signals utilized and / or processed by these types of systems. In this context, the term “noise” is used to designate unwanted signals that tend to disturb the transmission and processing of desired signals in electrical, electro-optic, and electroacoustic systems. However, since many of these types of systems are relatively unaffected by amplitude variations, phase variations (denoted “phase noise”) generally involve more problems. As an example, an oscillator is a device capable of generating an oscillating output signal having an oscillating frequency (eg, a signal having a sine waveform, generally known as a “sine signal”). In general, oscillators limit certain types of amplitudes, reducing the effects of any possible fluctuations in amplitude, such that phase noise is typically a major noise factor on the resulting oscillator output signal. Includes features.

  Since noise, such as phase noise, is a very important factor in the design and / or performance of electrical, electro-optic, and / or electroacoustic systems, designers typically do for a given system Requires a measure of phase noise. In the past, various approaches have been utilized to characterize the phase noise of a given system. For example, an amplifier is characterized by first injecting an input signal of a known frequency into the input of the amplifier and then measuring the resulting amplified output signal with a spectrum analyzer, the spectrum analyzer being A device that can display the spectral waveform composition of several electrical, acoustic, or optical signals, including the amplitude intensity and frequency of each of the given signals. Unfortunately, the measurement sensitivity of this approach is limited by the relatively poor sensitivity of the spectrum analyzer. Furthermore, it is difficult to measure phase noise at a frequency value close to the carrier frequency of the signal, and the carrier frequency is the frequency of the carrier signal.

  Unlike spectrum analyzers, phase locked discriminator systems have relatively good sensitivity and generally allow measurements near the carrier frequency. However, configuring a phase locked discriminator system is cumbersome and time consuming. In an attempt to solve this problem, as described in U.S. Patent No. 6,099,028 (incorporated by reference in its entirety), which seeks to alleviate the cumbersome nature of such a system, An automated phase-locked discriminator noise-test measurement system ("APD system") has been developed. FIG. 1 illustrates an example implementation of an APD system 100 that is in signal communication with a unit-under-test (“UUT”) 102 and a spectrum analyzer 104 via signal paths 106, 108, and 110, respectively. A functional block diagram of is shown.

  The APD system 100 includes a variable low noise source 112, a variable phase shifter 114, a variable amplifier 116, a mixer 118, a variable low noise matching amplifier 120, an analog-to-digital converter (“ADC”) 122, Controller 124. UUT 102 is in signal communication with both variable low noise source 112 and variable amplifier 116 via signal paths 106 and 108, respectively. Mixer 118 is in signal communication with variable phase shifter 114, variable amplifier 116, and variable low noise matching amplifier 120 via signal paths 126, 128, and 130, respectively. Variable phase shifter 114 is also in signal communication with variable low noise source 112 via signal path 132. The ADC 122 is in signal communication with the variable low noise matching amplifier 120 and the controller 124 via signal paths 134 and 136, respectively. Controller 124 includes spectrum analyzer 104, variable low noise source 112, variable phase shifter 114, variable amplifier 116, and variable low noise matching amplifier 120 via signal paths 110, 138, 140, 142, and 144, respectively. Signal communication with.

  As an example of operation, the variable low noise source 112 generates a UUT input signal 146 (which is a low noise carrier signal) for driving the UUT 102. UUT 102 may be any device where a user requests a phase noise test measurement, such as an amplifier, phase shifter, diplexer, or other suitable device or system of devices. UUT 102 receives UUT input signal 146 via signal path 106 and processes it to generate UUT output signal 148. As an example, if UUT 102 is an amplifier, UUT output signal 148 is an amplified version of UUT input signal 146. The UUT output signal 148 is received via signal path 108 and amplified by variable amplifier 116 to produce variable amplifier signal 150 that is passed to mixer 118 via signal path 128. Variable low noise source 112 also generates a variable phase shifter input signal 152 that is passed to variable phase shifter 114 via signal path 132. Variable phase shifter input signal 152 is identical to UUT input signal 146 and has the same frequency as UUT input signal 146. The variable phase shifter 114 shifts the variable phase shifter input signal 152 by 90 degrees to generate a variable phase shift signal 154 that is passed to the mixer 118 via the signal path 126. In this manner, the carrier signal of the variable amplifier signal 150 is removed from the mixer output signal 156 generated by the mixer 118. In order to keep the mixer output signal 156 within the proper dynamic range of the ADC 122, the mixer output signal 156 is passed through the signal path 130 to the variable low noise matched amplifier 120 and processed by the amplifier to route the signal path 134. Through which the variable low noise matched output signal 158 is generated.

  In order to remove the carrier signal of the variable amplifier signal 150, the variable phase shift signal 154 must be orthogonal (ie, 90 degrees shifted) to the carrier signal. If the quadrature relationship between the variable amplifier signal 150 and the variable phase shift signal 154 is not established, there will be a DC offset in the digital ADC output signal 160 from the ADC 122.

  The controller 124 monitors the ADC output signal 160 and controls the variable phase shifter 114 using the variable phase shifter control signal 162 to maintain the quadrature relationship between the variable amplifier signal 150 and the variable phase shift signal 154. . Variable phase shifter control signal 162 is transmitted to variable phase shifter 114 via signal path 140.

  Removal of the carrier signal from the variable amplifier signal 150 is also equal when the carrier signal (ie, UUT input signal 146) and the phase shift version of the carrier (ie, variable phase shift signal 154) enter the mixer 118. It depends on whether it is power. Thus, similar to the control of variable phase shifter 114, controller 124 also controls variable amplifier 116 in response to processing of ADC output signal 160 using variable amplifier control signal 164 via signal path 142. Also maintain equal power for variable phase shift signal 154 and variable amplifier signal 150. These powers need not be kept exactly the same, but instead may simply be within a sufficient range of each other so that linear operation of the mixer 118 is ensured. One skilled in the art will appreciate that the variable amplifier 116 may be attenuated in response to the variable amplifier control signal 164 rather than simply amplifying it. As an example, if UUT 102 is an amplifier, variable amplifier 116 may have to attenuate UUT output signal 148 to keep the comparison power of both variable phase shift signal 154 and variable amplifier signal 150 equal. is there. The controller 124 also controls the variable low noise matching amplifier 120 using the variable low noise matching amplifier control signal 166 via the signal path 144 to generate the variable low noise matching output signal 158 as appropriate for the ADC 122. It can be kept within the dynamic range.

  When the component control for quadrature operation is performed, the controller 124 removes the carrier signal from the ADC output signal 160 so that the ADC output signal 160 simply represents phase noise. The phase noise injected by the variable low noise source 112 may be supplemented by a calibration operation so that the UUT 102 is removed and the variable low noise source 112 simply feeds directly to the variable amplifier 116, but this Such a direct supply can occur through a delay line (not shown). The resulting phase noise in the ADC output signal 160 during calibration can be stored in a memory (not shown) associated with the controller 124. Thus, during testing of UUT 102, controller 124 (or spectrum analyzer 104 associated with controller 124) may perform a Fourier analysis of ADC output signal 160 to determine phase noise power. The measured phase noise may then be adjusted by the phase noise injected by the variable low noise source 112 to determine the additional phase noise provided by the UUT 102.

  The phase noise measured in the ADC output signal 160 depends on the frequency of the UUT input signal 146. For example, the UUT 102 may be noisy at one frequency but not so much at another frequency. In order to measure phase noise across the frequency range, the controller 124 commands the variable low noise source 112 to transmit the UUT input signal 146 using the variable low noise source command signal 168 via the signal path 138. The frequency may be changed, the resulting phase noise may be measured, the frequency may be changed again, the resulting phase noise may be measured after changing the frequency, and so on. Advantageously, such measurements are made automatically and accurately, and there is no manual intervention or tracking as required by conventional phase noise test measurement systems.

  Unfortunately, although the APD system 100 represents a significant advance in the art, many factors are involved in properly biasing or driving a given component for optimal low noise performance. Some challenges still remain. As an example, if the UUT 102 is a fiber optic link, the fiber optic link typically requires a manual bias for optimal performance so that it can be properly tested by the APD system 100. Without significant manual intervention from the user, the APD system 100 would not be able to test the fiber optic link.

  As an example, FIG. 2 shows a functional block diagram of a conventional optical fiber link 200. The fiber optic link 200 may include an input amplifier 202, a laser diode 204, a fiber optic channel 206, a photodetector 208, and an output amplifier 210. In this example, laser diode 204 may be in signal communication with input amplifier 202 and fiber optic channel 206 via signal paths 212 and 214, respectively. Photodetector 208 may be in signal communication with fiber optic channel 206 and output amplifier 210 via signal paths 216 and 218, respectively.

In one example of operation, input amplifier 202 amplifies electrical input signal s _in (t) 220 and generates laser input signal 222 to drive laser diode 204. As a result, the laser diode 204 feeds the input optical signal 224 into the optical fiber channel 206 (which can be an optical fiber). After passing through the fiber optic channel 206, the output optical signal 226 is converted into an electrical signal 228 within the photodetector 208. Output amplifier 210 then amplifies electrical signal 228 to provide output signal s _out (t) 230. Unfortunately, many factors are involved in proper biasing of the fiber optic link 200 for optimal performance with an APD system. For example, input amplifier 202 and output amplifier 210 to fiber optic link 200, including appropriately biasing the transistors in input amplifier 202 and output amplifier 210 and appropriately biasing laser diode 204 and photodetector 208, respectively. Is a factor that affects the additional phase noise that the fiber optic link 200 injects into the output signal s _out (t) 230. However, the designer of optical fiber link 200 does not have an intelligent way to set these factors. A similar situation exists for the proper setting of variables in many other systems and devices.

  In addition, the APD system 100 can advantageously be used to characterize the phase noise performance of many different types of UUTs 102, but unfortunately the APD system 100 can carry, for example, a source signal. There is a need for a non-radio signal path to the variable low noise source 112, such as a transmission line that may include a coaxial cable or coplanar waveguide. There are transmitters such as those used in mobile phones where the use of source signals through such signal paths is very inconvenient or impossible.

US Pat. No. 6,793,372

  Therefore, there is a need in the art for an automated system that can measure the phase noise performance of a wireless transmitter through reception and analysis of the transmitted wireless signal.

(Overview)
A noise test measurement system configured to measure a noise component of a transmitted RF signal is disclosed. The noise test measurement system includes an antenna, a low noise amplifier, a frequency source (which may be a local oscillator), a first coupler, a first variable phase shifter, a first mixer, a controller (processor). May be included). The antenna may be configured to receive the transmitted RF signal and provide a received RF signal, and the low noise amplifier may be configured to amplify the received RF signal and provide an amplified RF signal. May be. The frequency source may be configured to provide a frequency reference signal (which may be a LO signal). The first combiner may be configured to combine the first version of the LO signal and the amplified RF signal to provide a combined RF signal, and the first variable phase shifter includes the LO The second version of the signal may be configured to phase shift to a quadrature LO signal responsive to the control signal. The first mixer may have an RF port configured to receive the combined RF signal and an LO port configured to receive the quadrature LO signal. The first mixer may be further configured to mix the signals received at its RF and LO ports to provide a first mixed output signal. The controller may be configured to analyze the digitized version of the first mixed output signal to generate a control signal and to measure a noise component.

  Other systems, methods, features, and advantages of the present invention will become apparent to those skilled in the art from consideration of the following figures and detailed description. All such additional systems, methods, features, and advantages are intended to be included within the scope of this description and the invention and protected by the accompanying claims.

  The present invention can be more fully understood with reference to the following figures. The components in the figures are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. In the figures, like reference numerals designate corresponding parts throughout the different views.

FIG. 1 is a functional block diagram of an example implementation of a known automatic phase locked discriminator measurement noise test system (“APD system”) that is in signal communication with a unit under test (“UUT”) and a spectrum analyzer. FIG. 2 is a functional block diagram of a conventional optical fiber link. FIG. 3 is a functional block diagram of an example implementation of an automated phase-noise measurement system (“ANM system”) according to the present invention. FIG. 4 is a functional block diagram of the UUT shown in FIG. FIG. 5 is a functional block diagram of an example of another implementation of an ANM system according to the present invention. FIG. 6 is a functional block diagram of an example of another implementation of an ANM system according to the present invention. FIG. 7 is a functional block diagram of an example implementation of the adjustable delay line shown in FIG. FIG. 8 is a functional block diagram of an example implementation of a direct downconversion receiver. FIG. 9 is a functional block diagram of an example of another implementation of an ANM system according to the present invention. FIG. 10 is a functional block diagram of an example of another implementation of an ANM system according to the present invention. FIG. 11 is a process diagram showing an example of the operation method of the ANM system according to the present invention.

(Detailed explanation)
In the following description of the preferred embodiments, reference is made to the accompanying drawings that form a part hereof, and which is shown by way of illustration of specific embodiments in which the invention may be practiced. It should be understood that other embodiments may be utilized and changes may be made without departing from the scope of the present invention.

(Unit under test with at least one controllable variable)
FIG. 3 shows a functional block diagram of an example implementation of an automatic phase noise measurement system (“ANM system”) 300 according to the present invention. The ANM system 300 utilizes one or more of the UUTs 302 that respond to phase noise measurements using feedback to minimize the additional phase noise provided by the unit under test (“UUT”) 302. Tune the controllable variables. UUT 302 is in signal communication with ANM system 300 via signal paths 304, 306, and 324, respectively.

  The ANM system 300 includes a variable low noise source 310, a variable phase shifter 312, a variable amplifier 314, a mixer 316, a variable low noise matching amplifier 318, an analog-to-digital converter (“ADC”) 320, And a controller 322. In this example, UUT 302 may be in signal communication with variable low noise source 310, variable amplifier 314, and controller 322 via signal paths 304, 306, and 324, respectively. Mixer 316 may be in signal communication with variable phase shifter 312, variable amplifier 314, and variable low noise matching amplifier 318 via signal paths 326, 328, and 330, respectively. Variable phase shifter 312 is also in signal communication with variable low noise source 310 via signal path 332. ADC 320 is in signal communication with variable low noise matched amplifier 318 and controller 322 via signal paths 334 and 336, respectively. Controller 322 is in signal communication with UUT 302, variable low noise source 310, variable phase shifter 312, variable amplifier 314, and variable low noise matching amplifier 318 via signal paths 324, 338, 340, 342, and 344, respectively. To do. The controller 322 may be a controller device, microcontroller, processor, microprocessor, application specific integrated circuit (“ASIC”), digital signal processor (“DSP”), or other similar programmable device.

  One skilled in the art can refer to the term “signal communication” to pass signals (including both non-information and information-carrying signals) and / or information from one device to another, wireless, wired, analog, and It should be understood to mean any communication and / or electromagnetic, acoustic, digital, or information carrying connection and / or coupling along any signal path between two devices, including / and digital signal paths. “Signal path” refers to, for example, conductive wires, electromagnetic transmission lines, and / or waveguides, attached and / or electromagnetically or mechanically coupled terminals, semiconductive or dielectric materials or devices, or It may be material, such as other similar physical connections or couplings. In addition, the signal path is a free space (in the case of electromagnetic propagation) through digital components where communication information is not necessarily passed directly through an electromagnetic connection, but is passed in various digital formats from one device to another. It may be non-material such as an information transport route.

  UUT 302 may be any device where a user requires a phase noise test measurement, such as an amplifier, phase shifter, diplexer, fiber optic link, or other suitable device or system of devices. Unlike the UUT shown in FIG. 1, the example UUT 302 of the implementation has at least one controllable variable that affects the phase noise that the UUT 302 injects into its output signal.

  As an example of operation, variable low noise source 310 generates a UUT input signal 346 (which may be a low noise carrier signal) for driving UUT 302. UUT 302 may be any device where a user requires a phase noise test measurement, such as an amplifier, phase shifter, diplexer, fiber optic link, or other suitable device or system of devices. UUT 302 receives UUT input signal 346 via signal path 304 and processes it to generate UUT output signal 348. As an example, if UUT 302 is an amplifier, UUT output signal 348 is an amplified version of UUT input signal 346. UUT output signal 348 is received via signal path 306 and amplified by variable amplifier 314 to produce variable amplifier signal 350 that is passed to mixer 316 via signal path 328. Variable low noise source 310 also generates a variable phase shifter input signal 352 that is passed to variable phase shifter 312 via signal path 332. Variable phase shifter input signal 352 is identical to UUT input signal 346 and has the same frequency as UUT input signal 346. The variable phase shifter 312 shifts the variable phase shifter input signal 352 by 90 degrees to generate a variable phase shift signal 354 that is passed to the mixer 316 via the signal path 326. In this method, the carrier signal of the variable amplifier signal 350 is removed from the mixer output signal 356 generated by the mixer 316. In order to keep the mixer output signal 356 within the proper dynamic range of the ADC 320, the mixer output signal 356 is passed through the signal path 330 to the variable low noise matching amplifier 318 and processed by the amplifier to route the signal path 334. To produce a variable low noise matched output signal 358 that is passed to ADC 320.

  In order to remove the carrier signal of the variable amplifier signal 350, the variable phase-shifted signal 354 must be orthogonal (ie, 90 degrees shifted) with respect to the carrier signal. If the quadrature relationship between the variable amplifier signal 350 and the variable phase shift signal 354 is not established, there will be a DC offset in the digital ADC output signal 360 from the ADC 320.

  Similar to the APD system shown in FIG. 1, controller 322 monitors ADC output signal 360 and controls variable phase shifter 312 using variable phase shifter control signal 362 to provide variable amplifier signal 350 and The orthogonal relationship with the variable phase shift signal 354 is maintained. Variable phase shifter control signal 362 is transmitted to variable phase shifter 312 via signal path 340.

  The removal of the carrier signal from the variable amplifier signal 350 is also at equal power when the carrier signal (ie, UUT input signal 346) and the phase-shifted version of the carrier (ie, variable phase-shifted signal 354) enter the mixer 316. It depends on whether there is. Thus, similar to the control of variable phase shifter 312, controller 322 also controls variable amplifier 314 responsive to processing of ADC output signal 360 using variable amplifier control signal 364 via signal path 342. Also hold equal power for the variable phase shift signal 354 and the variable amplifier signal 350. These powers need not be kept exactly the same, but instead may simply be within a sufficient range of each other so that the linear operation of the mixer 316 is ensured. Again, those skilled in the art will appreciate that the variable amplifier 314 may be attenuated in response to the variable amplifier control signal 364 rather than simply amplifying. As an example, if UUT 302 is an amplifier, variable amplifier 314 may have to attenuate UUT output signal 348 to keep the comparison power of both variable phase shift signal 354 and variable amplifier signal 350 equal. is there. Controller 322 also controls variable low noise matching amplifier 318 using variable low noise matching amplifier control signal 366 via signal path 344 to provide variable low noise matching output signal 358 as appropriate for ADC 320. You may hold | maintain within a dynamic range.

  When the component control for quadrature operation is performed, the controller 322 removes the carrier signal from the ADC output signal 360 so that the ADC output signal 360 represents only phase noise. The phase noise injected by the variable low noise source 310 may be supplemented by a calibration operation so that the UUT 302 is removed and the variable low noise source 310 simply feeds directly to the variable amplifier 314, but this Such a direct supply may occur through a delay line (not shown). The resulting phase noise in the ADC output signal 360 during calibration may be stored in a memory (not shown) associated with the controller 322. Accordingly, during testing of UUT 302, controller 322 may perform a Fourier analysis of ADC output signal 360 to determine phase noise power. The measured phase noise may then be adjusted by the phase noise injected by variable low noise source 310 to determine the additional phase noise provided by UUT 302.

  The phase noise measured at the ADC output signal 360 depends on the frequency of the UUT input signal 346. For example, UUT 302 may be noisy at some frequencies but not so much at others. In order to measure phase noise across the frequency range, the controller 322 commands the variable low noise source 310 and the UUT input signal 346 using the variable low noise source command signal 368 via the signal path 338. The frequency may be changed, the resulting phase noise may be measured, the frequency may be changed again, the resulting phase noise may be measured after changing the frequency, and so on.

  In the situation where the UUT 302 is a device with at least one controllable variable that affects the phase noise that the UUT 302 injects into its output signal, the controller 322 sends a UUT command sent to the UUT 302 via the signal path 324. Signal 370 is used to control the value of at least one controllable variable.

  For example, the UUT 302 may include the amplifier 400 shown in FIG. The amplifier may include a set of variable attenuators, shown as first variable attenuator 402 and second variable attenuator 404 for convenience. However, it should be understood that the set of variable attenuators may be more than two variable attenuators. In this example, the first variable attenuator 402 and the second variable attenuator 404 change the amount of additional phase noise generated in the amplifier output signal (ie, UUT output signal 348), Their attenuation may be varied. The first variable attenuator 402 functions to match the amplifier 400 to the input line carrying the amplifier input signal (ie, the UUT input signal 346), while the second variable attenuator 404 is an amplifier. 400 functions to match the output line carrying the amplifier output signal (ie, UUT output signal 348).

  Returning to FIG. 3 of the present example, the controller 322 can control the variable attenuators 402 and 404 as further described herein. As a result, the phase noise measurement relies on the control of at least one controllable variable in the UUT 302. For example, if the UUT 302 includes the amplifier 400 of FIG. 4 within the UUT 302, the measured phase noise depends on the settings of the variable attenuators 402 and 404. The controller 322 thus uses the UUT command signal 370 to change these settings (and therefore the corresponding attenuation provided by the variable attenuators 402 and 404) and result in the digital ADC output signal 360. You may observe the phase noise which arises as follows.

  For example, if the phase noise increases when an increase in attenuation is commanded, the controller 322 repeats the measurement, but may command a smaller amount of attenuation. It will be appreciated that the degrees of freedom and interaction of the variables controlled within the UUT 302 increase in complexity as the number of variables increases. However, the controller 322 may be configured to determine optimal settings for these variables using techniques common to the wireless industry. Since the controller 322 can detect and store the optimized data set, the method of toggling control for such maximum efficiency and noise floor may be automated. Any signal processing method capable of determining an “optimal fit” method or “optimal data fit” may be used to determine the optimal setting for the controllable variable.

  As discussed above, the phase noise measured in the ADC output signal 360 depends on the frequency of the UUT input signal 346. For example, UUT 302 may be noisy at some frequencies but not so much at others. To measure phase noise across the frequency range, the controller 322 uses the variable low noise source command signal 368 to command the variable low noise source 310 to change the frequency of the UUT input signal 346. May be. The process of optimizing phase noise in response to controllable variable variations within UUT 302 is then repeated to record the resulting phase noise, re-frequency, etc.

  In general, the amount and type of variables that can be controlled by the controller 322 within the UUT 302 is virtually infinite. For example, if UUT 302 includes a fiber optic link such as the known fiber optic link 200 discussed in FIG. 2, amplifiers 202 and 210 may be configured with the attenuators discussed with respect to FIG. 4 and introduced by each attenuator. Four variables may be provided, including the amount of attenuation to be performed. In addition (or alternatively), the bias of the transistor (or transistors) in amplifiers 202 and 210 may be a controllable variable. Similarly, the bias of laser diode 204 and photodetector 208 may also be designed as a controllable variable. Once the optimal settings for the controllable variables are determined, the manufacturer then selects a system or device that is configured as determined using the ANM system 300 rather than those that are not controllable. To manufacture.

(Combined UUT and variable low noise source)
One skilled in the art may combine the UUT and variable low noise source into a single device (known as a “source UUT”), such as an oscillator, so that in FIG. It will be understood that the separation from 310 is merely a conceptual separation. In the oscillator example, the variable low noise source is a UUT. Thus, driving this type of source UUT with a separate low noise source is redundant, and therefore no externally generated UUT input signal to transmit to the UUT is required.

  Referring now to FIG. 5, a functional block diagram of an example of another implementation of the ANM system 500 is shown. In this example, ANM system 500 is in signal communication with source UUT 502 via signal paths 504, 506, 508, and 510.

  The ANM system 500 may include a variable phase shifter 512, a delay line 514, a variable amplifier 516, a mixer 518, a variable low noise matching amplifier 520, an ADC 522, and a controller 524. In this example, variable amplifier 516 may be in signal communication with delay line 514, mixer 518, and controller 524 via signal paths 526, 528, and 530, respectively. Mixer 518 may also be in signal communication with variable phase shifter 512 and variable low noise matching amplifier 520 via signal paths 532 and 534, respectively. Variable phase shifter 512 is also in signal communication with source UUT 502 via signal path 510. The ADC 522 is in signal communication with both the variable low noise matching amplifier 520 and the controller 524 via signal paths 536 and 538, respectively. Controller 524 is in signal communication with source UUT 502, variable phase shifter 512, variable amplifier 516, and variable low noise matching amplifier 520 via signal paths 504, 508, 540, 530, and 542, respectively. Again, the controller 524 may be a controller device, microcontroller, processor, microprocessor, ASIC, DSP, or other similar programmable device.

In an example of an operation that analyzes the phase noise of source UUT 502, delay line 514 may generate a delayed output signal 544 of source UUT output signal 546. Source UUT output signal 546 is generated by source UUT 502. Those skilled in the art will appreciate that in the situation where source UUT 502 is a complete source capable of producing a sinusoidal output signal equal to cos (ωt), the phase difference between any time t ₁ and t ₂ is It should be understood that it depends only on the delay period between these times. However, in real world sources, there will also be some phase noise that affects this phase difference. In general, the choice of delay period may be shown to affect the ability of ANM system 500 to measure phase noise at a smaller frequency offset relative to the frequency of the carrier signal, and the sensitivity of the phase noise measurement. As the delay provided by delay line 514 increases, the ability to measure phase noise at a smaller offset from the carrier frequency, and the sensitivity, increase. However, the attenuation through delay line 514 becomes too sensitive and can affect the measurement, so the delay cannot be increased arbitrarily.

  The discussion of FIGS. 3 and 5 shows basic similarities between the characterization of source UUT and non-source UUT. Regardless of whether the UUT is a source UUT or a non-source UUT, the variable amplifier 314 or 516, the mixer 316 or 518, the variable low noise matching amplifier 318 or 520, the ADC 320 or 522, and the variable phase shifter 312 Or, the control and operation of 512 are the same. Thus, similar to FIG. 3, source UUT 502 provides a variable phase shifter input signal 548 to phase variable phase shifter 512, similar to the discussion regarding variable low noise source 310 of FIG. Variable amplifier 516 receives delayed output signal 544 and generates variable amplifier signal 550 that is passed to mixer 518, which includes variable amplifier signal 550 and a 90 degree phase shift of variable phase shifter input signal 548. Mix with the variable phase shift signal 552 generated by the variable phase shifter 512 (which is a version) to generate the mixer output signal 554. Controller 524 controls variable amplifier 516 with variable amplifier control signal 556 to maintain the linear operation of mixer 518. Controller 524 additionally controls variable phase shifter 512 by variable phase shifter control signal 558 and variable low noise matching amplifier control signal 559 by variable phase shifter control signal 559 via signal paths 540 and 542, respectively. To control. The controller 524 uses at least one controllable in the source UUT 502 using the first UUT command signal 560 via the signal path 508, similar to the discussion regarding tuning of the non-source UUT 302 of FIG. Operates to tune variables. Similarly, the controller 524 uses the second UUT command signal 562 via the signal path 504 to control the carrier frequency used by the source UUT 502 as discussed with respect to FIG. Again, similar to the example of FIG. 3, the controller 524 receives the receive ADC via signal path 538 generated by the ADC 522 in response to receipt of the variable low noise matched amplifier output signal 566 via signal path 536. Responsive to monitoring the output signal 564, the devices and / or modules in the ANM system 500 are controlled.

  FIG. 6 shows a functional block diagram of an example of another implementation of an ANM system 600 that is in signal communication with a source UUT 602 via signal paths 604, 606, 608, and 610. Since the amount of delay provided by the delay line has a strong impact on the characterization of the phase noise from the source UUT 602, the ANM system 600 (unlike the ANM system 500 shown in FIG. 5) is an adjustable delay line. Includes a selectable delay feature shown as 612. However, like the ANM system 500 of FIG. 5, the ANM system 600 also includes a variable phase shifter 614, a variable amplifier 616, a mixer 618, a variable low noise matching amplifier 620, an ADC 622, and a controller 624. But you can. In this example, variable amplifier 616 may be in signal communication with adjustable delay line 612, mixer 618, and controller 624 via signal paths 626, 628, and 644, respectively. Mixer 618 may also be in signal communication with variable phase shifter 614 and variable low noise matching amplifier 620 via signal paths 632 and 634, respectively. Variable phase shifter 614 is also in signal communication with source UUT 602 via signal path 610. The ADC 622 is in signal communication with both the variable low noise matching amplifier 620 and the controller 624 via signal paths 636 and 638, respectively. Adjustable delay line 612 is also in signal communication with source UUT 602 and controller 624 via signal paths 608 and 640, respectively. The controller 624 is also in signal communication with the source UUT 602, variable phase shifter 614, variable amplifier 616, variable low noise matching amplifier 620, and ADC 622 via signal paths 604, 642, 644, 646, and 648, respectively. Again, the controller 624 may be a controller device, microcontroller, processor, microprocessor, ASIC, DSP, or other similar programmable device.

  The operation of the ANM system 600 is similar to the operation of the ANM system 500 shown in FIG. 5 except for differences in the operation and control of selectable delay features for the uncontrolled delay line 514 shown in FIG. In FIG. 6, this selectable delay feature is provided by an adjustable delay line 612 that delays the source UUT output signal 650 received via signal path 608 generated by source UUT 608. Based on the desired sensitivity and frequency offset of the phase noise measurement, the controller 624 uses the adjustable delay line control signal 652 transmitted via the signal path 640 to delay the delay line within the adjustable delay line 612. (Not shown) is selected.

As an example, adjustable delay line 612 may include a plurality of delay lines that provide a corresponding plurality of delays from delay τ ₁ to delay τ _N shown in FIG. For clarity, a plurality of delay lines included within the adjustable delay line 612 includes a first delay line 700 having a delay τ ₁ , a second delay line 702 having a delay τ ₂ , and a delay τ _N. Only the Nth delay line 704 having is shown in FIG. These delay lines 700, 702, and 704 are selected by the controller 324 via an adjustable delay line control signal 652 through the operation of a switch such as a field effect transistor (“FET”) (not shown). May be.

  In one example of operation, returning to FIG. 6, regardless of which delay line within the adjustable delay line 612 is selected, the variable amplifier 616 then (signal path), similar to the process discussed with respect to FIG. The received delayed output signal 654 from the adjustable delay line 612 (via 626) is amplified and responsive to producing a variable amplifier output signal 656 that is passed to the mixer 618. Similarly, variable phase shifter input signal 658 is generated by source UUT 602 and passed to variable phase shifter 614 via signal path 610. Variable phase shifter 614 receives variable phase shifter input signal 658 and shifts it 90 degrees to produce a variable phase signal 660 that is also received by mixer 618. In response, mixer 618 mixes variable amplifier output signal 656 and variable phase shift signal 660 to produce mixer output signal 662 that is passed to variable low noise matched amplifier 620 via signal path 634.

  In this example, source UUT 602 includes at least one controllable variable, such as a transistor bias voltage or other variable that can be controlled by source UUT control signal 664 from controller 624 transmitted via signal path 604, for example. including. Accordingly, ANM system 600 operates similarly to the discussion of operation of ANM system 500 of FIG. In response to measuring the phase noise of source UUT 602 using digital ADC output signal 666 (via signal path 648) generated by ADC 622, controller 624 optimizes the phase noise performance of ANM system 600. As such, the second source UUT control signal 668 is driven (via signal path 606) to tune at least one controllable variable in source UUT 602.

  Although the invention has been described with reference to specific implementation examples, it is to be understood that the description is only an example of the invention's application and should not be taken as a limitation. As an example, in both FIGS. 5 and 6, the order of variable amplifier (514 or 612) and delay line 514 or adjustable delay line 612 is such that the delay line (or adjustable delay line) is the variable amplifier output signal. May be reversed, such that a delayed version of is provided to the mixer (either 518 or 618).

(Radio phase noise measurement)
As an example of another implementation, the ANM system can measure the noise performance of a wireless transmitter through reception and analysis of the transmitted wireless signal. US Pat. No. 6,745,020 (named “Direct Downconversion Receiver”, filed Aug. 29, 2002), which is hereby incorporated by reference in its entirety to implement this implementation. May be adapted to allow a user to remotely measure the noise floor of a wireless transmitter.

  Referring to FIG. 8, a functional block diagram of an example implementation of a direct downconversion receiver 800 (described in US Pat. No. 6,745,020) is shown. Direct downconversion receiver 800 may include a signal source 802, a variable phase shifter 804, a mixer 806, a first combiner 808, a second combiner 810, and an antenna 812. In this example, the first combiner 808 may be in signal communication with the signal source 802 and the second combiner 810 via signal paths 814 and 816, respectively. Second combiner 810 may also be in signal communication with both mixer 806 and antenna 812 via signal paths 818 and 820, respectively. Variable phase shifter 804 may also be in signal communication with first combiner 808 and mixer 806 via signal paths 822 and 824, respectively.

  Direct downconversion receiver 800 can mitigate DC offset components in baseband signal output 826 from mixer 806. Mixer 810 has its baseband signal output 826 whose product is the product of mixer input signal 828 (via signal path 818) to mixer 806 and frequency reference signal 830 (local oscillator (local) for mixer 806 via signal path 824). oscillator: “LO”) signal, etc.), which may be either a double sideband or a single sideband mixer.

  In one example of operation, direct downconversion receiver 800 receives an incoming radio frequency (“RF”) signal 832 having a given carrier frequency at antenna 812. In response, antenna 812 generates a received RF signal 834 that is passed to second combiner 810 via signal path 820. The second combiner 810 then combines the received RF signal 834 with the mixer input signal 828 and passes it to the mixer 806. Those skilled in the art will appreciate that the incoming RF signal 832 is received by antenna 812 and processed by a filter and low noise amplifier (not shown) before being coupled to mixer 806, as is conventional in receiver architecture. It will be understood that it may be.

  The signal source 802 may be a voltage controlled oscillator (“VCO”) or another similar generator with a suitable LO signal output. As an example, signal source 802 may generate a sine output signal 836. One skilled in the art will appreciate that the type of LO signal output depends on the particular modulation present in the incoming RF signal 832. Prior to passing mixer input signal 828 to mixer 306, first combiner main output signal 838 and received RF signal 834 are combined through second combiner 810. First combiner main output signal 838 is generated by passing sine output signal 836 through first combiner 808. The first combiner 808 also generates a variable phase shifter input signal 840 that is passed to the variable phase shifter 804 via the signal path 822. It will be appreciated that a signal distributor may also be utilized in place of the first combiner 808. Variable phase shifter 804 provides a quadrature local oscillator signal (ie, frequency reference signal 830) to LO input port 842 of mixer 806 (ie, a port that receives frequency reference signal 830 via signal path 824). The sine output signal 836 is configured to shift 90 degrees. In this example, variable phase shifter 804 is shown as a functional block of discrete components, but it should be understood that variable phase shifter 804 may also be included in mixer 806. In the example of variable phase shifter 804 that is part of mixer 806, frequency reference signal 830 is coupled to RF input port 844 of mixer 806 (ie, the port that receives mixer input signal 828 via signal path 818), Thereby, the opportunity to self-couple and generate unwanted DC offset components in the baseband signal output 826 is minimized. Alternatively, variable phase shifter 804 may be located as close as possible to LO output port 842 of mixer 806 so as to minimize radiative or reactive coupling of frequency reference signal 830 to RF input port 844.

  The design of the first and second couplers 808 and 810 is based on the type of waveguide selected to propagate the received RF signal 834 and the sinusoidal output signal 836 using RF techniques known to those skilled in the art. For example, it should be understood that it depends on stripline, coaxial cable, or microstrip waveguide, etc.). As a result, the type of waveguide implemented depends on the carrier frequency, signal power level, spatial concerns, and other known design choices.

  In this implementation, the baseband signal output 826 includes a frequency offset term that is the sine of the frequency difference between the RF frequencies for the received RF signal 834 and the sine output signal 836. In addition, various dual frequency terms are generated and may be easily filtered. In this way, the baseband signal output 826 can be achieved without the DC offset problem present with other direct downconversion receivers.

  Referring to FIG. 9, an ANM system 900 is shown that includes adaptations of the direct downconversion receiver architecture discussed with respect to FIG. The ANM system 900 includes a source LO 902, a variable amplifier 904, a variable phase shifter 906, a mixer 908, a variable low noise matching amplifier 910, an ADC 912, a controller 914, an antenna (or multiple antennas) 916, A low-noise amplifier (“LNA”) 918, a combiner 920, and an optional isolator 922 may be included. Variable amplifier 904 may be in signal communication with mixer 908, controller 914, and combiner 920 via signal paths 924, 926, and 928, respectively. Mixer 908 may also be in signal communication with variable phase shifter 906 and variable low noise matching amplifier 910 via signal paths 930 and 932. Variable low noise matching amplifier 910 may be in signal communication with ADC 912 and controller 914 via signal paths 934 and 936, respectively. Controller 914 may be in signal communication with variable phase shifter 906 and ADC 912 via signal paths 938 and 940, respectively. Source LO 902 may be in signal communication with variable phase shifter 906 and optional isolator 922 via signal paths 942 and 944, respectively. Combiner 920 may be in signal communication with LNA 918 and optional isolator 922 via signal paths 946 and 948, respectively. LNA 918 may also be in signal communication with antenna 916 via signal path 930.

  In one example of operation, wireless signal 950 is received at antenna 916 (or multiple antennas) and processed through LNA 918 to generate RF received signal 952. In this example, source LO 902 is configured similarly to the discussion with respect to FIG. Source LO 902 generates two identical signals, referred to as source LO output signal 954 and variable phase shifter input signal 956. However, in this example, source LO output signal 954 or variable phase shifter input signal 956 functions as an LO for received RF signal 952. Thus, the variable phase shifter input signal 956 is variable phase shift to generate a quadrature LO signal (ie, variable phase shift output signal 960) that is received at LO port 962 of mixer 908 via signal path 930. The phase is shifted through the device 906. Received RF signal 952 is combined with source LO output signal 954 using combiner 920 to form composite combiner output signal 964. In this example, source LO output signal 954 may be processed by optional isolator 922 to form a disconnected source LO output signal 966. If no optional isolator 922 is utilized, the disconnected source LO output signal 966 is identical to the source LO output signal 954.

  Similar to the discussion with respect to FIG. 5, the resulting variable amplified output signal 968 applied to the RF port 970 of the mixer 908 when compared to when applied to the LO port 962 to the extent that the mixer 908 operates linearly. The variable amplifier 904 may amplify the combined combiner output signal 964 so that is of sufficient power. In this regard, depending on the design and linearity of the mixer 908, the variable amplifier 904 may not be necessary in some implementations. Mixer output signal 972 from mixer 908 on signal path 932 may then be processed as discussed with respect to FIG. In this implementation, using control signal 974 via signal path 938, 976 via signal path 926, and 978 via signal path 936, respectively, controller 914 uses variable phase shift as previously discussed. 906, variable amplifier 904, and variable low noise matching amplifier 910 are controlled. To prevent the received RF signal 952 from contaminating the quadrature LO signal (ie, the source LO output signal 954), the source LO 902 may be disconnected from the coupler 920 using the optional isolator 922. Good. However, if the coupler 920 is sufficiently directional, the optional isolator 922 may not be necessary. By performing a spectral analysis, such as FFT, on the digitized noise signal 980 generated by the ADC 912 on the signal path 940, the controller 914 provides the noise floor of a wireless transmitter (not shown) that generates the wireless signal 950. May be characterized. In this example, the noise measurement is performed completely wirelessly without the need for any transmission line connecting the ANM system 900 to the wireless transmitter.

  In another implementation, a delay line (not shown) may also be used as in the discussion with respect to FIG. This delay line may be inserted in front of or behind the coupler 920. However, the function of the quadrature discriminator with respect to the received RF signal 952 is that the resulting discriminating function is applied to the source LO output signal 954 and the variable phase shifter input signal 956 instead of the received RF signal 952. It will be understood that this is not achieved through the inclusion of such delay lines. However, there may also be implementations where it is advantageous to perform a discrimination function with respect to source LO902. In any case, the source LO 902 should have a lower noise floor than the radio transmitter (not shown) used to transmit the radio signal 916 so that additional noise measurements may be performed. An example of a low noise source that may be configured as source LO902 is described in US patent application Ser. No. 11 / 494,884 filed Jul. 21, 2006, the contents of which are referenced. Is incorporated herein by reference in its entirety. However, it will be understood that any suitable low noise source may be used to form source LO902. Initial calibration may be performed to determine the noise floor of source LO 902 so that additional noise due to combination with received RF signal 952 may be determined. With respect to the received RF signal 952, the radio signal 950 will be represented as a frequency difference (offset from LO) in the mixer output signal 972, since a downconversion architecture has been created instead of an orthogonal discriminator. Between this signal and DC is the transmitter noise spectrum. The dual frequency component may be removed using an appropriate signal filter (not shown) as discussed with respect to FIG. Unlike the discriminator discussed in FIG. 5, the spectrum of this digitized noise signal 980 will represent amplitude noise and phase noise.

  As a result, FIG. 10 shows a functional block diagram of an example implementation of the ANM system 1000 that can measure the amplitude noise and phase noise of the wireless transmitter 1002 separately. The ANM system 1000 includes a source LO 1004, a variable amplifier 1006, a first variable phase shifter 1008, a second variable phase shifter 1010, a first mixer 1012, a controller 1014, and a first combination. A combiner 1016, a second combiner 1018, an antenna 1020, a first LNA 1022, an amplitude and phase (“AM and PM”) modulator 1024, an optional isolator 1026, a second mixer 1028, A distributor 1030, a second LNA 1032, a third LNA 1034, a first ADC 1036, and a second ADC 1038 may be included. Source LO 1004 is also in signal communication with first variable phase shifter 1008, second variable phase shifter 1010, controller 1014, and optional isolator 1026 via signal paths 1040, 1042, and 1044, respectively. Good. Distributor 1030 converts any signal generated by source LO 1004 (and transmitted via signal path 1040) via first variable phase shifter 1008 and second variable via signal paths 1045 and 1046, respectively. It can be divided into two distribution signals transmitted to the variable phase shifter 1010. The variable amplifier 1006 may be in signal communication with the first combiner 1016, the second combiner 1018, and the controller 1014 via signal paths 1048, 1050, and 1052, respectively. Second combiner 1018 may be in signal communication with first mixer 1012 and second mixer 1028 via signal paths 1053 and 1054, respectively. First variable phase shifter 1008 may be in signal communication with first mixer 1012 and controller 1014 via signal paths 1056 and 1058, respectively. Similarly, second variable phase shifter 1010 may be in signal communication with second mixer 1028 and controller 1014 via signal paths 1060 and 1062, respectively. First coupler 1016 may be in signal communication with optional isolator 1026 and AM and PM modulator 1024 via signal paths 1064 and 1066, respectively. First LNA 1022 may be in signal communication with antenna 1020 and AM and PM modulator 1024 via signal paths 1067 and 1068, respectively. The wireless transmitter 1002 may perform signal communication with the antenna 1020 via a signal path 1069 which is a wireless signal path in this example. In addition, the controller 1014 may be in signal communication with the transmitter 1002 via a signal path 1070 that may also be a wireless signal path.

  In one example of operation, the wireless transmitter 1002 transmits the wireless signal 1071 to the antenna 1020 (or multiple antennas) that receives the wireless signal 1071 and generates a received signal 1072 that is passed to the LNA 1022 via the signal path 1067. . The LNA 1022 processes the received signal 1072 to generate a received RF signal 1073 that is transmitted to the AM and PM modulator 1024 via the signal path 1068. The AM and PM modulator 1024 receives the received RF signal 1073 and generates AM modulation and / or phase PM techniques to produce a modulated received signal 1074 that is passed to the first combiner 1016 via the signal path 1066. This is applied to the received RF signal 1073.

  The source LO 1004 generates a source LO output signal 1075 that is passed through the optional isolator 1026 via signal paths 1044 and 1064 to the first combiner 1016. If the optional isolator 1026 is present, the optional isolator 1026 generates a disconnect source LO output signal 1076. In the absence of optional isolator 1026, disconnect supply LO output signal 1076 is the same as supply LO output signal 1075. Again, as described in FIG. 9, the purpose of the optional isolator 1026 is to prevent the modulated received signal 1074 from contaminating the source LO output signal 1075, and therefore the optional isolator 1026 Disconnecting the source LO 1004 from one coupler 1016. However, if the first coupler 1016 is sufficiently directional, the optional isolator 1026 may not be necessary.

  The first combiner 1016 supplies a disconnected source LO output signal 1076 (or an optional isolator 1026 if not present) to generate a combined received signal 1077 that is passed to the variable amplifier 1006 via signal path 1048. The source LO output signal 1075) and the modulated received signal 1074 are combined. The variable amplifier 1006 receives the combined received signal 1077 and amplifies it so as to generate an amplifier received signal 1078 that is passed to the second combiner 1018 via the signal path 1050. The second combiner 1018 outputs the amplified received signal 1078 to the first amplified received signal 1079 (via signal path 1053) passed to the first mixer 1012 and the second (via signal path 1054). This is distributed to the second amplified received signal 1080 passed to the mixer 1028.

  Source LO 1004 also generates a variable phase shifter input signal 1081 that is transmitted via signal path 1040 to distributor 1030 (or other suitable device that can distribute the signal into two signals). Divider 1030 provides variable phase shifter input signal 1081 to generate first variable phase shifter input signal 1082 and second variable phase shifter input signal 1083 along signal paths 1045 and 1046, respectively. Receive and distribute it. The first variable phase shifter input signal 1082 is processed within the first variable phase shifter 1008 to provide a quadrature LO signal 1084 to the first mixer 1012 as discussed with respect to FIG. . In addition, the second variable phase shifter input signal 1083 is processed within the second variable phase shifter 1010 to provide an in-phase LO signal 1085 to the second mixer 1028. As described above, both mixers (1012 and 1028) receive a distributed version of the amplified received signal 1078.

  Since the second mixer 1028 mixes the in-phase signal LO 1085 and the second amplified received signal 1080, the resulting second mixer output signal 1086 is similar to that discussed with respect to FIG. Represents the in-phase (“I”) component after processing this second mixer output signal 1086 through the LNA 1032 and the first ADC 1036. This resulting I component signal 1087 may be passed to controller 1014 via signal path 1088. Similarly, the first mixer 1012 mixes the quadrature LO signal 1084 and the first amplified received signal 1079 to generate a first mixer output signal 1089. The first mixer output signal 1089 represents a digital quadrature (“Q”) component signal 1090 that is processed by the third LNA 1034 and the second ADC 1038 and passed to the controller 1014 via the signal path 1091.

  In this implementation, the determination of whether amplitude or phase noise is being measured along with these I and Q component signals 1087 and 1090 is the modulation of the modulation utilized in the received RF signal 1073 within the AM and PM modulator 1024. Controlled by type. As an example, if the AM and PM modulator 1024 then applies a predetermined AM modulation to the received RF signal 1073, the ANM system 1000 may add additional amplitude noise and addition within the wireless transmitter 1002. The effect of these modulations on the I and Q component signals 1090 and 1087 may be analyzed to determine the amount of phase noise that is likely.

  In addition to determining the noise floor within the wireless transmitter 1002, the ANM system 1000 can also tune the wireless transmitter 1002 using the techniques described in US patent application Ser. No. 11 / 134,546. As an example, the wireless transmitter 1002 may be a transistor (not shown) or other suitable tunable device that may be controlled by the controller 1014 using the wireless transmitter control signal 1091 via the signal path 1070. Bias settings. By observing the effect on the noise spectrum through analysis of the I and Q component signals 1090 and 1087 (or their combined effects discussed with respect to FIG. 9), the controller 1014 allows the wireless transmitter 1002 to transmit wirelessly. Radio transmitter control signal 1091 (which may be designated as a control variable (CV)) may be adjusted to be tuned to lower the noise floor of transmitter 1002. Controller 1014 analyzes the digitized noise signal (or signal) and adjusts the corresponding control variable (CV) accordingly to keep its output orthogonal (via signal path 1058). Via the first variable phase shifter control signal 1092, where the first variable phase shifter control signal 1092 is also another CV) and the first variable phase shifter 1008, and , So that the output is held in phase with the source LO output signal 1075 (via signal path 1062, via second variable phase shifter control signal 1093, where second variable phase shifter The control signal 1093 also controls the second variable phase shifter 1010 (which is another CV).

As an example, the first variable phase shifter 1008 is first variable to be orthogonal to the amplified signal 1078 from the variable low noise amplifier 1006 (essentially a related version of the source LO output signal 1075). It is controlled by a phase shifter control signal 1092. However, if the carrier signal is not removed due to an error in the first variable phase shifter control signal 1092, the carrier signal is present as a DC offset in the resulting noise spectrum. This DC offset has a polarity that is in one of the quadrature phases and an opposite polarity to the other of the quadrature phases, so that the first variable phase shifter 1008 has an approximately orthogonal angle (eg, 80-100 degrees). ) Change the sign when scanning. The effect of CV on measurable variables ("MV") such as DC offset may be used for various control algorithms. For example, a change of sign within the DC offset may be used for a zero crossing search. In general, a CV that produces a zero crossing MV may have its range divided into several intervals. The controller 1014 steps the CV through these intervals and observes the impact on the zero crossing MV. For example, the zero crossing MV may change the sign for the two values MV ₀ and MV ₁ , corresponding to the values of CV ₀ and CV ₁ , respectively. Considering the crossing of the zero crossing, the optimum setting of CV (“CV _optimal ”) can be expressed by the following equation.

一般に、第１の可変移相器制御信号１０９２等のＣＶ_{ｏｐｔｉｍａｌ}は、温度変化および他の影響によって時間とともに変化する。この時間に対する変化は、収束アルゴリズムを使用して追跡してもよい。例えば、（ＣＶ_１およびＣＶ_０に対応する）またがる間隔は、２のような収束因子によって低減されてもよい。ＣＶ_{ｏｐｔｉｍａｌ}の新しい値は、次いで、例えば、式（１）を使用して計算される。連続する測定値間の差は、次いで、時間的に変化する補正因子を提供するように、以前に得られた差とともに平均されてもよい。算出されたＣＶ_{ｏｐｔｉｍａｌ}は、次いで、時間的に変化する補正因子に従って調整されてもよい。ＣＶを更新する前に、許容因子に関するＭＶの測定を実行してもよい。

In general, the CV _optimal , such as the first variable phase shifter control signal 1092, changes over time due to temperature changes and other effects. This change over time may be tracked using a convergence algorithm. For example, the span span (corresponding to CV ₁ and CV ₀ ) may be reduced by a convergence factor such as 2. The new value of the CV _optimal can then, for example, is calculated using Equation (1). The difference between successive measurements may then be averaged along with previously obtained differences to provide a time-varying correction factor. The calculated CV _optimal may then be adjusted according to a time-varying correction factor. Prior to updating the CV, a MV measurement for the tolerance factor may be performed.

The second variable phase shifter 1010 is controlled in a complementary manner that the output signal of the second mixer 1028 should have maximum carrier power. A second variable phase shifter control signal 1093 (also another CV) that controls the second variable phase shifter 1010 is controlled by the controller 1014 to find the maximum value of the corresponding MV. Also good. For example, the output signal of the second mixer 1028 may be such that if five consecutive increments of CV vary from CV ₀ to CV ₄ , the corresponding value of MV is MV ₂ > MV ₁ > MV ₀ and also MV _It may be increased over its entire range so that ₂ > MV ₃ > MV ₄ is generated. The value of CV to generate a maximum of MV, therefore, the CV _2. If no maximum value is found over the entire available range of CVs, the interval between successive CVs may be too wide, and a new search is performed with the interval reduced in half. Once the maximum is found, the interval between CV ₀ and CV ₄ is sampled at twice the previous rate so that the interval between successive CV points used in previous searches is halved. . If the maximum pattern can no longer be identified, it may be assumed that the algorithm has zoomed to noise at the maximum value of the MV variable. The maximum value of MV for the last iteration of CV that can be identified by the pattern provides the corresponding CV _optimal value. This CV _optimal value may be tracked as discussed for zero crossing MV control.

  Additional control is provided via the control signal 1094 along the control path 1042 to the source LO 1004 (if desired for frequency change), to the variable amplifier 1006 (via the control signal 1095 along the signal path 1052), And other components such as LNAs 1032 and 1034 (which may be variable LNAs). Empirical observations in the ANM system 1000 may indicate that a single CV can have a dominant impact on other CVs. For example, the first variable phase shifter control signal 1092 can be the dominant CV for the zero crossing in the first mixer output signal 1089. Thus, in this example, the dominant CV may be tuned first, followed by a less dominant CV. However, it will be appreciated that this tuning method can be easily extended to parallel control of multiple CVs.

  Referring to FIG. 11, a flow diagram illustrating an example of how an ANM system operates using feedback to tune one or more controllable variables in a unit under test (“UUT”) is shown. The example process 1100 begins at step 1102, where a variable low noise source generates an RF signal and transmits it to a UUT and a variable phase shifter. In step 1106, the UUT receives the RF signal and processes it to produce an output signal, which is then passed to the variable amplifier, which in step 1108 amplifies and variably amplifies the output signal. Generate a signal. In step 1110, the variable phase shifter receives the RF signal, and the variable phase shifter shifts the RF signal by 90 degrees to generate a variable phase shift signal.

  The variable amplification signal and variable phase shift signal are then passed to the mixer at step 1112. At step 1114, the mixer generates a mixer output signal that is transmitted to a variable low noise matching amplifier, and at step 1116, the variable low noise matching amplifier is passed to an analog-to-digital converter ("ADC"). A matched output signal is generated. The ADC then generates a digital ADC output signal that is passed to the controller at step 1118. In step 1120, the controller monitors the digital ADC output signal and utilizes the variable phase shifter control signal, variable amplifier control signal, variable low noise matching amplifier control signal, respectively, to control the variable phase shifter, variable amplifier, and variable. Controls the low noise matching amplifier and commands the variable low noise source to change the frequency of the UUT input signal utilizing the variable low noise source command signal.

  In addition, the controller is configured to utilize the UUT command signal to control the value of at least one controllable variable of the UUT. As described above with respect to FIG. 4, an example of such a controllable variable may be a variable attenuator included within the UUT. At decision step 1122, the controller determines whether an optimal setting of at least one controllable variable has been obtained, eg, there is minimal phase noise. If the optimal settings have been obtained, the process 1100 ends at step 1124. If not, the process returns to step 1104 and repeats itself.

  Those skilled in the art will understand and appreciate that one or more of the functions, modules, units, blocks, processes, subprocesses, or process steps described above may be performed by hardware and / or software. Let's go. If the process is performed by software, the software may reside in software memory (not shown) in any of the devices described above. The software in the software memory may include an ordered list of executable instructions for implementing the logic function (ie, in digital form, such as a digital circuit or source code, or an analog circuit, or analog electrical "Logic", which may be implemented in analog form, such as an analog source such as an audio or video signal), and selectively fetching instructions from an instruction execution system, apparatus or device and executing the instructions Any computer readable (or signal) for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, a system including a processor, or other system (Supported) may be selectively incorporated in the medium. In the context of this document, “computer-readable medium” and / or “signal-bearing medium” includes a program for use by or in connection with an instruction execution system, apparatus, or device. Any means that can, communicate, propagate or carry. The computer readable medium may be, for example but not limited to, an electronic, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, device, or propagation medium. More specific examples of computer readable media are still a non-limiting list, but electrical connections (electrical), portable computer diskettes (magnetic), RAM (electrical) with one or more wires. ), Read-only memory “ROM” (electric), erasable and programmable read-only memory (EPROM or flash memory) (electric), optical fiber (optical), and portable compact disk read-only memory “CDROM” (optical) ). A computer readable medium, for example, electronically captures the program via optical scanning of paper or other media, then compiles, translates, or otherwise processes in a suitable manner as necessary, and then processes the computer Note that the program may be printed paper or another suitable medium, since it can be stored in memory.

  While various embodiments of the invention have been described, it will be apparent to those skilled in the art that many more embodiments and implementations are possible within the scope of the invention. Accordingly, the invention is not limited except in light of the attached claims and their equivalents.

Claims (12)

A noise test measurement system configured to measure a noise component of a transmission RF signal, comprising:
An antenna for receiving the transmitted RF signal and providing a received RF signal;
A low noise amplifier configured to amplify the received RF signal and provide an amplified RF signal;
A local oscillator configured to provide an LO signal;
A first combiner configured to combine a first version of the LO signal and the amplified RF signal to provide a combined RF signal;
A first variable phase shifter configured to phase shift a second version of the LO signal to a quadrature LO signal responsive to the control signal;
A first mixer having an RF port configured to receive the combined RF signal and an LO port configured to receive the quadrature LO signal, the first mixer comprising the RF port A first mixer further configured to mix the signals received at the port and the LO port to provide a first mixed output signal;
A processor configured to analyze a digitized version of the first mixed output signal to generate the control signal and to measure the noise component.   The system of claim 1, further comprising an analog-to-digital converter that provides the digitized version.   The system of claim 2, further comprising a second variable amplifier configured to variably amplify the mixer output and provide a signal to the analog-to-digital converter.   The system of claim 3, wherein the processor is further configured to control the second variable amplifier to maintain its output within a desired dynamic range of the analog-to-digital converter. A second variable phase shifter configured to phase shift a third version of the LO signal to provide an in-phase LO signal responsive to a second control signal;
A second mixer having an RF port configured to receive the combined RF signal and an LO port configured to receive the quadrature LO signal, wherein the second mixer has its RF A second mixer further configured to mix the signals received at the port and the LO port to provide a second mixed output signal, the processor comprising the second mixed output The system of claim 1, further configured to analyze a digitized version of a signal to generate the second control signal and to measure the noise component.   The system of claim 5, wherein the processor is configured to determine an amplitude noise measurement and a phase noise measurement from the noise component.   The wireless transmitter for providing the transmitted RF signal includes a tunable variable, and the processor is configured to tune the tunable variable to reduce the measurement noise component. The system described.   The system of claim 7, wherein the tunable variable is a transistor bias setting. A method for measuring a noise floor for a radio transmitter, comprising:
Transmitting an RF signal from the wireless transmitter to provide a transmitted RF signal;
Receiving the transmitted RF signal to provide a received RF signal;
Combining the received RF signal with a version of the LO signal to provide a combined RF signal;
Mixing the combined RF signal and an orthogonal version of the LO signal to provide a mixer output signal;
Analyzing the spectrum of the mixer output signal to determine the noise floor.   The method of claim 9, further comprising adjusting a power of the combined RF signal to ensure that the mixing is linear. Analyzing the spectrum includes
Digitizing the mixer output signal to form a digitized signal;
The method of claim 9, comprising: fast Fourier transforming the digitized signal.   The method of claim 11, further comprising controlling phase shift based on the spectral analysis.

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