KR20090067919A - Method of estimating the parameter of induction motor - Google Patents

Abstract

A method for estimating a parameter of an induction motor is provided to remove the influence of ZZC(Zero Current Clamping) by preventing the current from flowing around 0 and applying the current with a DC component and an AC component together. The first equivalent impedance including the first equivalent resistance of the induction motor according to the first current and the first equivalent inductance is measured by applying a first current including the d axis current and the q axis current of a first frequency to the induction motor(310). The second equivalent impedance including the second equivalent resistance of the induction motor according to the second current and the second equivalent inductance is measured by applying a second current including the d axis current and the q axis current of a second frequency to the induction motor(320). The stator resistance of the induction motor is estimated based on the relation between the frequency of the first and second current and the measured first and second equivalent resistance(330).

Description

Method of Estimating the Parameter of Induction Motor

The present invention relates to a method for estimating the constant of an induction motor, and more particularly, while maintaining the induction motor in a stopped state, the stator resistance of the induction motor is applied to the induction motor by applying a current in which the DC component and the AC component are combined. And a method for estimating the constants of an induction motor based on the estimated stator resistance.

In recent years, the use of induction motors has been prominent in many fields of the industry because of their simple structure and low cost. However, induction motors have the disadvantage of being difficult to control despite these advantages.

In general, the flow motor independently controls the magnetic flux (FLUX) and the torque (TORQUE) through a vector control inverter. In order to ensure the performance of the induction motor high performance drive system, it is very important to set the motor constants correctly and initially. Measure the constants of stator resistance (Rs), rotor resistance (Rr), excitation inductance (Lm), stator leakage inductance (L _ls ), rotor leakage inductance (L _lr ) and apply them You're in control. In particular, there is the constant of the rotor leakage inductance (L _lr) in a considerable part of the use, in which the rotor leakage inductor's (L _lr) is a sensitive parameter to the large change in the number - several orders of magnitude depending on the temperature and the rotational speed of the motor It is necessary to estimate without disturbing the operation of the car.

The study of induction motor constant estimation can be divided into two areas: offline and online estimation.

Off-line estimation methods include estimating the constants necessary for the initial control of the motor from classical no-load tests and restraint tests, or from motor nameplate information. This off-line method helps to calculate the approximate motor constants, but it is not easy to use the constants obtained directly for motor control. Recently, a method of measuring the rotor resistance by forcibly injecting a pseudo random binary signal (PRBS) or a sine wave in a stationary state has been studied. This method has a skin effect according to the injected signal. This causes an error in the rotor resistance. On the other hand, the method of measuring the rotor time constant by applying and blocking the DC power supply and analyzing the motor information in the stationary state in the time and frequency domain is provided with a separate device or instrument such as a DC power supply or a frequency response analyzer. There is a downside to it. Moreover, since these equipments must be directly connected to the motor, there is a problem that it is difficult to put the inverter driving system into practical use in the industrial field directly connected to the motor.

In addition, the method of setting the controller gain as well as all the necessary motor constants in the initial stage by the self commissioning method has been studied. This method has the advantage of performing constant acquisition while the inverter is mounted, but there is a disadvantage that the error of the constant obtained initially accumulates due to the error of other constants obtained later. In particular, it is not the value acquired during operation. Occurs.

Therefore, most of the studies on motor constant estimation are focused on the online estimation method that directly compensates for the constant fluctuations that occur during operation. However, the existing online estimation method also requires the attachment of a separate sensor and is difficult to implement due to the large amount of calculation. There is a problem that precision is guaranteed only under operating conditions.

On the other hand, in an industrial site such as a steel making line that requires high-performance motor operation, an unknown motor may be connected to a mechanical load, and thus, unexpected operation of an induction motor for initializing or setting a constant of the motor may not be possible.

In this case, the conventional stationary off-line constant estimating method is affected by the error of other constants when estimating individual constants, and also has a problem in that sensitivity is inferior to the constant error. In addition, since the constant initial setting must be made at the stop state, there is a problem that the existing offline and online constant estimation methods that require rotation cannot be used. Therefore, a simple and accurate constant estimation method is required to perform the initial setting of the motor constant within a short time from the standstill state.

The problem to be solved by the present invention is to provide an induction motor by applying a current in which the induction motor connected to the load maintains the stop state and the AC component having a predetermined magnitude and frequency and the DC component having a predetermined magnitude are combined. It is to provide a method for estimating constants.

In order to achieve the above object, the induction motor constant estimation method according to the characteristics of the present invention includes (a) a first current including a d-axis current in which a high frequency AC component and a DC component of a first frequency and a q-axis current having a current value of 0 are zero. Applying a to an induction motor to measure a first equivalent impedance including a first equivalent resistance and a first equivalent inductance of the induction motor according to the first current; (b) a second equivalent of the induction motor according to the second current is applied to the induction motor by applying a second current including a d-axis current in which the high frequency AC component and the DC component of the second frequency and the q-axis current with a current value of 0 are applied. Measuring a second equivalent impedance comprising a resistance and a second equivalent inductance; And (c) estimating the stator resistance of the induction motor based on the relationship between the measured first and second equivalent resistances and the frequencies of the first and second currents.

Advantageously, the method may further comprise (d) estimating the rotor resistance, rotor leakage inductance, and stator leakage inductance according to the first current from the estimated stator resistance.

Preferably, the method comprises (e) applying a third current including the d-axis current of the low frequency alternating current component and the direct current component of the third frequency and the q-axis current having a current value of zero to the induction motor, Measuring a third equivalent inductance of the induction motor according to the invention; And (f) estimating the rotor leakage inductance according to the third current using the difference between the measured third equivalent inductance and the estimated stator leakage inductance.

Preferably, the method further comprises: (g) rated rotor resistance according to rated slip frequency and rated rotor frequency based on the relationship between the rotor leakage inductance according to the estimated first current and the rotor leakage inductance according to the estimated third current. The method may further include estimating the rotor leakage inductance according to the slip frequency.

Preferably, the step (g) comprises: (g1) extracting a first proportional coefficient K _{1 obtained} by multiplying the rotor leakage inductance according to the estimated first current and the double root of the first frequency; (g2) extracting a second proportional coefficient K _{2 obtained} by multiplying a constant constant by a depth of the rotor bar from the rotor leakage inductance according to the estimated third current; (g3) Resistance of the rotor bar to direct current and leakage inductance of rotor bar to direct current, based on the second proportional coefficient K ₂ , the first rotor resistance, and the first rotor leakage inductance. Extracting; And (g4) rotor resistance and rated slip according to the rated slip frequency based on the first proportional coefficient, the second proportional coefficient, resistance of the rotor bar to direct current, and leakage inductance of the rotor bar to direct current. Estimating the rotor leakage inductance with respect to frequency.

According to the present invention, when an induction motor including an inverter is connected to a load and independent operation is impossible, the induction motor constant can be estimated easily and accurately within a short time while maintaining the induction motor in a stopped state.

In addition, according to the present invention, the DC component current and the high frequency AC component current are applied together, and the current is set so as not to pass around zero, thereby eliminating the influence of the ZCC (Zero Current Clamping) effect. Differently, injection during operation does not affect operation performance.

In addition, according to the present invention, unlike the off-line constant setting method, the error of other constants is not involved in setting the individual constants, and has high accuracy since the frequency characteristic of the rotor is used.

In addition, according to the present invention, since it can be performed at a long distance through communication with the inverter, it is possible to reduce the motor constant estimation time.

1 is a view showing a core-shaped rotor bar.

The rotor of most squirrel cage induction motors is in the form of a deep-bar rotor as shown in FIG. d is the total depth of the rotor bar, w is the width of the slot, w _b is the width of the rotor bar, y is the height of any point on the rotor bar, and dy is the displacement of y (fine change) to be.

The total impedance of the rotor bar can be divided into a resistance part and a leakage inductance component, and the total voltage drop of the rotor can be expressed by Equation 1 below.

Vr: Voltage drop due to resistance component Vx: Voltage drop due to linkage flux

J: current density (A / m ² ) ρ: resistivity of rotor bar

Φ: chain link

In addition, Equation 1 may be summarized as in Equation 2 below.

w: width of the slot, where slot width (w) and rotor bar width (w _b ) are assumed to be the same

α:

In Equation 2, ρ / (wd) is equal to the resistance R _r0 per unit length of the rotor bar with respect to a direct current, and thus Equation 2 may be expressed as follows.

In Equation 3, since the total impedance Z is V / I, the total impedance Z of the rotor bar is expressed by dividing the resistance component R _r and the leakage reactance component X _lr as follows. .

이다(f:회전자 바에 흐르는 전류의 주파수, μ₀:진공투자율(4π*10^-7),ρ:회전자 바의 저항률). Where Z is the total impedance of the rotor, R _r is the rotor resistance component, X _lr is the rotor leakage reactance component, R _r0 is the rotor resistance to direct current, and d is the depth of the rotor bar. ,

(F: frequency of current flowing through the rotor bar, μ ₀ : vacuum permeability (4π * 10 ^-7 ), ρ: resistivity of the rotor bar)

On the other hand, since the leakage inductance L _lr0 of the rotor with respect to the DC current is expressed as Equation 5 below, Equation 4 is expressed as Equation 6 below.

Where X _lr0 is the rotor leakage reactance for direct current.

2 is a graph showing the change of the nonlinear terms included in [Equation 4] and [Equation 6] according to the frequency change.

In the graph of FIG. 2, the horizontal axis represents a change in αd, and the vertical axis represents a change of nonlinear terms included in Equations 4 and 6 above. Since α is as described above, αd eventually changes as the frequency changes as a function of frequency.

Referring to FIG. 2, when α d, which is a function of frequency, is 3 or more, all nonlinear terms included in Equation 6 converge to one. Therefore, when αd is 3 or more, the rotor resistance Rr and the rotor leakage reactance X _lr of Equation 6 become linear functions with respect to frequency. As will be described later, the rotor characteristics in this linear region (if αd is 3 or more) will be used to estimate the induction motor constant at the non-linear region (if αd is less than 3), ie at the rated slip frequency.

On the other hand, when αd is 3 or more, since the nonlinear terms of Equation 4 and Equation 6 converge, the following equation is also established.

(단, )

(only, )

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings.

3 is a flowchart of a method of estimating induction motor constants according to a preferred embodiment of the present invention.

First, a first current including a d-axis current obtained by adding a high frequency alternating current component I _h and a direct current component I _{ds at} a first frequency f ₁ = ω _h / (2π) and a q-axis current having a current value of 0. Is applied to an induction motor to measure a first equivalent impedance (Z _eq1 ) including a first equivalent resistance (R _eq1 ) and a first equivalent inductance (L _eq1 ) of the induction motor according to the first current (310).

In order to estimate the motor constant at standstill, the d-axis current (torque current: i ^e _ds ) and the q-axis current (magnetic flux current: i) in the synchronous coordinate system (the synchronous coordinate system and the stationary coordinate system are identical because the motor is stopped). ^e _qs ) is controlled as follows.

의 조건을 만족하도록 제1 주파수(f₁)를 결정하는 것이 바람직하다.If I _h increases at high frequencies, the difference in inductance between d-axis and q-axis increases, which causes reluctance torque to cause pulsation of current. Therefore, it is preferable to set I _h to about 1/10 of I _ds . Frequency depends on the capacity of the induction motor

It is preferable to determine the first frequency f ₁ to satisfy the condition of.

Since the magnitude of the direct current (I _ds ) is large enough for the magnitude of the alternating current (I _h ) and the direction of the current is always constant, all phase currents do not pass near zero current. Therefore, the zero current clamping (ZCC) effect does not occur, and the motor maintains the stop state because the q-axis current i ^e _qs is controlled to zero.

의 조건을 만족하는 주파수로 제1 주파수(f₁)를 결정하는 것이 바람직하다. αd는 회전자 바의 재질, 회전자 바의 깊이, 및 회전자에 흐르는 전류의 주파수에 관련된 함수이므로 회전자 바의 재질과 깊이를 알 수 있다면 최소 주입 주파수를 결정할 수 있다. On the other hand, to remove the nonlinearity of the rotor bar frequency as described above

It is preferable to determine the first frequency f ₁ as a frequency satisfying the condition of. αd is a function of the material of the rotor bar, the depth of the rotor bar, and the frequency of the current flowing through the rotor, so that the minimum injection frequency can be determined if the material and depth of the rotor bar are known.

을 만족하는 주입 주파수를 결정한다면 회전자 바의 재질에 대한 문제는 고려하지 않아도 된다. The material of the rotor bar is typically aluminum and copper, based on the aluminum rotor bar with a smaller αd value.

The problem with the material of the rotor bar does not have to be taken into account when determining the injection frequency that satisfies.

For induction motors less than 5 [kW], the depth of the rotor bar is less than 1.5 [cm], and for induction motors greater than 7 [kW], the depth of the rotor bar is more than 2 [cm]. As such, the larger the dose, the lower the minimum injection frequency. In the case of an induction motor of 5 [kW] or less, the most suitable injection frequency is preferably about 500 [Hz] in consideration of the minimum injection frequency, and about 200 [Hz] is preferable for an induction motor of 7 [kW] or more. . When the injection frequency is lowered, the value of αd becomes smaller, but even if the value of αd decreases to about 2, it can be seen that the error is about 5% and the error is not so large (see FIG. 2).

4 is an equivalent circuit when a current in which a DC component and a high frequency AC component are combined is injected into the d-axis.

의 형식을 가지는 전류이다. 주입 주파수가 충분히 크면 교류성분(I_hcosw_ht)의 전류는 대부분 회전자를 통해서만 흐르게 되고 직류성분(I_ds)의 전류는 여자회로(L_m)를 통해서 흐르게 된다. 유도전동기의 등가 임피던스(Z_eq)는 등가 저항(R_eq)과 등가 인덕턴스(L_eq)의 합으로서, 고주파에서의 등가 임피던스(Z_eq)를 구하는 방법은 페이져(Phasor)를 이용하는 방법 및 저역통과필터(LPF)를 이용한 신호처리 방법 등 다양한 방법이 알려져 있다. 본 발명은 다양한 방법에 의해 제1 전류에 따른 유도전동기의 제1 등가 저항(R_eq1)과 제1 등가 인덕턴 스(L_eq1)를 포함한 제1 등가 임피던스(Z_eq1)를 측정하게 된다.The current in Figure 4

A current of the form If the injection frequency is large enough, the current of the alternating current component (I _h cosw _h t) flows mostly through the rotor and the current of the direct current component (I _ds ) flows through the excitation circuit (L _m ). The equivalent impedance (Z _eq ) of the induction motor is the sum of the equivalent resistance (R _eq ) and the equivalent inductance (L _eq ), and the method of obtaining the equivalent impedance (Z _eq ) at high frequency is a method using a phaser and a low pass. Various methods are known, such as a signal processing method using a filter (LPF). According to the present invention, the first equivalent impedance Z _eq1 including the first equivalent resistance R _eq1 and the first equivalent inductance L _eq1 of the induction motor according to the first current is _measured by various methods.

3 will be described again.

의 조건을 만족하도록 제2 주파수를 결정하는 것이 바람직하다.In step 310, the induction motor constant estimating method according to the preferred embodiment of the present invention includes a sum of a high frequency AC component I _h and a DC component I _{ds of} a second frequency (f ₂ = ω _{h 2} / (2π)). A second current including an axial current and a q-axis current of zero current is applied to the induction motor, so that the second equivalent resistance R _eq2 and the second equivalent inductance L _eq2 of the induction motor according to the second current are applied. The second equivalent impedance Z _eq2 is measured (320). In this case, the second frequency must be different from the first frequency f1 used in step 310. At this time, as in step 310, I _h is preferably set to about 1/10 of I _ds , and the frequency depends on the capacity of the induction motor.

It is preferable to determine the second frequency to satisfy the condition of.

The stator resistance R _s of the induction motor based on the measured relationship between the first equivalent resistance R _eq and the second equivalent resistance R _eq2 and the frequencies f ₁ and f ₂ of the first and second currents. ) Is estimated (330).

Since the stator resistance (R _s ) is equal to the following Equation 8, the equivalent resistances R _eq1 and R _eq2 at frequencies f ₁ and f ₂ are represented by Equations 9 and 10, respectively. (R _eq = R _r + R _s ).

(여기서, K_r은 상수임)

Where K _r is a constant

Equation 9 and Equation 10 are combined to remove K _r and R _r0 , and the stator resistance R _s is summarized as in Equation 11 below. In the following Equation 11, the stator resistance R _s is calculated by substituting the equivalent resistances R _eq1 and R _eq2 and the frequencies f ₁ and f ₂ measured in steps 310 and 320.

The rotor resistance R _r , the rotor leakage inductance L _lr , and the stator leakage inductance L _ls according to the first current are estimated from the estimated stator resistance R _s (340).

5 is a flowchart illustrating step 340 of FIG. 3 in detail, and step 340 includes the following steps.

The rotor resistance R _r1 according to the first current is estimated using the difference between the measured first equivalent resistance R _eq1 and the estimated stator resistance R _s (510) (R _r1 ). = R _eq1 R _s ).

The rotor leakage inductance L _lr1 according to the first current is estimated based on the estimated rotor resistance R _r1 and the first frequency f ₁ (520). The rotor leakage inductance L _lr according to the first current is substituted by the rotor resistance R _r1 and the first frequency f ₁ according to the first current obtained in step 510 in Equation 12 below. Calculate

).The stator leakage inductance L _ls is estimated using the difference between the measured first equivalent inductance L _eq1 and the rotor leakage inductance L _lr1 according to the estimated first current (530) (

).

Among the estimated induction motor constants, stator resistance (R _s ) and stator leakage inductance (L _ls ) have a constant value regardless of frequency, and thus, the estimated stator resistance ( R _s ) and stator leakage inductance L _ls will be used.

A third current including a d-axis current in which the low frequency AC component I _h and the direct current component I _{ds at} the third frequency f ₃ and the q-axis current having a current value of 0 is applied to the induction motor, The third equivalent inductance L _eq3 of the induction motor according to the current is measured (350). At this time, as in step 310, the size of I _h is preferably set to about 1/10 of I _ds . On the other hand, the third frequency f ₃ is preferably determined such that αd is less than two.

식을 이용하면 상기 제3 전류에 따른 회전자 누설 인덕턴스(L_lr3)가 계산된다.Using the difference between the measured third equivalent inductance L _eq3 and the estimated stator leakage inductance L _ls , the rotor leakage inductance L _lr3 according to the third current is estimated (360). L _ls is a frequency-independent value that we already obtained in step 530,

Using the equation, the rotor leakage inductance L _lr3 according to the third current is calculated.

Based on the relationship between the rotor leakage inductance according to the estimated first current and the rotor leakage inductance according to the estimated third current, rotor resistance and rotor leakage inductance according to the rated slip frequency are estimated (370).

6 is a flowchart illustrating step 370 of FIG. 3 in detail, and step 370 includes the following steps.

The first proportional coefficient K _{1 obtained} by multiplying the rotor leakage inductance L _lr1 and the _{double root} of the first frequency f ₁ according to the estimated first current is extracted (610).

Since the first frequency f ₁ satisfies the condition of αd> 3, the nonlinear term converges to 1 in Equation 6 above. Therefore, the rotor leakage inductance L _lr1 according to the first current is represented by the following Equation 13 and Equation 14, and as a result, the first proportional coefficient K ₁ is represented by Equation 15 Is defined as].

Where L _lr0 is the rotor leakage inductance for direct current and αd is a function of frequency

임)(here,

being)

From the rotor leakage inductance L _lr3 according to the estimated third current, a second proportional coefficient K _{2 obtained} by multiplying a depth of the rotor bar by a constant is extracted (620).

Since the rotor leakage inductance L _lr3 according to the third current is low frequency (αd <2), the nonlinear term in [Equation 6] does not converge to 1 and exists. Therefore, L _lr3 is _represented by the following [Equation 16] and [Equation 17]. L _lr3 , f ₃ , K ₁ present in the right term of Equation 17 Since all the values are known, the value of αd is determined from Equation 17 (where αd is a value that changes with frequency as a function of frequency).

From Equation 18, a second proportional coefficient K ₂ is derived as in Equation 19, and since αd and f ₃ have already been determined in Equation 19, K ₂ is calculated therefrom.

On the basis of the extracted second proportional coefficient K ₂ , the rotor resistance R _r1 according to the estimated first current, and the rotor leakage inductance L _lr1 according to the estimated first current, The rotor resistance R _r0 and the rotor leakage inductance L _{lr 0} for the DC current are extracted (630).

The rotor resistance (R _r0 ) for DC current is calculated by [Equation 21] derived by Equation (20) below. The rotor leakage inductance (L _lr0 ) for DC current is Calculated by Equation 23 derived by Equation 22 (where α d is a value that changes with frequency as a function of frequency).

The extracted first proportional coefficient K ₁ , the extracted second proportional coefficient K ₂ , the rotor resistance R _{r0 for the} extracted DC current, and the rotor leakage inductance L _lr0 for the extracted DC current. Estimate the rotor resistance (R _{r _ slip} ) and the rotor leakage inductance (L _{lr_slip} ) according to the rated slip frequency on the basis of 640 (where α d is a value that changes with frequency as a function of frequency). ).

From Equation 6, Equations 24 and 25 are derived, respectively. As shown in [Equation 24] and [Equation 25], the rotor resistance (R _{r _ slip} ) according to the rated slip frequency and the rotor leakage inductance (L _{lr_slip} ) according to the rated slip frequency are expressed as a function of frequency. Therefore, not only the rated slip frequency but also the rotor resistance and the rotor leakage inductance according to an arbitrary value of frequency can be obtained.

로부터 구해질 수 있다(여기서, αd는 주파수에 대한 함수로서 주파수에 따라 변경되는 값임).Also, since the stator leakage inductance is a constant value that does not change with frequency, the value obtained at high frequency and the value at slip frequency are the same. Therefore, stator transient inductance ( _{σL s_slip} ) at rated slip frequency

(Where αd is a value that changes with frequency as a function of frequency).

In the rotor constant estimation method according to the present invention, it can be seen that the variation of the rotor constant with respect to the total rotor frequency is different. This has the effect of knowing the frequency characteristics of the rotor constant when injecting a separate high frequency into the rotor circuit, such as high frequency voltage injection sensorless control. In addition, since the frequency characteristics of the rotor are used, all processes can be performed in a state where the motor to which the load is connected is stopped. In addition, the present invention can be performed at a long distance through communication with the inverter, thus contributing to the reduction of time in setting the motor constant.

7 is a block diagram of an induction motor control apparatus according to another embodiment of the present invention.

In general vector control, current control is performed by converting an AC component of a stationary coordinate system from a synchronous coordinate system to a DC component. However, in case of controlling the high frequency current by using the current type of current controller, since the controller itself is involved in the delay factor, the control performance is lowered. Actual current is reduced in magnitude and significantly delayed in phase compared to the command current. Increasing the bandwidth to increase control performance will generate vibration and noise, which will reduce the stability of the system.

Referring to FIG. 7, the induction motor controller according to another embodiment of the present invention includes a first current controller 415, a second current controller 445, a first converter 460, and a first inverse converter 425. The second transform unit 475, the second inverse transform unit 450, the band pass filter (BPF, 465), the band stop filter (BSF, 480), the global pass filter 470, the first subtractor 410, and And a subtractor 440, an adder 420, and a signal processor 485.

By adjusting the d-axis current of the command currents I ^* _dqs , I ^e _dqh ^* input to the first subtractor 410 and the second subtractor 440, a current in which a DC component and an AC component are combined is finally applied to the induction motor. can do. At this time, the frequency of the AC component is the control angle θ _h provided to the second transform unit 475 for converting from the two-phase stationary coordinate system to the synchronous coordinate system and the second inverse transform unit 450 for inverting the two-phase stationary coordinate system from the synchronous coordinate system. Adjusted by). On the other hand, the q-axis currents of the command currents I ^* _dqs and _Ie ^* _dqh are all set to 0 so that the induction motor is stopped (the high frequency synchronous coordinate system is expressed by the synchronous coordinate system).

The first converter 460 includes a current sensing unit (not shown) and a phase converter (not shown). The three-phase current output from the induction motor 490 is detected by the current sensing unit. The three-phase current is converted from the three-phase to the two-phase stationary coordinate system (abc to dq axis) by the phase converter, and finally I _dqs is output.

I _dqs output from the first converter 460 Only the d-axis current is input to the middle band pass filter BPF 465 to extract only the AC component, and all of the dq-axis current is input to the band stop filter BSF 480 to output only the DC component. In order to obtain the DC component by converting the AC component into the synchronous coordinate system, the d-axis and the q-axis must have the same magnitude and have a phase difference of 90 °. However, since the actual q-axis current is controlled to 0 and cannot be used for coordinate transformation, a virtual q-axis current is generated and used for coordinate transformation by delaying the phase of the d-axis current by 90 ° using an all pass filter (APF, 470). .

The first current controller 415, which is a PI controller, controls the current in which the DC component _extracted from the band blocking filter (BSF, 480) and I ^* _dqs are combined to output V ^* _dqs . In addition, the second current controller 445, which is a PI controller, is configured independently of the first current controller 415 to extract the AC component (high frequency) extracted using the band pass filter (BPF) 465 and the global pass filter 470. Component is converted into a synchronous coordinate system to make a DC component and then controlled. When the output of the second current controller 445 of the AC component is inversely converted into the two-phase stationary coordinate system through the second inverse transform unit 450 and added to the output of the first current controller 415 of the DC component, the current of the AC component is consequently added. It can be controlled in the form of current of direct current component.

In addition, the q-axis voltage output of the second current controller 445 that appears during the inverse conversion is not added to the final voltage output, so that the actual q-axis current can be controlled to 0 only in the first current controller 415.

The signal processor 485 receives V ^* _dh , V ^* _dqs , and I _dqs and calculates the same, thereby performing an operation related to induction motor constant estimation.

The induction motor control apparatus independently configures a current controller for controlling an AC component and a DC component, so that the control performance is greatly improved compared to a conventional current controller, and it is possible to control the high frequency current more stably.

The method may also be embodied in computer readable code on a computer readable recording medium. The computer-readable recording medium includes all the recording devices in which data read by the computer is stored. Examples of computer-readable recording media include ROM, RAM, CD-ROM, magnetic tape, optical data storage, and the like, and also include a carrier wave (for example, transmission over the Internet). The computer readable recording medium can also be distributed over network coupled computer systems so that the computer readable code is stored and executed in a distributed fashion.

The technical scope of the present invention described above through the embodiments is not limited to the above-described embodiments, and various modifications and changes may be made without departing from the spirit and scope of the present invention. It is evident to those who have knowledge. Therefore, such modifications or variations will have to be belong to the scope of the invention described in the claims of the present invention.

1 is a view showing a deep ball cage rotor.

2 is a graph showing the change of the nonlinear terms included in [Equation 5] and [Equation 7] according to the frequency change.

3 is a flowchart of a method of estimating induction motor constants according to a preferred embodiment of the present invention.

4 is an equivalent circuit when a current in which a DC component and a high frequency AC component are combined is injected into the d-axis.

5 is a flowchart illustrating step 340 of FIG. 3 in more detail.

6 is a flowchart illustrating step 570 of FIG. 3 in more detail.

7 is a block diagram of an induction motor control apparatus according to another embodiment of the present invention.

Claims (10)

(a) applying a first current including a d-axis current in which a high frequency alternating current component and a direct current component of a first frequency and a q-axis current having a current value of 0 is applied to an induction motor, whereby a first current of the induction motor according to the first current is applied. Measuring a first equivalent impedance comprising an equivalent resistance and a first equivalent inductance; (b) applying a second current including a d-axis current in which a high frequency AC component and a direct current component of a second frequency and a q-axis current having a current value of 0 are applied to the induction motor, whereby a second of the induction motor according to the second current is applied. Measuring a second equivalent impedance comprising an equivalent resistance and a second equivalent inductance; And (c) estimating the stator resistance of the induction motor based on the relationship between the measured first and second equivalent resistances and the frequencies of the first and second currents. Way. The method of claim 1 wherein the estimated stator resistance is Where Rs is the estimated stator resistance, R _eq1 is the first equivalent resistance, R _eq2 is the second equivalent resistance, f ₁ is the first frequency, and f ₂ is the second frequency. A method for estimating a conductive synchronous constant, characterized in that calculated by the formula. The method of claim 1, (d) estimating the rotor resistance, the rotor leakage inductance, and the stator leakage inductance according to the first current from the estimated stator resistance.  The method of claim 3, wherein step (d) (d1) estimating the rotor resistance according to the first current using the difference between the measured first equivalent resistance and the estimated stator resistance; (d2) estimating rotor leakage inductance according to the first current based on the estimated rotor resistance and the first frequency; And (d3) estimating stator leakage inductance using the difference between the measured first equivalent inductance and the rotor leakage inductance according to the estimated first current. The method of claim 2, (e) applying a third current including a d-axis current in which a low frequency alternating current component and a direct current component of a third frequency and a q-axis current having a current value of 0 are applied to the induction motor, whereby a third of the induction motor according to the third current is applied. Measuring an equivalent inductance; And (f) estimating the rotor leakage inductance according to the third current using the difference between the measured third equivalent inductance and the estimated stator leakage inductance. . The method of claim 5, (g) Estimating the rotor resistance and the rotor leakage inductance according to the rated slip frequency based on the relationship between the rotor leakage inductance according to the estimated first current and the rotor leakage inductance according to the estimated third current Induction motor constant estimation method further comprising the step of. According to claim 6, wherein (g) step, (g1) extracting a first proportional coefficient K _{1 obtained} by multiplying the rotor leakage inductance according to the estimated first current by the double root of the first frequency; (g2) extracting a second proportional coefficient K _{2 obtained} by multiplying a constant constant by a depth of the rotor bar from the rotor leakage inductance according to the estimated third current; (g3) based on the second proportional coefficient K ₂ , the rotor resistance R _r1 according to the estimated first current, and the rotor leakage inductance L _lr1 according to the estimated first current Extracting the rotor resistance against the DC current and the rotor leakage inductance for the DC current; And (g4) Rotor resistance and rated slip frequency according to the rated slip frequency based on the first proportional coefficient, the second proportional coefficient, the rotor resistance to the DC current, and the rotor leakage inductance for the DC current. Induction motor constant estimation method comprising the step of estimating the rotor leakage inductance according to. The method of claim 7, wherein The first proportional coefficient is

Where K ₁ is the first proportional coefficient, L _lr1 is the first rotor leakage inductance, f ₁ is the first frequency, and The second proportional coefficient is

(here,

And μ ₀ is the vacuum permeability (4π * 10 ⁻⁷ ), ρ is the resistivity of the rotor bar, and d is the depth of the rotor bar. The method of claim 7, wherein The rotor resistance according to the rated slip frequency is

(Where R _{r _ slip} is the rotor resistance according to the rated slip frequency, K ₂ is the second proportional coefficient, R _r0 is the rotor resistance to DC current, and f _slip is the rated slip frequency) Is estimated by Rotor leakage inductance according to the rated slip frequency

(Where L _{lr _ slip} is the rotor leakage inductance according to the rated slip frequency, K ₂ is the second proportional coefficient, L _lr0 is the rotor leakage inductance for DC current, and f _slip is the rated slip frequency) Induction motor constant estimation method, characterized in that estimated by the equation. The method of claim 6, (h) estimating the stator transient inductance by summing the rotor leakage inductance and the estimated stator leakage inductance according to the estimated rated slip frequency.

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