KR20080077845A - Apparatus for receiving a signal and method for receiving a signal - Google Patents

Abstract

A signal receiving device and a signal receiving method are provided to obtain synchronization of a reception signal accurately in spite of the presence of a frequency offset. A signal including the first noise signal, among perpendicular noise signals is received(S10). The first noise signal is generated according to a generation rule of the perpendicular noise signals(S20). Values obtained by phase-shifting conjugate complex numbers of the first noise modulated according to a carrier frequency offset and values obtained by correlating reception signals including the first noise are added into a sum, and synchronization of the reception signals are obtained from a peak value of the sum(S30).

Description

Apparatus for receiving a signal and method for receiving a signal}

1 is a flow chart showing an embodiment of a signal receiving method according to the present invention

2 is a flowchart illustrating another embodiment of a signal receiving method according to the present invention;

Figure 3 is a flow chart showing another embodiment of a signal receiving method according to the present invention

4 is a configuration diagram showing a brief configuration of a transmitting apparatus of the DMB-T;

4 is a structural diagram of a frame having a guard interval of 1/9 as an example of a transmission signal frame of the DMB-T;

5 is a view showing an embodiment of a transmission signal receiving apparatus of a TDS-OFDM scheme;

6 is a view showing an embodiment of a DMB-T receiving apparatus;

7 schematically illustrates an example of measuring a frequency offset using a PN sequence in a DMB-T receiving apparatus;

8 illustrates a correlation magnitude between PN sequences according to frequency offsets.

9 is a diagram showing the magnitude of the correlation between two PN sequences according to the spacing ε of the PN sequences generated by the carrier frequency offset.

FIG. 10 is a diagram illustrating a result of summing correlation values calculated in FIG. 9 with respect to a carrier frequency offset. FIG.

11 is a view schematically showing the concept of a signal receiving apparatus and a signal receiving method according to the present invention

12A to 12D show the results of the calculation according to FIG. 11 according to a carrier frequency offset.

FIG. 13 is a diagram illustrating a frame sync period of a DMB-T to explain another embodiment of a signal receiving method according to the present invention; FIG.

14 is a diagram illustrating a signal calculation unit for calculating the frequency offset described in FIG. 13.

15 is a view showing an embodiment of a signal receiving apparatus according to the present invention

16 is a view showing an embodiment of a signal receiving apparatus according to the present invention

<Explanation of symbols in the main part of the drawing>

10: channel encoding unit 15: TPS generation unit

20: modulator

30: reverse DFT section 40: PN generation section

50: multiplexer 60: SRRC unit

70: RF transmitter 110: tuner

120: automatic gain control unit 130: A / D converter

140: phase separator 145: multiplier

150: resampler 160: SRRC unit

170: frame synchronization unit 171: PN correlation unit

172: signal acquisition unit 174: signal tracking unit

177: automatic frequency control unit 179: multiplier

180,182: DFT section 190: equalizer

260: PN generation unit 290: Frequency offset detection unit

310: temporary storage unit 320: correlation unit

330 coefficient calculation unit

410: multiplier 420: noise generator

430: signal calculator 435: offset calculator

440: peak detection unit 445: reference value storage unit

450: converter 460: position calculator

470: frequency offset estimation

The present invention relates to a signal receiving method and a signal receiving apparatus, and more particularly, to provide a signal receiving method and a signal receiving apparatus capable of receiving a transmission signal including a pseudo noise signal orthogonal to each other.

With the development of digital communications, broadcast or communication systems have emerged that transmit and receive using digital signals. In digital communication, noise signals having orthogonality orthogonal to each other may be used to code and spread a transmission signal, or may be used as a pilot signal of a transmission signal to easily obtain synchronization of a reception signal. A typical noise signal used for this purpose is a pseudo noise (PN) signal, and in addition, a quadrature noise signal can be generated. Such a noise signal may be used as a training signal in consideration of orthogonal characteristics.

When receiving a signal including a noise signal having an orthogonal phase, the noise signal generated by the receiver is correlated with the received signal in consideration of the characteristics of the orthogonal noise signal, and the noise signal included in the received signal is correlated. You can find it. However, when another noise signal is included from the receiving device or when another noise signal is included during transmission, there is a problem that the orthogonality of the noise signal inserted in the transmission signal is broken and synchronization cannot be easily obtained. For example, if a frequency offset occurs when the signal receiver of the signal receiver converts the received signal to an intermediate frequency, the orthogonality of the noise signal included in the received signal may be destroyed by the frequency offset. Then, there is a problem in that the synchronization of the received signal cannot be accurately obtained in the subsequent processing of the received signal.

The present invention is to solve the above problems and to provide a signal receiving method and a signal receiving apparatus that can more easily and accurately process a received signal when receiving a signal using a orthogonal noise signal.

Another object of the present invention is to provide a signal receiving method and a signal receiving apparatus capable of accurately obtaining a synchronization of a received signal even when there is a frequency offset.

Still another object of the present invention is to provide a signal receiving method and a signal receiving apparatus capable of accurately calculating and compensating a frequency offset even when a frequency offset occurs in a received signal.

In order to achieve the above object, the present invention comprises the steps of: receiving a signal including a first noise signal of orthogonal noise signals, generating a first noise signal according to the generation principle of the orthogonal noise signals; And calculating the sum of the values of the phase complex values of the first noise modulated according to the carrier frequency offset and the values correlated with the received signals including the first noise, respectively, and calculating the sum of the sum values. It provides a signal receiving method comprising the step of obtaining the synchronization of the received signal from a peak value.

In another aspect, the present invention provides a method comprising: receiving a signal including a first noise signal among orthogonal noise signals, generating a first noise signal according to a generation principle of the orthogonal noise signals, and modulating according to a carrier frequency offset Calculating the sum of the values of the complex conjugates of the first noise, respectively, of which phase-shifted values are correlated with the received signal including the first noise, and the received signal from the peak value of the sum value. Acquiring a synchronization signal; accumulating a value obtained by multiplying a received signal including the first noise signal by a conjugate complex number of the generated first noise signal and a signal modulated by a carrier frequency offset; When the synchronization is obtained in step, the phase shift of the received signal according to the frequency offset from the accumulated result is determined. Output, and using the calculated phase shift value to provide a signal reception method comprising the step of compensating the carrier frequency offset.

In yet another aspect, the present invention provides a method for receiving a signal including a first noise signal among orthogonal noise signals, generating a first noise signal according to a generation principle of the orthogonal noise signals, and according to a carrier frequency offset. Calculating a sum of the values of the phase-conjugated complex numbers of the modulated first noise and values correlated with the received signal including the first noise, respectively, and receiving the received value from the peak value of the sum. Acquiring a synchronization of a signal, converting the calculation result signals into a frequency domain in the synchronization acquiring step, and calculating a carrier frequency offset from a position shift of a known signal among the result signals of the frequency domain; And compensating for the received signal including the first noise signal using the calculated offset. It provides a method of receiving a signal comprising a.

In another aspect, the present invention provides a noise generating unit for generating a conjugate complex number of a first noise signal among orthogonal noise signals, values of the conjugate complex numbers of the first noise modulated according to a carrier frequency offset, respectively, 1 a signal calculator for calculating the sum of the values correlated to the received signal including noise, a peak detector for detecting peak values of the results calculated by the signal operator, and the results calculated by the signal operator in the frequency domain. A conversion unit for converting, a position calculating unit for calculating a change in the position of a frequency domain of a known signal among the signals converted by the converting unit, and a carrier frequency offset from the position change of the known signal calculated by the position calculating unit A frequency offset estimator for estimating and outputting a frequency and a frequency estimated by the frequency offset estimator Using the offset provides a signal receiving apparatus including a multiplier for compensating the reception signal.

In another aspect, the present invention provides a noise generating unit for generating a conjugate complex number of a first noise signal among orthogonal noise signals, values of the conjugate complex numbers of the first noise modulated according to a carrier frequency offset, respectively, 1 a signal calculator for calculating the sum of the values correlated to the received signal including noise, a peak detector for detecting peak values of the results calculated by the signal operator, and the results calculated by the signal operator in the frequency domain. A converting unit for converting, a position calculating unit for calculating a change in the position of a frequency domain of a known signal among the signals converted by the converting unit, and a carrier frequency offset from the change in the position of the known signal calculated by the position calculating unit A frequency offset estimator for estimating and outputting a frequency and a frequency estimated by the frequency offset estimator Using the offset provides a signal receiving apparatus including a multiplier for compensating the reception signal.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, embodiments of the present invention that can be specifically realized are described with reference to the accompanying drawings.

1 is a flowchart illustrating an embodiment of a signal receiving method according to the present invention. An embodiment of a signal reception method according to the present invention will be described with reference to FIG. 1.

First, a signal including a first noise signal among orthogonal noise signals is received (S10). In operation S20, the first noise signal is generated according to a principle of generating orthogonal noise signals.

Compute the sum of the values of the conjugate complex numbers of the first noise modulated according to the carrier frequency offset and the values correlated with the received signal including the first noise, respectively, and the peak of the sum value. A synchronization of the received signal is obtained from the value (S30). Step S30 will be described in detail with reference to FIG.

Another embodiment of the present invention can remove the carrier frequency offset included in the received signal using the embodiment illustrated in FIG. Another embodiment of the invention is disclosed in FIGS. 2 and 3.

2 is a flowchart illustrating an embodiment of a signal receiving method according to the present invention. Hereinafter, an embodiment of a signal receiving method according to the present invention will be described with reference to FIG. 2.

A signal including a first noise signal among orthogonal noise signals is received (S110). The first noise signal is generated according to the generation principle of the orthogonal noise signals (S120).

Compute the sum of the values of the conjugate complex numbers of the first noise modulated according to the carrier frequency offset and the values correlated with the received signal including the first noise, respectively, and peak of the sum value. The synchronization of the received signal is obtained from the value S130. Step S130 is described in detail with reference to FIG.

In operation S140, the operation result signals are converted into a frequency domain.

The carrier frequency offset is calculated from a position shift of a known signal among the result signals of the frequency domain, and the received signal including the first noise signal is compensated using the calculated offset (S150). 15 is an embodiment of a signal receiving apparatus according to the present invention, but discloses the steps S 140 and S 150 in detail.

3 is a flowchart illustrating still another embodiment of a signal receiving method according to the present invention. Referring to Figure 3 another embodiment of a signal receiving method according to the present invention will be described.

A signal including a first noise signal among orthogonal noise signals is received (S210). The first noise signal is generated according to the generation principle of the orthogonal noise signals (S220).

A first complex signal including at least one of a conjugate complex number of the generated first noise signal and a signal complex modulated by a carrier frequency offset and simultaneously modulated by the conjugate complex number to be phase-shifted; Compute with the received signal and obtain the synchronization of the received signal from the peak value of the operation result (S230).

The received signal including the first noise signal and the complex number of the generated first noise signal are accumulated by multiplying the signals modulated by the carrier frequency offset (S240). When the synchronization is acquired in the synchronization acquisition step, the phase shift of the received signal according to the frequency offset is calculated from the accumulated result, and the carrier frequency offset is compensated using the calculated phase shift value (S250). 16 is an embodiment of a signal receiving apparatus according to the present invention, but discloses the steps S 240 and S 250 in detail.

In order to more easily understand the present invention, a noise signal having a plurality of phases and a signal orthogonal to each phase is an example of a noise signal, and the first noise signal is a signal of a specific phase of the PN sequence.

A case of transmitting and receiving a signal according to a TDS-OFDM (TDS-OFDM) system in which a PN sequence is used as a training signal will be described as an example.

This system is used in terrestrial digital television (hereinafter referred to as terrestrial DTV) broadcasting for China, and is used as a broadcasting standard called Terrestrial Digital Multimedia / Television Broadcasting (DMB-T).

Data transmitted after being modulated at the transmitting end of the TDS-OFDM is applied with an Inverse Discrete Fourier Transform (IDFT), as used in a cyclic prefix OFDM (CP-OFDM) scheme. However, the guard interval is used as a training signal by inserting pseudonoise (PN) instead of using a cyclic prefix (CP). Such a method can reduce overhead in transmitting a room transmission call, improve channel usage efficiency, and improve performance of a synchronization unit and a channel estimator of a broadcast signal receiver.

In order to easily describe the present invention, a transmission / reception system of the DMB-T will be described.

4 is an exemplary diagram illustrating a brief structure of a DMB-T transmitting apparatus. Referring to Figure 4 describes the operation of the transmitter of the DMB-T as follows.

The channel encoder 10 outputs a bitstream encoded by the channel in order to detect an error at the receiving end.

The modulator 20 receives an encoded bit stream and a transmission parameter signal output by the TPS generation unit 15, and modulates the received signal with quadrature amplitude modulation such as four, sixteen, or sixty-four values. Modulation is performed using quadrature amplitude modulation (QAM).

The inverse DFT unit 30 modulates a signal modulated by the OFDM scheme in the frequency domain into an OFDM signal in the time domain. In general, the DMB-T method converts a frequency domain signal of 3780 points of transmission data into a time domain signal.

The PN generator 40 generates a PN sequence to be used as a training signal of a broadcast signal to be transmitted.

The multiplexer 50 distributes the generated PN sequence and the OFDM signal converted by the inverse DFT unit 30 in a time domain, and multiplexes the same.

The filter unit 60 outputs the bandwidth of the multiplexed DMB-T signal by limiting it. The filter unit 60 may use a SRRO (Square Root Rasied Cosine) filter. In this case, 0.05 may be used as a roll-off factor (α) used for bandwidth limitation.

The RF transmitter 70 up-converts the output signal of which the bandwidth is limited to an RF (Radio Frequency) transmission band of frequency fc to transmit a broadcast signal.

FIG. 5 illustrates a structure of a frame in which a guard interval is 1/9 of a data interval among frames of a signal transmitted by the DMB-T transmitting apparatus of FIG. 4. The frame structure of the transmission signal will be described with reference to FIG. 5.

The frame includes frame sync and frame body.

The frame body is a place where data to be transmitted is a DFT block to which a Discrete Fourier Transform (DFT) is applied to be transformed into a frequency domain, and the DFT block generally includes 3780 stream data.

The frame sink includes a PN sequence, and the PN sequence used for the frame sink may use a sequence having an order of 8 (m = 8). When m = 8, 255 different sequences can be generated, and each sequence is extended to a preamble and a postamble to be used for a guard interval.

The preamble and the postamble are repetition intervals of the PN sequence for cyclic extension of the PN sequence.

Of the 255 PN sequences of the frame sync, the first 115 PNs of the PN sequence are added as a postamble to the end of the 255 PN sequence, and the last 50 PNs of the PN sequence are added as preambles before the 255 PN sequence. To expand.

The polynomial of the PN sequence is P (x) = x ⁸ + x ⁶ + x ⁵ + x + 1, and the phase changes from 0 to 254 according to the initial state of the PN sequence.

If the guard interval is 1/9 long, a preamble and postamble may be added to the 255 PN sequences before and after the frame sync including 420 data. In other words, 420 data, which is 1/9 of 3780 data of the DFT block, are used for frame sync. One OFDM frame may include a frame sink including 420 data and a frame body including 3780 data. The frame body may include 3744 symbols of data, 32 symbols of system information, and 4 symbols of a carrier mode. System information and carrier mode are modulated with BPSK.

The structure of the data frame of the DMB-T may vary depending on the protection period, and the number of data distributed in each frame may also be distributed differently. In addition, the guard interval may be defined as 1/4 length or 1/9 length, in addition to that, 1/6 length guard interval may be used. That is, the guard interval may include data of 420, 595, 945, etc., and the frame structure of the transmission signal forming the system may vary according to the length of the guard interval.

6 is a diagram illustrating an embodiment of a DMB-T receiving apparatus. The operation of the DMB-T receiving apparatus illustrated in FIG. 6 is as follows.

DMB-T receiver The tuner 110 of the receiver converts the signal of the RF transmission band into a base band signal and outputs it.

The automatic gain controller (AGC) 120 normalizes and outputs the power of the output signal.

The analog to digital converter 130 converts the output signal into an analog signal and outputs the digital signal.

The phase splitter 140 divides an inphase component signal (hereinafter referred to as I signal) and a quadrature component signal (hereinafter referred to as Q signal) from the signal output from the A / D converter 130. Print separately.

The automatic frequency control (FCC) unit 177 compensates the estimated frequency error of the separated I and Q signals, and the filter unit 160 filters the bandwidth of the received signal. Do this.

The frame synchronization unit may include an signal acquisition unit 172, a signal tracking unit 174, and an AFC unit 177.

First, the AFC unit 177 may calculate a frequency error of the received signal and compensate the frequency error of the received signal by calculating the product of the received signal and the signal from which the frequency error is calculated through the multiplier 145.

The acquisition unit 172 synchronizes the PN sequence sent by the transmitter. The signal tracking unit 174 compensates for the symbol error using the captured PN sequence.

The frame synchronization units described above all use correlation results of the PN correlator 171.

The data estimated as a result of the frame synchronization unit is transformed into the frequency domain through the fast fourier transform (FFT) process in the DFT units 180 and 182, and the channel is compensated through the equalizer 190 to the channel decoder (not shown). Is output.

FIG. 7 is a diagram schematically illustrating an example of measuring a frequency offset using a PN sequence in a DMB-T receiving apparatus. A method of measuring the frequency offset will be described with reference to FIG. 7 as follows.

First, the PN generator 260 of the signal receiver may arbitrarily generate a first PN sequence sample among PN sequences used as a training signal of a transmission signal. In the example of FIG. 7, the PN sequence generated by the generation polynomial has a length of 255.

The correlator 270a of the correlator 270 correlates the generated PN sequence with 255 length sample data of the received signal. The correlator 270a moves and correlates along the input data. In FIG. 7, the correlator 270a represents a moving window as a moving window.

The frequency offset detection unit 290 divides the values correlated by the correlator 270a into a plurality of sub blocks, correlates correlation values among the plurality of sub blocks, and correlates the values that correlate the sub blocks. The carrier frequency offset may be measured as a peak. When the subblock is one frame section of the DMB-T system, one frequency offset can be measured in one frame on the assumption that the frequency offset is constant in one frame.

In general, when the size of the carrier frequency offset is small, the above-described frequency offset estimation method can be used. In the case of the DMB-T system, a frequency offset of -8 kHz to +8 kHz generated by the tuner can be measured and compensated by the frequency offset estimation method described above.

However, when the magnitude of the frequency offset is large, the orthogonality between the PN sequence included in the transmission signal and the PN sequence generated in the receiving apparatus is destroyed, and the magnitude of the correlation value is reduced. In addition, the method of removing the frequency offset by squaring the PN sequence modulated with BPSK as described above has a severe deviation of the estimated frequency offset, and requires a separate multiplier as many as the number of correlators. Also, if the synchronization is not properly obtained, the frequency offset cannot be accurately removed. Accordingly, an embodiment in which a frequency offset can be removed by accurately acquiring synchronization of a received signal will be described below.

8 is a diagram illustrating a correlation magnitude between PN sequences according to frequency offsets. The horizontal axis represents the magnitude of the frequency offset and the vertical axis represents the magnitude of the correlation value. As illustrated in FIG. 8, when the size of the frequency offset increases, the magnitude of the correlation value between the PN sequence of the received signal and the PN sequence generated by the reception device may decrease, and the correlation value may have a null position. In FIG. 8, an arrow indicates a position of a null position, and an interval between PN sequences correlated by a carrier frequency offset may be represented by the following equation. That is, in Equation 1, the carrier frequency offset becomes a normalized value according to the interval of symbols.

Herein, N is the length of the frame, which is 3780 for DMB-T, and M represents the length of the PN sequence. Alternatively, when fc is a carrier frequency offset (in HZ) and fs is a sub-symbol spacing, ε of Equation 1 may be expressed as fc / fs. As shown in FIG. 8, the correlation value has a null position similar to a sync function according to a frequency offset.

In order to know the effect of the frequency offset on the correlation of the PN sequence, the sequence r (n) when the frequency offset exists in the PN sequence C (n) included in the received signal is as follows.

When the receiver receives and correlates the PN sequence shown in Equation 2, the integral value P (m) of the correlation value is as follows.

* Represents a conjugate complex number of C (n).

When there is a carrier frequency offset in the PN sequence of the received signal, the PN sequence and the PN sequence generated by the receiving apparatus may be expressed by the following equation when the synchronization is identical (that is, m = 0 in Equation 3).

Α in Equation 4 is an arbitrary constant according to the square of C (n), and Equation 4 may be expressed as follows.

According to Equation 5, if the received signal has a frequency offset, the PN sequence of the received signal and the PN sequence generated by the receiving device have a value of the correlation value (P (0)) even if they are synchronized with each other (m = 0). Has a different value if there is no frequency offset.

For example, the magnitude of P (0) changes according to the frequency offset [epsilon] as follows.

Therefore, if there is a frequency offset, the magnitude of the correlation value of each PN sequence may change, so it may not be possible to determine whether synchronization is consistent only by the magnitude of the correlation peak value of the PN sequence. According to Equation 6, the position of the null position according to the frequency offset illustrated in FIG. 8 may be obtained. That is, since the correlation value is close to zero in the magnitude of the frequency offset at which the null position occurs in FIG. 8, it is difficult to determine depending on the correlation value of the PN sequence under such conditions. In addition, although the PN sequence included in the signal transmitted according to the transmission / reception system and the PN sequence generated by the receiving device receiving the signal are generated by the same generation principle, the peak of the correlation does not occur as described above.

9 shows the magnitude of the correlation of two PN sequences according to the interval ε of the PN sequence generated by the carrier frequency offset. Here, the two PN sequences refer to the PN sequence included in the received signal and the PN sequence generated by the receiving device. In FIG. 9, the correlation size of two PN sequences shows a null position at a predetermined position.

FIG. 10 illustrates a result of adding correlation values calculated in FIG. 9 with respect to a carrier frequency offset. In the case of two PN sequences correlated in FIG. 10, it is assumed that a lag between two sequences is 0 if there is no influence due to a frequency offset. According to the result of FIG. 10, when a carrier frequency offset occurs, correlation values of two PN sequences have a similar shape to a sync function as shown in FIG. 10. And, according to FIG. 10, even though a carrier frequency offset occurs, the sum of correlation values for PN sequences is almost similar. Thus, the value of the sum of the correlations of the two PN sequences that all possible frequency offsets have is of approximately similar magnitude. Since the sum of the sum is a large value of a scale similar to the correlation peak value, even when a frequency offset occurs, the synchronization of the received signal can be obtained.

11 is a view schematically showing the concept of a signal receiving apparatus and a signal receiving method according to the present invention. The received signal includes a first noise signal, and the first noise signal has a correlation peak in the same signal as itself and has a small correlation value in a signal different from itself. The example of FIG. 11 is a signal according to the DMB-T system, and a received signal includes a PN sequence which is a first noise signal. In the signal according to the DMB-T system, a 255-length PN sequence and its PN sequence extend back and forth in a frame sync section.

When receiving and processing this signal, a PN sequence P '(n) identical to the PN sequence included in the received signal is generated to obtain synchronization even when a carrier frequency offset occurs.

By the possible phase change values of the PN sequence (e ^ {± jπp (M-1) / M}; (b) term), the conjugate complex of the PN sequence can be phase shifted, and the carrier frequency offset is normalized to the symbol interval. Values (e ^ {± j2πpn) / M}; In (c) term, the complex conjugate of the PN sequence is modulated according to the carrier frequency offset.

Therefore, the PN sequence P '(n), the conjugate complex number P * (n) (a) of the generated PN sequence P' (n), and the possible phase change values e ^ of the PN sequence. {± jπp (M−1) / M}; (b) term) and carrier frequency offset values that normalize the symbol interval (e ^ {± j2πpn) / M}; (c) correlate and sum the conjugate complex numbers modulated by term), respectively.

In order to facilitate this calculation, a product of the PN sequence P '(n) and its conjugate complex number P * (n) is a common factor, and the common complex conjugate of the PN sequence has phase change values. And, at the same time, multiply the sum of the values that the conjugate complex numbers are modulated to have frequency offsets.

That is, the example of the operation described above is the same principle as that of calculating (a + ab + abc + ...) to a (1 + b + bc + ...). Where a is a common factor, and the sum of parentheses (1 + b + bc + ..) is the sum of the values that cause the complex complex to be modulated by the carrier frequency offset and the values that cause the complex complex to phase change. do.

12A to 12D are diagrams illustrating the result of the calculation according to FIG. 11 according to a carrier frequency offset.

First, FIG. 12A shows a result of correlating a PN sequence included in a received signal with a generated PN sequence having a frequency offset of 0 kHz along a time axis (horizontal axis). 12A is an ideal method when there is no frequency offset in a conventionally used method. However, when the frequency offset occurs, it is difficult to obtain synchronization and compensate for the frequency offset because the result is not shown in FIG. 12A.

FIG. 12B illustrates a result of correlating a PN sequence included in a received signal and a PN sequence generated by a receiving apparatus in a manner described in FIG. 11 with a frequency offset of 0 kHz.

FIG. 12C shows a result of correlating a PN sequence included in a received signal with a PN sequence generated by a receiving apparatus in a manner described in FIG. 11 with a frequency offset of 80 kHz, and FIG. 43D shows a frequency offset of 160 kHz, see FIG. The result of the calculation in the manner described above is shown. 12B to 12D, even when the frequency offset occurs, since the magnitude of the correlation peak value is almost similar, synchronization can be obtained at the portion where the correlation peak value occurs.

FIG. 13 is a diagram illustrating a frame sync period of a DMB-T in order to explain another embodiment of a signal receiving method according to the present invention. According to the embodiment of the method of receiving a signal according to the present invention, even if there is a frequency offset, synchronization can be obtained, and the frequency offset can be estimated and compensated. A method of compensating for the frequency offset will be described with reference to the following equation and FIG. 13.

R (l) obtained by summing the noise signal r (k) included in the received signal and the result of correlating the noise signal r * (kl) generated by the receiving apparatus and delayed by l by the received signal and the frequency offset. ) May be expressed as follows (where l denotes a distance on a signal delayed by a carrier frequency offset from synchronization of a PN sequence generated from a PN sequence of a received signal).

In Equation 7, since η (k) means a noise component of 0, the sum operation of Equation 7 may be expressed as Equation 8. In Equation 8, the value of the term including η (k) is 0. to be. ;

Accordingly, the phase of R (l) in Equation 8 is as follows. ]

When the phase difference 2επl / N group is within 180 degrees, the range of ε is as follows.

For example, if the delay length l is 32, the maximum ε is 118.125 according to Equation 10, in which case the frequency offset (f _offset ) is 235.25 (KHz) (= 2 kHz (sub-symbol spacing) × ε). . Therefore, when the length l delayed by the frequency offset is known, the magnitude of the frequency offset can be calculated and compensated for.

FIG. 14 is a diagram illustrating a signal calculator that calculates a frequency offset described with reference to FIG. 13. An example of the signal calculator of FIG. 14 may be included in a signal receiving apparatus according to the present invention disclosed in FIGS. 15 and 16. In the example of FIG. 14, (a) shows an embodiment of the signal calculator.

The operation as shown in FIG. 13 may require a large number of multiplexers. Therefore, in order to calculate this efficiently, the embodiment according to the present invention may have a device structure as shown in FIG.

The operation of modulating the PN sequence according to the phase and the possible frequency offset according to the value normalized at the symbol interval, and then calculating the modulated PN sequence with the received signal and summing again, is finally performed in (b) of FIG. The same coefficients as shown are computed in the PN sequence through a finite impulse response (FIR) filter.

When the received signal is correlated, the shift operation and the addition operation are performed by the first filter unit 331 of the coefficient calculator 330 to perform a correlation operation without adding a multiplier. Since coefficients used as filter coefficients are symmetrical as shown in FIG. (C) in FIG. 14 is a diagram showing the FIR coefficients in the time domain in the frequency domain.

14 illustrates a temporary storage unit 310, a correlation unit 320, and a coefficient operation unit 330.

The temporary storage unit 310 stores the received signal and outputs the received signal. The temporary storage unit 310 may store the received signal in the form of first-in first-out (fifo). The temporary storage unit 310 stores and outputs received data in an arbitrary length. The example of FIG. 14 illustrates an example of storing and outputting a received signal in units of 10 bits. The correlator 320 stores a pair complex part of the generated noise signal having a length of 1 bit in a storage unit therein. The noise signal stored in the temporary storage unit 310 and the one-bit noise signal stored in the storage unit in the correlation unit 320 are correlated with each other.

The first calculator 331 of the coefficient calculator 330 stores coefficients of a function as shown in FIG. 14B. As illustrated in FIG. 11, the coefficient of FIG. 14B is generated by the sum of values excluding a product of P ′ (n) and P * (n), which are common factors. The sum of the signals obtained by modulating the conjugate complex number of the first noise signal into phase shifted values and frequency offset values has a form as shown in FIG.

Therefore, the coefficient calculating unit 330 multiplies the correlation value output by the correlator 320 in the form of a coefficient as shown in FIG. In this case, since the coefficient of FIG. 14B is constantly changed with time, the product of the coefficient can be converted into a shift operation and a sum operation.

The storage unit 333 of the coefficient calculator 330 stores a value calculated through a shift operation and a sum operation, and the second calculator 335 adds the values stored in the storage unit 333. To print. Therefore, the operation disclosed in FIG. 11 can be easily performed. In FIG. 14, the storage unit 333 is represented by fifo for temporarily storing 10-bit length data.

15 is a view showing an embodiment of a signal receiving apparatus according to the present invention. An embodiment of a signal receiving apparatus according to the present invention will be described with reference to FIG. 46 as follows.

An embodiment of the signal receiving apparatus according to the present invention includes a multiplier 410, a noise generator 420, a signal calculator 430, a peak detector 440, a converter 450, a position calculator 460 and A frequency offset estimator 470 may be included.

First, the noise generator 420 generates a noise signal by the same generation principle as that of the noise signal included in the received signal.

The signal calculator 430 may correlate the received signal with the noise signal generated by the noise generator 420. The signal operator 430 may include a temporary storage unit 431, a correlation unit 433, and a coefficient operator 435.

The temporary storage unit 431 stores the received signal, and the correlation unit 433 may correlate the signal stored in the temporary storage unit 431 with the signal generated by the noise generator 420. The coefficient calculating unit 435 calculates the filter coefficient calculated using the phase change value of the noise signal included in the received signal and the values obtained by normalizing the frequency offset in symbol units with the result of correlation by the correlation unit 433. . Signal operation of the signal calculator 430 has been described with reference to FIG. 14.

The peak calculator 440 may calculate the position of the peak value by using an appropriate threshold value from the result calculated by the coefficient calculator 435.

The converter 450 may convert the result calculated by the signal calculator 430 into the frequency domain. The position calculator 460 calculates how far a known signal in a received signal such as a pilot signal is located from the original position of the frequency domain on the frame from the signal in the frequency domain output from the converter 450. can do. For example, in the case of the DMB-T system, the position calculator 460 determines how far the position on the frame of the transmission parameter signal or system information is from the original position when there is no frequency offset. You can judge.

The frequency offset estimator 470 may estimate the frequency offset from the position of the signal calculated by the position calculator 460. The frequency offset estimator 470 converts the frequency offset in the frequency domain into a phase change in the time domain and outputs the result to the multiplier 410.

The multiplier 410 may remove the frequency offset of the received signal including the noise signal by multiplying the received signal by a phase shift value on the time axis according to the frequency offset estimated by the frequency offset estimator 470.

16 is a view showing an embodiment of a signal receiving apparatus according to the present invention. An embodiment of a signal receiving apparatus according to the present invention will be described with reference to FIG. 47 as follows.

16 illustrates an embodiment of a signal receiving apparatus according to the present invention, a multiplier 410, a noise generator 420, a signal calculator 430, an offset calculator 435, a peak detector 440, and a reference value storage unit. 445 may be included.

The noise generator 420 generates a noise signal by the same generation principle as that of the noise signal included in the received signal.

The signal calculator 430 correlates the received signal with the noise signal generated by the noise generator 420, and calculates the signal using a phase change value of the noise signal included in the received signal and values obtained by normalizing the frequency offset in units of symbols. The filter coefficient can be calculated with the result of the correlation unit 433 being correlated. The signal operation unit 430 receives a signal of at least one of a complex complex number of the noise signal generated through the above described operation and a signal modulated such that the complex complex number is modulated by a carrier frequency offset and simultaneously modulated by the complex complex number. You can output the signal and the result of calculation

The signal calculator 430 may include a temporary storage 431, a correlator 433, and a coefficient calculator 435. Details of each component have already been described in FIG. 15.

The peak detector 440 may detect peak values of the results output from the signal calculator 430 using a threshold value. The peak detection unit 440 outputs a signal for referring to the reference value storage unit 445 when a peak value is detected in the results calculated by the signal operation unit 440 (the enable signal in FIG. 16).

The offset calculation unit 435 may multiply the received signal by the complex conjugate of the received signal shifted by the frequency offset, and output the sum of the product according to the noise signal (in the case of DMB-T, the result of the product is PN sequences included in the frame sync are summed for PN sequences excluding postamble and preamble). The offset calculator 435 may perform the operation described in Equation 8.

The reference value storage unit 445 stores the phase value of the calculation result output by the offset calculation unit 435. When the reference value storage unit 445 receives the reference signal from the peak detection unit 440, the reference value storage unit 445 outputs a phase value corresponding to the calculation result output by the offset calculation unit 435. The reference value storage unit 445 may store the phase values according to the offset calculation result in a table form according to Equation (9). The reference value storage unit 445 outputs a phase value according to the result calculated by the offset calculation unit 435. This phase value may be a phase in the time domain according to the frequency offset included in the received signal.

The multiplier 410 may remove the carrier frequency offset included in the received signal by multiplying a phase output from the reference value storage 445 by a received signal including noise.

It is easy for a person skilled in the art to change or modify the present invention from the present specification. Therefore, although an embodiment of the present invention has been described above clearly, various modifications thereof should be made without departing from the spirit and the scope of the present invention.

The effects of the signal reception method and the signal reception device according to the present invention described above are as follows. According to the signal receiving method and the signal receiving apparatus according to the present invention, when the signal is received using the orthogonal noise signal, the noise signal can be processed more easily. In addition, even when there is a frequency offset, synchronization of a received signal can be accurately obtained. According to the signal receiving method and the signal receiving apparatus according to the present invention, even when a frequency offset occurs, the frequency offset can be accurately calculated and compensated.

Claims (19)

Receiving a signal comprising a first noise signal among orthogonal noise signals; Generating a first noise signal in accordance with the generation principle of the orthogonal noise signals; And Compute the sum of the values of the conjugate complex numbers of the first noise modulated according to the carrier frequency offset and the values correlated with the received signal including the first noise, respectively, and peak of the sum value. And obtaining the synchronization of the received signal from the value of. The method of claim 1, In the acquiring of the synchronization, the peak value is added by combining the values of the conjugate complex number modulated by the carrier frequency offset and the values of the conjugate complex number phase shift, and the conjugate complex number and the result of the summation. And calculating and multiplying the received signal including the first noise signal. The method of claim 2, The multiplication operation is performed by a shift operation and a sum operation of the sum result. The method of claim 1, And the noise signal is a pseudonoise sequence. The method of claim 1 The signal receiving method in the step of receiving the signal is a signal modulated according to the Time Domain Synchronous OFDM (TDS-OFDM). Receiving a signal comprising a first noise signal among orthogonal noise signals; Generating a first noise signal in accordance with the generation principle of the orthogonal noise signals; Compute the sum of the values of the conjugate complex numbers of the first noise modulated according to the carrier frequency offset and the values correlated with the received signal including the first noise, respectively, and peak of the sum value. Acquiring synchronization of the received signal from the value of; Accumulating values of a received signal including the first noise signal and a product of a complex complex number of the generated first noise signal multiplied by a carrier frequency offset; And Calculating a phase shift of a received signal according to a frequency offset from the accumulated result and compensating the carrier frequency offset by using the calculated phase shift value when synchronization is obtained in the synchronization acquisition step; Receiving method. The method of claim 6, And the noise signal is a pseudonoise sequence. The method of claim 6, And calculating the calculated phase shift as an arctangent function value of the accumulated result. Receiving a signal comprising a first noise signal among orthogonal noise signals; Generating a first noise signal in accordance with the generation principle of the orthogonal noise signals; Compute the sum of the values of the conjugate complex numbers of the first noise modulated according to the carrier frequency offset and the values correlated with the received signal including the first noise, respectively, and peak of the sum value. Acquiring synchronization of the received signal from the value of; Converting the operation result signals into a frequency domain in the synchronization acquisition step; And Calculating a carrier frequency offset from a position shift of a known signal among the result signals of the frequency domain, and compensating a received signal including the first noise signal using the calculated offset; How to receive the signal. The method of claim 9, The known signal is a transmission parameter signal (TPS). A noise generator for generating a complex conjugate of the first noise signal among orthogonal noise signals; A signal calculating unit calculating a sum of values of phase-conjugated complex conjugates of the first noise modulated according to a carrier frequency offset and values respectively correlated with a received signal including the first noise; A peak detector for detecting peak values of the results of the signal calculation unit; An offset calculation unit for multiplying a received signal including the noise signals and a signal complex number of the received signal modulated according to a carrier frequency offset, and outputting a result of adding up the multiplying results; A reference value storage unit for storing phase shift values of the signal with respect to the frequency offset; And And a multiplier for multiplying the received signal by a phase shift value read out from the reference value storage unit according to the output of the offset calculator. The method of claim 11, The signal operation unit may include a temporary storage unit that stores a received signal including the first noise signal; A correlator for correlating the conjugate complex number of the first noise signal generated by the noise generator with the received signal; And Multiply the values by which the conjugate complex number is modulated by the carrier frequency offset and the values by which the conjugate complex number is modulated by phase shift, and multiply the multiplied values according to the modulated values as coefficient values And a coefficient calculator for multiplying the coefficient values with the results output by the correlation unit. The method of claim 12, And the multiplication operation calculated by the coefficient calculating unit performs the coefficient value by a shift operation and a sum operation. The method of claim 11, And a phase shift value read out from the reference value storage by the offset calculator is an arctangent function value for the output of the offset calculator. The method of claim 11, And the noise signal is a pseudonoise sequence. A noise generator for generating a complex conjugate of the first noise signal among orthogonal noise signals; A signal calculating unit calculating a sum of values of phase-conjugated complex conjugates of the first noise modulated according to a carrier frequency offset and values respectively correlated with a received signal including the first noise; A peak detector for detecting peak values of the results of the signal calculation unit; A converter for converting the results calculated by the signal operator into a frequency domain; A position calculator for calculating a change in a position of a frequency domain of a known signal among the signals converted by the converter; A frequency offset estimator for estimating and outputting a carrier frequency offset from the position change of a known signal calculated by the position calculator; And And a multiplier for compensating for the received signal using the frequency offset estimated by the frequency offset estimator. The method of claim 16, The known signal is a transmission parameter signal (TPS). The method of claim 16, The signal operation unit may include a temporary storage unit that stores a received signal including the first noise signal; A correlator for correlating the conjugate complex number of the first noise signal generated by the noise generator with the received signal; And Multiply the values by which the conjugate complex number is modulated by the carrier frequency offset and the values by which the conjugate complex number is modulated by phase shift, and multiply the multiplied values according to the modulated values as coefficient values And a coefficient calculator for multiplying the coefficient values with the results output by the correlation unit. The method of claim 16, And the noise signal is a pseudonoise sequence.

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