KR20070119915A - Equalizing method and apparatus - Google Patents

Abstract

A channel equalizer and an equalizing method are provided to calculate a precise phase noise value of a reception signal and to equalize the same. A PN(Pseudo Noise) convolution unit(530) calculates a data signal of a time domain and a channel impulse response from a reception signal. The second calculating unit(570) compensates an estimated phase noise of the data signal. The first and second FFT(Fast Fourier Transform) units(572,574) convert an output signal of the second calculating unit(570) and the channel impulse response calculated by the PN convolution unit(530) into signals of a frequency domain. A phase noise estimating unit(580) estimates a phase noise from the data signal of the frequency domain and a channel transfer function and output them to the second calculating unit(570). A distortion compensating unit(590) receives the data signal of the frequency domain and the channel transfer function and compensates a distorted channel included in the data signal.

Description

Channel Equalizer and Equalization Method {Equalizing Method and Apparatus}

1 is a configuration diagram illustrating an example of a TDS-OFDM transmitter;

2 is a structural diagram showing a frame structure of a signal having a guard interval of 1/9 among TDS-OFDM transmission signals

3 is a block diagram of an embodiment of a broadcast receiving apparatus that may include a channel equalizer according to the present invention;

4 is a distribution diagram showing a distribution in the frequency domain of four sets of pilot signals

5 is a block diagram illustrating a channel equalizer for compensating for phase noise of a received signal using estimated phase noise according to an embodiment of the present invention.

6 is a block diagram showing the configuration of the phase noise estimation unit according to an embodiment of the present invention

7 is a signal diagram showing the actual phase noise of the received signal and the phase noise estimated by the phase noise estimator according to the present invention;

8A illustrates a constellation of a received signal including phase noise.

8B is a constellation diagram after compensating for the received signal of FIG. 8A by the channel equalizer according to the present invention.

<Explanation of symbols in the main part of the drawing>

500: frame timing recovery unit 510: channel estimation

520: PN (Pseudo Noise) generation unit 530: PN convolution unit

540: first operation unit 550: overlap unit

560: padding section 570: second operation section

572: First FFT Unit 574: Second FFT Unit

580: phase noise estimation unit 590: distortion compensation unit

610: signal detector 620: signal extractor

630: reference signal generation unit 640: third operation unit

650: first IFFT unit 660: second IFFT unit

670: phase calculation unit

The present invention relates to a channel equalizer and an equalization method, and more particularly, to a channel equalizer and an equalization method capable of solving phase rotation and intercarrier interference (hereinafter, referred to as ICI).

In China, a new standard for terrestrial digital television (hereinafter terrestrial DTV) broadcasting has been proposed. The proposal relates to a broadcast standard called Terrestrial Digital Multimedia / Television Broadcasting (DMB-T). In DMB-T, a new modulation scheme called Time Domain Synchronous OFDM (hereinafter referred to as TDS-OFDM) is used.

A signal transmitted after being modulated at the transmitting end of the TDS-OFDM is applied with an Inverse Discrete Fourier Transform (IDFT) as in the cyclic prefix OFDM (CP-OFDM) scheme.

However, Pseudo Noise (PN) is inserted in the guard interval instead of CP and used as a training signal.

The above-described method can reduce overhead when transmitting a broadcast signal, improve channel usage efficiency, and improve performance of a synchronizer and a channel estimator of a broadcast signal receiver.

1 is a configuration diagram showing an example of a DMB-T transmitting apparatus. Referring to FIG. 1, the operation of the transmitter of the DMB-T will be described.

The channel encoder 110 outputs a bitstream in which data is encoded so that an error can be detected at the receiving end.

The TPS generation unit 120 includes a TPS (transmission variable parameter signal) including channel coding or modulation information such as a frame group number, a forward error correction (FEC) code error ratio, and a time-deinterleaver mode. TPS) generates and outputs the data.

The modulator 130 receives the encoded bitstream and the TPS data output from the TPS generator 120, and modulates quadrature amplitude modulation such as 4-value (4QAM), 16-value (16QAM), or 64-value (64QAM). Quadrature Amplitude Modulation (QAM).

The IDFT unit 140 modulates the signal modulated by the OFDM scheme in the frequency domain into an OFDM signal in the time domain. In the DMB-T scheme, the IDFT unit 140 may simultaneously convert frequency domain signals for 3780 points of transmission data into time domain signals.

The PN generator 150 generates a PN sequence to be used as a training signal of a broadcast signal to be transmitted.

The multiplexer 160 distributes the generated PN sequence and the OFDM signal converted by the IDFT unit 140 in the time domain, and multiplexes and outputs the same.

And SRRC (Square Root Rasied Cosine: SRRC) unit 170 outputs by limiting the bandwidth of the multiplexed DMB-T signal. In general, the roll-off factor (α) used for the bandwidth limit is 0.05.

The RF transmitter 180 up-converts the output signal with the limited bandwidth to an RF transmission band of a predetermined carrier frequency fc and transmits a broadcast signal.

2 shows the structure of a frame of a signal having a guard interval of 1/9 among signals transmitted by the TDS-OFDM method. A transmission frame structure having a guard interval of 1/9 will be described with reference to FIG. 2.

The frame has frame sync and frame body. The frame body is a place where data to be transmitted is a DFT block to which a discrete fourier transform (DFT) is applied, and the DFT block generally includes 3780 stream data. The frame body, which is a data section, includes transmission parameter signals (TPS) in four sections.

The frame sync consists of a PN sequence, and the PN sequence used for the frame sync may use a sequence having an order of 8 (m = 8). When m = 8, 255 different sequences can be generated, and the sequences can be extended to preambles and postambles for use in guard intervals.

Accordingly, the preamble and the postamble may be a repetition period of the PN sequence for cyclic extension (cyclic extension) of the PN sequence.

Of the 255 PN sequences of the frame sync, the first 115 PN sequences of the PN sequence are added as a postamble to the end of the 255 PN sequences, and the last 50 PNs of the PN sequence are preambles before the 255 PN sequences. Can be added and extended.

The polynomial of the PN sequence is P (x) = x ⁸ + x ⁶ + x ⁵ + x + 1, and the generated phase varies from 0 to 254 according to the initial state of the PN sequence.

When the guard interval is 1/9, the preamble and the postamble may be added to the 255 PN sequences before and after to configure a frame sink including 420 data. In other words, 420 data, which is 1/9 of 3780 data of the DFT block, may be used for frame sync. One OFDM frame may be composed of a frame sink of 420 data and a frame body of 3780 data.

The structure of the data frame may vary depending on the protection period, and the number of data distributed in each frame may also be distributed differently.

In addition, the protective section may be defined as 1/4 or 1/9, in addition to the 1/6 protective section may be used, and thus, the length of the protective section may be formed differently depending on the standard forming the system.

When receiving the transmission signal modulated by the TDS-OFDM scheme, the receiver performs a process of compensating for frequency error and compensating for sampling error. The channel compensation is performed in the frequency domain with respect to the error compensated signal, and the process of decoding the compensated received signal is performed.

In addition, since the phase noise introduced by the oscillator may cause distortion of the signal during the modulation and demodulation of the signal, the receiving end compensates for the phase noise.

The oscillator of the transmitter / receiver stage changes in phase due to thermal noise and noise caused by an external power supply and control current, and the phase undulation and frequency undulation generate phase noise in the signal.

In particular, in the case of a multi-carrier method such as OFDM, since the sub-carrier spacing is very narrow, it is more sensitive to phase noise than the single carrier method. In systems that require high data rates at a given bandwidth and use complex modulation schemes to achieve these rates, it is more important to eliminate phase noise.

An object of the present invention is to provide a channel equalization device and an equalization method that can solve phase rotation and ICI due to phase noise without increasing the cost.

A feature of the channel equalizer according to the present invention for solving the above problems is a signal calculating unit for calculating a data signal and a channel impulse response in the time domain from a received signal, and noise for compensating the estimated phase noise in the data signal. A compensator, a first converter converting the output signal of the noise compensator and the channel impulse response calculated by the signal calculator into a signal in the frequency domain, and a data signal and a channel transfer function in the frequency domain output from the first converter A phase noise estimator for estimating phase noise from the phase noise estimator and outputting it to a noise compensator, and a distortion compensation for compensating for channel distortion included in the data signal by receiving a data signal and a channel transfer function in a frequency domain output from the first converter To include wealth.

The phase noise estimator extracts a signal detector for detecting a pilot signal from a data signal of a frequency domain received by the first converter, and extracts a section for estimating phase noise from the pilot signal detected by the signal detector, based on the signal of the interval. A signal extracting unit converting a band signal, a reference pilot signal generating a reference pilot signal and multiplying with a channel transfer function received by the first converting unit, and calculating the signal extracted from the signal extracting unit and the reference signal calculating unit It is preferable to include a phase noise calculation section for converting the received signal into a signal in the time domain to calculate phase noise.

A feature of the channel equalization method according to the present invention is a signal calculation step of calculating a time domain data signal and a channel impulse response from a received signal, and compensating the data signal of the time domain calculated in the signal calculation step with estimated phase noise. A frequency domain conversion step of converting a signal compensated in the phase noise compensation step, a phase noise compensation step and a channel impulse response calculated in the signal calculation step into a signal in a frequency domain, and data of the frequency domain converted in the frequency domain conversion step A phase noise estimation step of estimating phase noise from a signal and a channel transfer function, and a distortion compensation step of compensating channel distortions included in the data signal from the channel signal and the channel signal in the frequency domain transformed in the frequency domain conversion step; It is in

The phase noise estimating step includes a signal detection step of detecting a pilot signal from a data signal of the frequency domain transformed in the frequency domain conversion step, and extracting a section necessary for phase noise estimation from the pilot signal detected in the signal detection step. A signal extraction step of converting a signal into a baseband signal, a reference signal calculation step of calculating a reference pilot signal passing through a channel from a reference pilot signal generated at a receiver and a channel transfer function converted at a frequency domain conversion step, and the signal extraction step And a phase noise calculation step of converting the signal extracted from the signal and the signal calculated in the reference signal calculation step into a signal in the time domain to calculate phase noise.

Other objects, features and advantages of the present invention will become apparent from the following detailed description of embodiments taken in conjunction with the accompanying drawings.

In addition, the terms used in the present invention was selected as a general term widely used as possible now, but in certain cases, the term is arbitrarily selected by the applicant, in which case the meaning is described in detail in the corresponding description of the invention, It is to be clear that the present invention is to be understood as the meaning of terms rather than names.

The operation of the channel equalizer and the equalization method according to the present invention configured as described above will be described in detail with reference to the accompanying drawings.

3 is a block diagram of an embodiment of a broadcast receiving apparatus that may include a channel equalizer according to the present invention.

The broadcast receiving apparatus includes a tuner 300, an automatic gain control unit 310, an A / D converter 320, a phase separator 330, a multiplier 340, a resampler 350, and an SRRC unit 360. ), PN correlator 372, signal acquisition unit 374, signal tracking unit 376, automatic frequency control unit 378, channel estimation and PN removal unit 380, DFT unit 382, 384, and equalizer 390.

The tuner 300 converts a signal of a radio frequency (RF) transmission band into a base band signal and outputs the signal to an automatic gain controller (AGC) 310. The automatic gain control unit 310 normalizes the power of the output signal and outputs it to an A / D converter 320.

The A / D converter 320 converts the signal output from the automatic gain control unit 310 into an analog signal to a digital signal and outputs the converted signal.

The phase splitter 330 is an in-phase component signal (hereinafter referred to as I signal) and a quadrature component signal (hereinafter referred to as Q signal) from the signal output from the A / D converter 320. To print out.

When the resampler 350 samples and outputs the signal calculated by the multiplier 340, the SRRC unit 360 filters the signal output from the resampler 350. That is, the SRRC unit 360 performs a filter role of limiting the bandwidth of the signal as in the transmitting apparatus.

The frame synchronization unit may be roughly divided into three parts: an signal acquisition unit 374, a signal tracking unit 376, and an AFC unit 378. First, the AFC unit 378 calculates an estimated frequency error of the I signal and the Q signal separated from the received signal, and calculates the product of the received signal and the signal from which the frequency error is calculated by the multiplier 340. Frequency error can be compensated.

The signal acquisition unit 374 synchronizes the PN sequence transmitted by the transmitter. Finally, the signal tracking unit 376 compensates for the symbol error using the captured PN sequence.

In this case, the frame synchronizer uses a correlation result of the received signal and the PN sequence in the PN correlator 372.

The channel estimator and PN remover 380 calculates a signal of a data section in which the channel impulse response and the PN section are removed from the received signal.

The DFTs 382 and 384 convert a data signal and a channel impulse response (CIR) output as a result of the channel estimation and PN removal unit 380 into a frequency domain through Fourier transform, respectively. .

The equalizer 390 compensates and outputs a channel in the frequency domain. Channel compensation is inverse filtering of the estimated channel and is performed in the frequency domain. The data entered through the frame synchronizer generates a channel impulse response while passing through the channel estimation and PN remover 380. Therefore, the channel can be compensated in the frequency domain by DFTing the data and the channel impulse response signal, respectively.

Hereinafter, a concept of compensating for a channel by the channel equalizer and the equalization method according to the present invention will be described.

을 모델로 한다. 그 절대값 크기는 1이므로 위상에서의 잡음만이 포함된다. 상기 위상잡음 모델을 전제로 수신단에 수신된 신호 가운데 DFT(Discrete Fourier Transform)를 거치기 전의 프레임 바디(frame body) 구간 신호는 하기의 수학식 1과 같이 표시할 수 있다.Phase noise is one phase rotation

Is modeled after. Since the absolute magnitude is 1, only noise in phase is included. On the premise of the phase noise model, a signal of a frame body section before passing a Discrete Fourier Transform (DFT) among the signals received at the receiver may be expressed by Equation 1 below.

는 한 개의 위상회전, x(n)은 송신단에서 송신한 프레임 바디 구간의 신호를 의미한다.

은 제 n번째 샘플링 포인트에서의 위상잡음임을 나타낸다. 따라서 수신신호 r(n)은 시간영역에서 x(n)과 위상잡음에 의한 위상회전 값(

)을 곱셈한 값이 된다.In Equation 1, r (n) denotes a signal of a frame body section before passing through a DFT at a receiver,

Denotes one phase rotation, and x (n) denotes a signal of a frame body section transmitted by a transmitter.

Denotes phase noise at the nth sampling point. Therefore, the received signal r (n) is a phase rotation value due to x (n) and phase noise in the time domain.

) Is multiplied by.

The dispersion of phase noise is very small compared to the data signal of the frame. Therefore, the phase rotation value due to phase noise can be expressed by Equation 2 below.

If Equation 2 is substituted for Equation 1, it can be expressed as Equation 3 below.

When the signal of Equation 3 is DFTed and converted into a signal in the frequency domain, a common phase error (CPE) and an inter carrier interference (ICI) component exist as shown in Equation 4 below.

Y (k) is a value obtained by DFT of r (n) and represents a value in the frequency domain.

가 생긴다.When r = k for any k in Equation 4, a CPE is generated as shown in Equation 5 below as a result of the calculation later in Equation 4. That is, common phase rotation in one frame

Occurs.

where

where

가 된다. 즉 상기 수학식 5의 결과는 수신신호의 한 프레임 내에서 공통되는 위상회전을 나타낸다.Equation 5 is a value at the back of Equation 4 when r = k

Becomes In other words, the result of Equation 5 indicates a common phase rotation within one frame of the received signal.

When r ≠ k for any k in Equation 4, an ICI component is generated as shown in Equation 6 below as a result of the calculation later in Equation 4.

가 된다. 즉, 다른 부반송파(sub-carrier)들에 의한 간섭인 ICI가 송신단에서 송신한 신호와 함께 수신신호에 포함된 것을 나타낸다.Equation 6 is a value at the end of Equation 4 when r ≠ k

Becomes That is, ICI, which is interference by other sub-carriers, is included in the received signal together with the signal transmitted from the transmitter.

The CPE can be estimated by using an average of offsets of a pilot such as a conventional TPS. It is simple to calculate and convenient to implement, but only the CPE portion of the phase noise can be estimated and compensated for.

Therefore, by using the guard pilot period, the other subcarriers use the property to prevent the interference to the pilot.

As shown in Equation 7, the signal r (n) received at the receiving end is the signal x (n) sent from the transmitting end, the channel impulse response h (n), the white noise n '(n) and the phase noise φ. It is represented including (n).

The convolution of the signal x (n) transmitted from the transmitter and the channel impulse response h (n) represents the value of x (n) received at the receiver through a noiseless channel. In Equation 7, the second stage represents the value of the component in which the phase noise φ (n) is reflected, and the final stage represents that the white noise n '(n) is added.

The signal r (n) received at the receiver of Equation 7 undergoes a synchronization process and a channel estimation process in the time domain. After that, the FFT (Fast Fourier Transform) process is performed to convert the data section into the frequency domain. This can be expressed as Equation 8 below.

As a result of converting the frequency domain through the FFT of Equation 8, it can be seen that the signal transmitted from the transmitter, the convolutional form of the phase noise and the signal transmitted from the transmitter, and white noise are included.

4 shows a TPS signal in a frequency domain in a DMB-T system. 4 shows one frame body and shows that four sets of pilot signals are included in the frame body.

Since these TPS signals are subjected to deep fading while being transmitted through a channel, the power of four sets of TPS signals included in one frame is measured, and one set having the largest power is selected.

In order to estimate the amount of phase noise included in the received signal, the TPS signal having the largest power is extracted. At this time, in order to remove a section in which an interference occurs with a data section, only a necessary section of the TPS signal section having the largest power is filtered and extracted (Y _{TPS (i)} (k)).

The TPS signal extracted as described above is lowered to a base band to obtain a baseband signal in a frequency domain including phase noise. This is represented by Equation 9 below.

Since the information corresponding to the extracted TPS signal is known and the channel transfer function (CTF) estimated and converted by the channel estimator is known, the known TPS signal is multiplied by the channel transfer function, and V (k) (= H (k) TPS _i (k)).

Equation (9) is an equation representing a signal in the frequency domain of the TPS signal extracted by filtering only a necessary section. Y _{TPS (i)} (k) is a signal of the frequency domain of the TPS signal extracted by filtering only the necessary section at the receiver, TPS _i (k) is a signal at the transmitter corresponding to the pilot signal extracted by filtering only the necessary section at the receiver This is a signal in the frequency domain of the transmitted pilot signal.

V (k) is a signal corresponding to a pilot signal extracted by filtering only a necessary section at a receiver, and is a frequency domain signal when a pilot signal transmitted from a transmitter passes through a noiseless channel.

Phase noise can be calculated in the time domain. Therefore, the phase noise is calculated by converting to the time domain as shown in Equation 10 below.

The phase noise is a complex conjugate of a signal obtained by converting a pilot signal (Y _{TPS (i)} (k)) obtained from a receiver into a time domain and a signal converted from a V (k) into a time domain. can be obtained by multiplying the signal.

Alternatively, the estimated phase noise may be obtained by dividing the signal obtained by converting the TPS signal (Y _{TPS (i)} (k)) obtained by the receiver into the time domain of the V (k) signal.

5 is a block diagram illustrating a channel equalizer for compensating for phase noise of a received signal using estimated phase noise according to an embodiment of the present invention.

The channel equalizer of FIG. 5 includes a channel estimation and PN removal unit 380, a second operation unit 570, a first FFT unit 572, a second FFT unit 574, a phase noise estimation unit 580, and distortion compensation. Part 590 is included.

The channel estimation and PN removal unit 380 includes a frame timing recovery unit 500, a channel estimation unit 510, a PN (pseudo noise) generation unit 520, a PN convolution unit 530, and a first operation unit 540. , An overlap part 550, and a padding part 560.

The frame timing recovery unit 500 recovers and outputs the frame timing of the received signal received from the synchronization unit.

The channel estimator 510 calculates channel characteristics from the received signal transmitted from the synchronizer and outputs a channel impulse response of the received signal.

The PN generator 520 generates and outputs a PN sequence included in frame synchronization of a transmission signal using a predetermined PN initial value (PN number). The PN convolution unit 530 convolves and outputs the channel impulse response output from the channel estimator 510 to the PN sequence output by the PN generator 520.

The first operator 540 removes the PN sequence output by the PN convolution unit 530 from the received signal output from the frame timing recovery unit 500 and outputs a signal from which the PN sequence section is removed from the received signal.

The overlap unit 550 overlaps and outputs data using a periodic property of the Fourier operation in order to apply the Fourier operation to the frame body section from which the PN sequence output from the first operator 540 is removed.

The estimated phase noise should be compensated for the time domain signal. Therefore, the second operator 570 multiplies the complex signal of the estimated phase noise by the data signal calculated by the overlap unit 550 to compensate for the phase noise.

The first FFT unit 572 converts the data signal compensated by the second operation unit 570 into a signal in the frequency domain.

The padding unit 560 pads a section other than the channel impulse response to 0 to apply a Fourier operation to the channel impulse response output by the channel estimator 510. The second FFT unit 574 calculates a channel transfer function by converting the channel impulse response zero-padded from the padding unit 560 into the frequency domain.

The phase noise estimator 580 estimates phase noise included in the received signal by using the data signal and the channel transfer function, which are transformed into the frequency domain by the FFT units 572 and 574.

The distortion compensator 590 performs channel compensation on the received signal based on the signals calculated by the first FFT 572 and the second FFT 574.

6 is a diagram illustrating a configuration of a phase noise estimation unit according to an embodiment of the present invention.

The phase noise estimator 580 according to the present invention includes a signal detector 610, a signal extractor 620, a reference signal generator 630, a third operator 640, a first IFFT unit 650, and a second IFFT unit. 660, and a phase calculating unit 670.

The signal detector 610 detects the TPS signal having the largest power in the data signal converted and output from the signal in the frequency domain by the first FFT unit 572. One frame includes four sets of TPS signals, among which the highest power TPS signal is detected.

The signal extractor 620 extracts a section without interference from the TPS signal extracted by the signal detector 610 except for a section where interference occurs with a data section and converts the extracted signal into a baseband signal.

For example, in the distribution of pilot signals of FIG. 4, when the pilot signal set of TPS ₃ (k) has the largest power, the signal detector 610 detects the pilot signal set of TPS ₃ (k). In the pilot signal group of TPS ₃ (k), if the middle signal having the largest size and two signals adjacent to each other do not interfere with the data section, the signal extractor 620 performs the middle signal having the largest size. The signal and the two signals on both sides are extracted and converted into a baseband signal.

The signal extracted by the signal extractor 620 and converted into baseband is a TPS signal in a frequency domain including phase noise.

In order to estimate phase noise, a reference signal corresponding to the TPS signal of the received signal extracted by the signal extractor 620 must be known. The reference signal is a TPS signal received at the receiver and does not include phase noise. This is calculated from the channel impulse response and the convolution of the time-domain TPS signal transmitted by the transmitter.

The reference signal calculator that calculates the reference signal includes a reference signal generator 630 and a third operator 640.

The reference signal generator 630 receives the information about the TPS signal from the signal detector 610 and generates a reference TPS signal for calculating the reference signal. The generated reference TPS signal is a signal in the frequency domain.

The third operator 640 multiplies the reference TPS signal generated by the reference signal generator 630 with the channel transfer function converted by the second FFT unit 574 to calculate a result. The calculated signal is a TPS signal in the frequency domain received at the receiving end that does not include phase noise.

The phase noise calculator for calculating phase noise includes a first IFFT unit 650, a second IFFT unit 660, and a phase calculator 670.

The signals extracted or calculated by the signal extractor 620 and the reference signal calculator 630 and 640, respectively, are signals in the frequency domain.

Accordingly, the first IFFT unit 650 converts the signal extracted from the signal extractor 620 into a signal in the time domain, and the second IFFT unit 660 converts the signal calculated by the third operator 640 of the reference signal calculator. Convert to a signal in the time domain.

The phase noise is calculated by subtracting the phase value of the TPS signal not including the phase noise calculated by the second IFFT unit 660 from the phase value of the TPS signal including the phase noise calculated by the first IFFT unit 650. do.

To calculate this, the phase calculator 670 divides the signal calculated by the first IFFT unit 650 into the signal calculated by the second IFFT unit 660 and calculates a phase difference value of both signals.

Alternatively, the phase calculator 670 multiplies the complex conjugate signal of the signal calculated by the second IFFT unit 660 with the signal calculated by the first IFFT unit 660 to calculate a phase difference value of both signals.

In the channel equalizer and the equalization method, the FFT unit or the IFFT unit is used to perform DFT and IDFT operations in a short time, so that the embodiment may be implemented as a unit that performs DFT and IDFT, respectively.

7 is a signal diagram showing the actual phase noise of the received signal and the phase noise estimated by the phase noise estimator according to the present invention.

In order to experiment the performance of the estimation of phase noise using the estimated phase noise according to the present invention, the specification of a specific tuner was used. The tuner's phase noise PSD (Power Spectral Density) has a distribution of 1 kHz (-65 dBc / Hz), 10 kHz (-75 dBc / Hz), and 100 KHz (-96 dBc / Hz). 7 shows a state of estimating phase noise occurring in this case. In FIG. 7, phase noise estimation for a low frequency region of 2 KHz or less can be seen.

Normally occurring phase noise is present over the entire frequency band, but at higher frequencies. In FIG. 7, the actual phase noise in which the signals of the high frequency and the low frequency are mixed can be seen. The thick line above the actual phase noise is a result of estimating the phase noise from the embodiment according to the present invention. It can be seen that this is a result of estimating phase noise of a low frequency region among actual phase noises.

It can be seen from the phase noise estimation result of FIG. 7 that the estimated value is discontinuous, since the reference signal is extracted from one TPS signal among four TPS signals included in one frame to estimate phase noise. .

FIG. 8A is a constellation of a received signal including phase noise, and FIG. 8B is a constellation after compensating the received signal of FIG. 8A by a channel equalizer according to the present invention.

In FIG. 8A, the phases are rotated by the CPE and ICI phenomena generated when the phase noise occurs and the scattered points. In FIG. 8B, it can be confirmed that the estimated phase noise is compensated and the above phenomenon is reduced.

In estimating the phase noise of the received signal, the PN sequence or other pilot signals among the received signals can be calculated. However, the channel equalizer and the equalization method according to the present invention preferably use the TPS signal of the received signal.

As described above, the channel equalizer and the equalization method according to the present invention have an effect of calculating and equalizing accurate phase noise of a received signal. By using the equalizer and the equalization method, problems of phase rotation and ICI, which are problematic in a multi-carrier system, can be solved.

In addition, by using the equalizer and the equalization method, even if an expensive oscillator is not used, the phase noise of the received signal can be accurately compensated and the reception performance of the receiver can be improved.

Claims (5)

A signal calculation unit for calculating a data signal and a channel impulse response in the time domain from the received signal; A noise compensator for compensating the phase noise estimated in the data signal; A converter for converting an output signal of the noise compensator and a channel impulse response calculated by the signal calculator to a signal in a frequency domain; A phase noise estimator for estimating phase noise from a data signal and a channel transfer function of the frequency domain output from the converter and outputting the phase noise to the noise compensator; And And a distortion compensator for receiving the data signal and the channel transfer function of the frequency domain output from the converter to compensate for channel distortion included in the data signal. The method of claim 1, Phase noise estimator, A signal detector for detecting a pilot signal from the data signal in the frequency domain received by the converter; A signal extracting unit extracting a section required for phase noise estimation from the pilot signal detected by the signal detecting unit and converting a signal of the section into a baseband signal; A reference signal calculator for generating a reference pilot signal and multiplying the channel transfer function received by the converter; And And a phase noise calculator for converting the signal extracted by the signal extractor and the signal calculated by the reference signal calculator into a signal in a time domain to calculate phase noise. The method of claim 2, And the pilot signal is a transmitter parameter signal (TPS). A signal calculation step of calculating a data signal and a channel impulse response in the time domain from the received signal; A phase noise compensation step of compensating the data signal of the time domain calculated in the signal calculation step with an estimated phase noise; A frequency domain conversion step of converting the signal compensated in the phase noise compensation step and the channel impulse response calculated in the signal calculation step into a signal in a frequency domain; A phase noise estimation step of estimating phase noise from the data signal and the channel transfer function of the frequency domain transformed in the frequency domain conversion step; And And a distortion compensation step of compensating for channel distortion included in the data signal from the data signal and the channel transfer function of the frequency domain transformed in the frequency domain conversion step. The method of claim 4, wherein Phase noise estimation step, A signal detection step of detecting a pilot signal from the data signal of the frequency domain converted in the frequency domain conversion step; A signal extraction step of extracting a section required for phase noise estimation from the pilot signal detected in the signal detection step and converting a signal of the section into a baseband signal; A reference signal calculation step of calculating a reference pilot signal passing through the channel from the reference pilot signal generated by the receiver and the channel transfer function converted in the frequency domain conversion step; And And a phase noise calculation step of converting the signal extracted in the signal extraction step and the signal calculated in the reference signal calculation step into a signal in a time domain to calculate phase noise.

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