KR20080036508A - Method for transmitting data using cyclic delay diversity - Google Patents

Abstract

A method for transmitting data is provided to reduce the complexity of a signal transmitting and receiving device and improve the efficiency of communication by using a generalized cyclic delay diversity operation. A transmission symbol corresponding to each of sub-carriers is spatially processed in a frequency region. The spatially processed transmission signal is converted into a time region. The converted transmission signal is multiplied by a predetermined weight value. Cyclic delay diversity is applied to the transmission signal according to each of antennas. A first pilot symbol corresponding to each antenna is added to the spatially processed transmission signal.

Description

Method for transmitting data using Cyclic Delay Diversity

The present invention relates to a method for transmitting data using cyclic delay in a multi-antenna system using a plurality of subcarriers.

Recently, the demand for wireless communication services is rapidly increasing due to the generalization of information communication services, the appearance of various multimedia services, and the emergence of high quality services. To cope with this actively, the capacity of the communication system must be increased.In order to increase the communication capacity in the wireless communication environment, a method of finding new available frequency bands and increasing the efficiency of limited resources can be considered. have. Among them, the transceiver is equipped with a plurality of antennas to secure additional spatial area for resource utilization, thereby obtaining diversity gain, or transmitting data in parallel through each antenna to increase transmission capacity. The so-called multi-antenna transmit and receive technology has recently been actively developed with great attention.

Referring to FIG. 1, a general structure of a multi-input multi-output (MIMO) system using orthogonal frequency division multiplexing (OFDM) among the multiple antenna transmission / reception techniques will be described. Same as

In the transmitting end, the channel encoder 101 attaches redundant bits to the transmitted data bits to reduce the influence of the channel or noise, and the mapper 103 converts the data bit information into data symbol information, and the serial-parallel converter. 105 parallelizes the data symbols for loading onto multiple subcarriers, and multi-antenna encoder 107 converts the parallelized data symbols into space-time signals. The multiple antenna decoder 109, the parallel-to-serial converter 111, the de-mapper 113, and the channel decoder 115 at the receiving end are the multiple antenna encoder 107, the serial-to-parallel converter 105, the mapper ( 103 and the reverse function of the channel encoder 101, respectively.

In a multi-antenna OFDM system, various techniques are required to increase data transmission reliability. Among them, a space-time code (STC) scheme and a cyclic delay diversity scheme are used to increase the spatial diversity gain. Diversity (CDD) and the like, and techniques for increasing the signal-to-noise ratio (SNR) include beamforming (BF) and precoding. Here, space-time code and cyclic delay diversity are mainly used to increase transmission reliability of an open loop system in which feedback information is not available at a transmitter, and beamforming and precoding are applicable in a closed loop system in which feedback information is available at a transmitter. It is used to maximize the signal to noise ratio through the feedback information.

Among the above-described techniques, a technique for increasing the spatial diversity gain and a technique for increasing the signal-to-noise ratio, in particular, look at cyclic delay diversity and precoding as follows.

In the cyclic delay diversity scheme, in the OFDM signal transmission system, all the antennas transmit signals with different delays or different sizes to obtain frequency diversity gains at the receiver. 2 illustrates a configuration of a transmitter of a multiple antenna system using a cyclic delay diversity scheme.

After the OFDM symbol is separately transmitted to each antenna through a serial-to-parallel converter and a multi-antenna encoder, a Cyclic Prefix (CP) for preventing interference between channels is attached and transmitted to the receiver. In this case, the data sequence transmitted to the first antenna is transmitted to the receiver as it is, but the data sequence transmitted to the next antenna is cyclically delayed by a predetermined bit compared to the antenna of the previous sequence.

On the other hand, if the cyclic delay diversity scheme is implemented in the frequency domain, the cyclic delay can be expressed as a product of a phase sequence. That is, as shown in FIG. 3, the data sequence in the frequency domain may be multiplied by a predetermined phase sequence (phase sequence 1 to phase sequence M) set differently for each antenna, and then may be transmitted to a receiver by performing a fast inverse Fourier transform (IFFT). This is called a phase shift diversity technique.

According to the phase shift diversity scheme, a flat fading channel can be changed into a frequency selective channel, thereby obtaining a frequency diversity gain or a frequency scheduling gain.

Precoding schemes include a codebook based precoding scheme used when the feedback information is finite in a closed loop system, and a method of quantizing and feeding back channel information. The codebook-based precoding is a method of obtaining a signal-to-noise ratio (SNR) gain by feeding back an index of a precoding matrix known to the transmitter / receiver to the transmitter.

4 illustrates a configuration of a transmitting and receiving end of a multiple antenna system using the codebook based precoding. Here, the transmitting end and the receiving end each have a finite precoding matrix ( P _1). ~ P _L ), and the receiver feeds back the optimal precoding matrix index ( l ) to the transmitter using channel information, and the transmitter sends the precoding matrix corresponding to the fed back index to the transmission data ( χ ₁ ~ χ _{Mt). )} it is applied to. For reference, Table 1 below shows an example of a codebook applicable to using 3 bits of feedback information in an IEEE 802.16e system having two transmit antennas and supporting spatial multiplexing rate 2.

In addition to the above-described advantages, the phase shift diversity scheme has received a lot of attention because of the advantages of obtaining frequency selective diversity gain in open loops and frequency scheduling gain in closed loops. There is a problem that it is difficult to obtain the gains if the data rate cannot be expected and the resource allocation is fixed.

In addition, the codebook-based precoding scheme described above has an advantage of enabling efficient data transmission because a high amount of spatial multiplexing rate can be used while requiring a small amount of feedback information (index information), but a stable channel must be secured for feedback. There is a problem that it is not suitable for a mobile environment in which the channel change is severe, and is applicable only in a closed loop system.

The present invention provides a phase shift based precoding technique that compensates for the shortcomings of the conventional cyclic delay diversity, phase shift diversity, and precoding scheme, and includes time-varying phase shift diversity and cyclic delay diversity. The objective of the present invention is to provide an improved phase shift based precoding technique or a cyclic delay diversity technique.

One aspect of the present invention for achieving the above object is a data transmission method in a multi-antenna system using a plurality of subcarriers, the spatial processing of the transmission symbol corresponding to each subcarrier in the frequency domain, and the spatial processing Converting a transmission signal into a time domain, multiplying a predetermined weight to the entire transmission signal converted into the time domain, and applying a predetermined cyclic delay to a transmission signal for each antenna. It is about a method.

According to another aspect of the present invention, there is provided a method of performing precoding in a frequency domain on transmission symbols corresponding to each subcarrier, converting the precoded transmission signal to a time domain, and transmitting a signal for each antenna. And applying a predetermined cyclic delay to the.

Another aspect of the present invention is to determine a phase shift based precoding matrix by multiplying a first matrix for phase shift and a second matrix for unitary matrixing the first matrix, and corresponding to each subcarrier. Multiplying the determined phase shift based precoding matrix by the transmission symbols, converting the transmitted signal on which the phase shift based precoding is performed into a time domain, and applying a predetermined cyclic delay to the transmission signal for each antenna The method may further include multiplying a predetermined weight to all transmission symbols converted into the time domain.

In another aspect of the present invention, a data transmission method in a multi-antenna system includes performing spatial processing associated with the multiple antenna on each transmission stream, and wherein each signal generated by the spatial processing is one of the multiple antennas. Performing precoding for transmission power allocation so as to be transmitted through the above; converting each signal generated by the precoding into a time domain signal for each antenna; and converting the converted time domain signal into at least one of the multiple antennas. Transmitting through.

In the above four aspects, the method may further include adding a first pilot symbol corresponding to each antenna before converting the transmission signal in the frequency domain to the time domain, wherein the first transformed signal is transformed in the time domain to the cyclically delayed transmission signal. The method may further include adding 2 pilot symbols for each antenna.

In the third aspect, the second matrix may be selected from a plurality of unit matrices previously stored in the codebook.

In addition, when the multiplexing rate of the phase shift based precoding matrix is 2 or more, each column of the phase shift based precoding matrix may be switched in a predetermined time unit, and the phase of the first matrix may be switched in a predetermined time unit. A predetermined offset may be added to each angle.

In addition, receiving a second matrix index and a column vector of the codebook from a plurality of receivers, selecting a second matrix of the fed back index, and selecting a column corresponding to the column vector from the matrix. The extracted second columns may be generated by combining the extracted columns.

Further, in the fourth aspect, the data transmission method includes one of applying phase shift diversity to each signal generated by the spatial processing and applying cyclic delay diversity to each transformed time domain signal. It may further include the above. Wherein the phase shift diversity is applied with a large cyclic delay value and the cyclic delay diversity is applied with a small cyclic delay value.

The data transmission method may further include adding a first pilot symbol for each antenna for each signal generated by the spatial processing, adding a second pilot symbol for each antenna for each signal generated by the precoding, and The method may further include at least one of adding, by antenna, a third pilot symbol transformed into a time domain for each signal to which the cyclic delay diversity is applied.

In the fourth aspect, the transmission power allocation precoding is performed by multiplying a unit matrix of N _t × N _t , wherein N _t is the number of the multiple antennas.

According to the phase shift based precoding scheme of the present invention, in addition to the conventional advantages of the phase shift diversity scheme and the precoding scheme, it is possible to adapt adaptively according to the channel or system situation regardless of the antenna structure and spatial multiplexing rate. There are advantages to it. In addition, the phase shift based precoding scheme can be selectively added to the phase shifting and cyclic delay diversity schemes over time to improve the complexity of the transceiver and to be used in combination with any antenna scheme of any structure. Can be applied differently to obtain the optimum communication performance.

Hereinafter, exemplary embodiments will be described in detail with reference to the accompanying drawings.

Example 1 Precoding Based on Phase Shift

Phase shift base Precoding  Generation of matrix

The phase shift based precoding matrix P can be expressed as follows.

(i = 1,...,N_t, j = 1,...,R)는 부반송파 인덱스 또는 특정 주파수 밴드 인덱스 k에 의해 결정되는 복소 가중치를 나타내고, N_t는 송신 안테나의 개수, R은 공간 다중화율을 각각 나타낸다. 여기서 송신 안테나는 물리적인 송신 안테나가 될 수도 있고, 가상의 송신 안테나를 의미할 수도 있다. 가상의 송신 안테나를 의미하는 경우 N_t와 R이 같다. 또한, 복소 가중치는 안테나에 곱해지는 OFDM 심벌 및 해당 부반송파의 인덱스에 따라 상이한 값을 가질 수 있다. 상기 복소 가중치는 채널 상황 및 피드백 정보의 유무 중 적어도 어느 하나에 따라 결정될 수 있다.here,

( i = 1, ..., N _t , j = 1, ..., R) represents a complex weight determined by the subcarrier index or the specific frequency band index k, N _t represents the number of transmit antennas, R represents the spatial multiplexing rate, respectively. Here, the transmission antenna may be a physical transmission antenna or may mean a virtual transmission antenna. In case of a virtual transmission antenna, N _t and R are the same. In addition, the complex weight may have a different value depending on the OFDM symbol multiplied by the antenna and the index of the corresponding subcarrier. The complex weight may be determined according to at least one of channel conditions and presence or absence of feedback information.

Meanwhile, the precoding matrix P of Equation 1 is preferably designed in a unitary matrix to reduce the loss of channel capacity in the multi-antenna system. Here, the channel capacity of the multi-antenna open-loop system in order to determine the conditions for constructing the unit matrix is represented as follows.

Where H is a multi-antenna channel matrix of size N _r x N _t and N _t is the number of transmit antennas, N _r represents the number of receiving antennas. The phase shift based precoding matrix P is applied to Equation 2 as follows.

As shown in Equation 3, in order to avoid loss in channel capacity, PP ^H must be a single matrix, so the phase shift based precoding matrix P must be a unit matrix satisfying the following conditions.

In order for the phase shift based precoding matrix P to become a unit matrix, two conditions, that is, a power constraint and an orthogonal constraint, must be satisfied at the same time. The power constraint is to make each column of the matrix have a size of 1, and the orthogonal constraint is to make orthogonality between each column of the matrix. Each of these expressions is expressed as follows.

Next, as an embodiment, a generalized equation of a 2 × 2 phase shift based precoding matrix will be presented, and the equations for satisfying the two conditions will be described. Equation 7 shows a general equation of a phase shift based precoding matrix having two transmit antennas and a spatial multiplexing rate of 2.

Here, α _i , β _i ( i = 1, 2) have a real value, θ _i (i = 1, 2, 3, 4) represents a phase value, and k represents a subcarrier index of an OFDM signal. In order to implement such a precoding matrix as a unit matrix, the power constraint of Equation 8 and the orthogonal constraint of Equation 9 must be satisfied.

Where the * symbol indicates a conjugate complex number. An embodiment of a 2 × 2 phase shift based precoding matrix that satisfies Equations 7 to 9 is as follows.

Here, θ ₂ and θ ₃ have the same relationship as in Equation 11 due to orthogonal constraint.

Meanwhile, the precoding matrix may be stored in a codebook form in memories of the transmitter and the receiver, and the codebook may include various precoding matrices generated through finite different θ ₂ values. Here, the value of θ ₂ may be appropriately set according to the channel condition and the presence or absence of feedback information. For example, when using feedback information, the frequency scheduling gain may be obtained by setting θ ₂ to be small, and the feedback information may be obtained. If not used, a high frequency diversity gain can be obtained by setting θ ₂ large.

On multiplexing rate  Based on phase shift Precoding  Reconstruction of matrix

On the other hand, even if a phase shift based precoding matrix is generated as shown in Equation 7, the spatial multiplexing rate may actually be set smaller than the number of antennas depending on the channel situation. In this case, a new phase shift based precoding matrix can be reconstructed by selecting a specific column corresponding to the current spatial multiplexing rate (smaller spatial multiplexing rate) among the generated phase shift based precoding matrices. . That is, instead of generating a new precoding matrix applied to the system whenever the spatial multiplexing rate is different, the precoding matrix is selected by using a phase shift-based precoding matrix that is originally generated, but selecting a specific column of the precoding matrix. Just reconfigure

As an example, the precoding matrix of Equation 10 assumes a case where the spatial multiplexing rate is 2 in a multi-antenna system having two transmit antennas, but for some reason, the spatial multiplexing rate of the system may decrease to 1. have. In this case, a specific column of the matrix of Equation 10 may be selected and reconstructed into a precoding matrix having a spatial multiplexing ratio 1. An exemplary phase shift based precoding matrix, in which the second column is selected, is represented by Equation 12 below. . This has the same form as the cyclic delay diversity scheme of two transmit antennas in the related art.

Here, the system having two transmit antennas is taken as an example, but it can be extended to a system having four transmit antennas, and the spatial multiplexing after generating a phase shift based precoding matrix when the spatial multiplexing rate is 4 Precoding can be performed by selecting a specific column according to the burn rate.

As an example of this, FIG. 5 illustrates a case where conventional spatial multiplexing and cyclic delay diversity are applied to a multiple antenna system having four transmit antennas and a spatial multiplexing rate of 2. FIG. 6 illustrates a case where cyclic delay diversity is applied to the multi-antenna system as in FIG. 5 together with the phase shift based precoding matrix of Equation 10. FIG. 5 and 6, the cyclic delay diversity is illustrated as an operation of multiplying a phase shift sequence, and assumes that the phase angle of the phase shift by the phase shift sequence is θ ₁ .

) 및 제2 시퀀스(

)가 전달되고, 제2 안테나 및 제4 안테나에는 소정 크기로 위상천이된 제1 시퀀스(

) 및 제2 시퀀스(

)가 전달된다. 따라서, 전체적으로는 공간 다중화율이 2가 됨을 알 수 있다.Referring to FIG. 5, the first and third antennas have a first sequence (

) And second sequence (

) Is transmitted, and the first sequence

) And second sequence (

) Is passed. Therefore, it can be seen that the spatial multiplexing ratio is 2.

가 전달되고, 제3 안테나에는

가 전달되며, 제2 안테나에는 소정 크기로 위상천이 된

가 전달되고, 제4 안테나에는 제2 안테나의 경우와 마찬가지로 소정 크기로 위상천이 된

가 전달된다. 따라서, 도 5의 시스템에 비해 프리코딩 기법의 장점을 가지면서도 단일한 프리코딩 행렬을 이용하여 4개의 안테나에 대해 순환지연(또는 위상천이)을 수행할 수 있으므로 순환지연 다이버시티 기법에 의한 장점까지 가질 수 있다.In contrast, the first antenna in FIG.

Is transmitted to the third antenna

Is transmitted to the second antenna and phase shifted to a predetermined size.

Is transmitted to the fourth antenna and phase shifted to a predetermined size as in the case of the second antenna.

Is passed. Therefore, while having the advantages of the precoding scheme compared to the system of FIG. 5, the cyclic delay (or phase shift) can be performed for four antennas using a single precoding matrix. Can have

The phase shift based precoding matrix according to spatial multiplexing rates for the two antenna systems and the four antenna systems discussed above are summarized as follows.

Here, θ _i ( i = 1, 2, 3) represents the phase angle according to the cyclic delay value, and K represents the subcarrier index of OFDM. In Table 2, each of the four precoding matrices can be obtained by taking a specific portion of the precoding matrix for a multi-antenna system having four transmit antennas and a spatial multiplexing ratio of two, as shown in FIG. Therefore, since there is no need to separately include each precoding matrix for the four cases in the codebook, the memory capacity of the transmitting end and the receiving end can be saved. In addition, the above-described phase shift based precoding matrix can be extended for a system having an antenna number M and a spatial multiplexing rate N by the same principle.

Example 2 Generalized Phase Shift Diversity

In the above description, a process of configuring a phase shift based precoding matrix in the case of four transmitting antennas and a spatial multiplexing ratio of 2 has been described. However, in the case of a phase shift based precoding, the number of transmitting antennas is N _t. ( N _t is a natural number of 2 or more) and a spatial multiplexing rate of R ( R is a natural number of 1 or more) is described. Hereinafter, generalized phase shift based precoding is called Generalized Phase Shift Diversity (GPSD).

8 is a block diagram showing the main configuration of a transceiver for performing generalized phase shift diversity.

The generalized phase shift diversity method transmits all streams to be transmitted through all antennas, but multiplies and transmits a sequence of different phases for each antenna.

For example, as shown in FIG. 8, stream 1 is transmitted through all the antennas provided from antenna 1 to antenna M, but when stream 1 is transmitted to antenna 1, it is transmitted without phase shift and is transmitted to antenna 2. In the case of transmission, the phase shift with respect to the phase angle of P1 (1) is applied to be transmitted. In this way, phase shifts of different phase angles are applied to antenna M to be transmitted.

Similarly, stream 2 is also transmitted through all the antennas provided from antenna 1 to antenna M, but when stream 2 is transmitted to antenna 1, it is transmitted without phase shift, and when transmitted to antenna 2, phase angle of P2 (1). A phase shift for is applied to allow transmission. In this way, phase shifts of different phase angles are applied to antenna M to be transmitted. Referring to FIG. 8, it can be confirmed that the same method is applied to the remaining streams up to stream S.

The generalized phase shift diversity method described above may be represented by a combination of matrices such as Equation 13.

는 N_t개의 송신 안테나와 R의 공간 다중화율을 가지는 MIMO-OFDM 신호의 k번째 부반송파에 대한 GPSD 행렬을 나타내며,

는

를 만족하는 단위 행렬(제2행렬)로서 각 송신 안테나를 통해 송신되는 전력 크기를 동일하게 할 수 있다. 여기서 k는 부반송파 인덱스뿐만 아니라 상황에 따라서 단위 자원 별로 할당되는 인덱스 또는 하나 이상의 부반송파를 포함하여 이루어지는 주파수 밴드 별로 할당되는 인덱스 정보가 될 수도 있다. GPSD 행렬은 각 송신 안테나 별로 서로 다른 위상천이 각도를 적용할 수 있도록 하는 위상천이 행렬(제1행렬)과 단위행렬(unitary matrix)(제2행렬)을 곱하여 구성할 수 있다. 대각 행렬인 제1행렬과 단위 행렬인 제2행렬을 곱한 결과인 GPSD 행렬 또한 단위행렬의 특징을 만족할 것이므로 좋은 효과를 내는 프리코딩 행렬로서 이용될 수 있다. 수학식 13에서 위상각 θ_i, i=1,...,N_t 은 지연 값 τ_i, i=1,...,N_t에 따라 아래의 수학식 14과 같이 얻을 수 있다.here,

Denotes the GPSD matrix for the k th subcarrier of the MIMO-OFDM signal having N _t transmit antennas and a spatial multiplexing rate of R,

Is

As a unit matrix (second matrix) that satisfies, power magnitudes transmitted through each transmission antenna may be equalized. Here, k may be not only a subcarrier index but also index information allocated for each unit resource according to a situation or index information allocated for each frequency band including one or more subcarriers. The GPSD matrix may be configured by multiplying a phase shift matrix (first matrix) and a unitary matrix (second matrix) so that different phase shift angles may be applied to each transmitting antenna. The GPSD matrix, which is a result of multiplying the first matrix, which is a diagonal matrix, and the second matrix, which is a unitary matrix, may also be used as a precoding matrix having a good effect because it will satisfy the characteristics of the unit matrix. In Equation 13, the phase angles θ _i , i = 1, ..., N _t may be obtained as shown in Equation 14 below according to the delay values τ _i , i = 1, ..., N _t .

Here, N _fft represents the number of subcarriers of the OFDM signal.

An example of the GPSD matrix generation equation in the case of using two 1-bit codebooks with two transmit antennas is as follows.

When the value of α is determined in Equation 15, the value of β is easily determined. Therefore, information on the value of α may be set to two appropriate values and the information may be fed back to the codebook index. For example, if the feedback index is 0, α may be 0.2, and if the feedback index is 1, α may be 0.8.

As an example of the second matrix, a predetermined characteristic matrix may be used to obtain a signal-to-noise ratio (SNR) gain. In particular, a GPSD matrix generation equation in which the Walsh code is used as the characteristic matrix is as follows. Same as

Equation (16) assumes a system having four transmit antennas and a spatial multiplexing rate of 4, wherein a particular transmit antenna is selected or the spatial multiplexing rate is adjusted by appropriately reconstructing the second matrix. )can do.

Equation 17 below shows the reconstruction of the second matrix to select two antennas in a system having four transmit antennas.

In addition, Table 3 below shows a method for reconstructing the second matrix to match the multiplexing rate when the spatial multiplexing rate changes according to time or channel conditions.

In this case, Table 3 shows a case where the first column, the first to second columns, and the first to fourth columns of the second matrix are selected according to the multiplexing rate. However, the present disclosure is not limited thereto and the multiplexing rate is 1. In this case, any one of the 1,2,3,4th columns may be selected, and when the multiplexing rate is 2, any one of the 1, 2, 2, 3, 3-4, and 4-1st columns may be selected. .

Meanwhile, the second matrix may be provided in the form of a codebook at the transmitting end and the receiving end. In this case, the transmitting end receives feedback of the index information of the codebook from the receiving end, selects the unit matrix (second half matrix) of the corresponding index from its own codebook, and constructs a matrix as shown in Equation (13).

In addition, the second matrix may be periodically changed such that carrier (s) transmitted in the same time slot have different precoding matrices for each frequency band.

Meanwhile, the above-described phase angle, ie, a cyclic delay value, for performing the generalized phase shift diversity (GPSD) may be a predetermined value to the transceiver, or may be a value transmitted by the receiver to the transmitter through feedback. In addition, the spatial multiplexing rate (R) may also be a predetermined value for the transceiver, but the receiver may periodically determine the channel state to calculate the spatial multiplexing rate and feed it back to the transmitter. Multiplexing rates can also be calculated and changed.

An example of the GPSD matrix using 2 x 2 and 4 x 4 Walsh codes as the unit matrix for obtaining the GPSD is as follows.

The flat fading channel can be changed to a frequency selective channel by the phase shift based precoding or the generalized phase shift diversity of the embodiments 1 to 2 described above, and the frequency diversity gain according to the delay sample size. Or frequency scheduling gain. 9 graphically illustrates two applications of phase shift based precoding or phase shift diversity.

As shown in the upper right of FIG. 9, when a large value of the cyclic delay (or delayed sample) is used, the frequency selectivity period is shortened, so that the frequency selectivity is increased, and thus the channel code can obtain the frequency diversity gain.

In addition, even when the channel size is smaller than the channel size required for transmission and reception in a flat fading channel situation, the use of the above-described large value of the cyclic delay may increase the frequency selectivity, thereby increasing the probability of transmission and reception success. This is preferably used in open loop systems where the temporal change of the channel is severe and the reliability of feedback information is poor.

In addition, as shown in the lower right side of FIG. 9, when a small value of cyclic delay is used, a period of frequency selectivity becomes long, so that in a closed loop system, a frequency scheduling gain can be obtained by allocating resources to a region having the best channel. . That is, in the case of generating a phase sequence using a small value of cyclic delay in applying phase shift based precoding or generalized phase shift diversity, as shown in FIG. There is a bigger and smaller part. Therefore, a certain subcarrier region of the OFDM signal has a larger channel size, and the other subcarrier region has a smaller channel size.

As shown in the figure, the transmitter may secure frequency diversity by allocating a user terminal to a portion in which channel strength is increased in a frequency band that fluctuates according to a relatively small cyclic delay value. In this case, a phase shift based precoding matrix is used to apply a cyclic delay value that increases or decreases uniformly for each antenna.

In this case, in an orthogonal frequency division multiple access (OFDMA) system that accommodates multiple users, the signal-to-noise ratio can be increased by transmitting a signal through a constant frequency band with a larger channel size for each user, and a frequency band with a larger channel size for each user. This different case often occurs, so the system gains multi-user diversity scheduling gains. In addition, since the receiver only needs to transmit CQI (Channel Quality Indicator) information of a subcarrier region that can be allocated to each resource simply as feedback information, the feedback information is relatively reduced.

Example 3 Time Varying Generalized Phase Shift Diversity

In the GPSD of Equation 13, the phase angle θ _i and the unitary matrix U may change with time. Such a time-variable GPSD can be expressed as follows.

는 특정 시간 t에서 N_t개의 송신 안테나와 R의 공간 다중화율을 가지는 MIMO-OFDM 신호의 k번째 부반송파에 대한 GPSD 행렬을 나타내며,

는

를 만족하는 단위 행렬(제4행렬)로서 위상천이 행렬(제3행렬)을 단위행렬(unitary matrix)화하기 위해 사용된다. 여기서도 k는 부반송파 인덱스뿐만 아니라 상황에 따라서 단위 자원 별로 할당되는 인덱스 또는 하나 이상의 부반송파를 포함하여 이루어지는 주파수 밴드 별로 할당되는 인덱스 정보가 될 수도 있다. here,

Denotes the GPSD matrix for the kth subcarrier of the MIMO-OFDM signal having N _t transmit antennas and R spatial multiplexing rate at a specific time t.

Is

It is used to unitize the phase shift matrix (third matrix) as a unit matrix (fourth matrix) that satisfies. Here, k may be not only a subcarrier index but also index information allocated for each unit resource according to a situation or index information allocated for each frequency band including one or more subcarriers.

Equation 18b below shows a result of obtaining a transmission signal by multiplying a data stream vector having a spatial multiplexing rate of R by using the GPSD matrix described in Equation 18a.

In Equation 18b, x (t) represents a data stream vector having the above-mentioned spatial multiplexing rate R, and y (t) represents a transmission signal vector.

In equations 18a and 18b, the phase angles θ _i (t), i = 1, ..., N _t are given by the following _{equations according} to the delay values τ _i (t), i = 1, ..., N _t : Equation 19 can be obtained.

Here, N _fft represents the number of subcarriers of the OFDM signal.

As shown in Equation 18 and Equation 19, the time delay sample value and the unit matrix may change over time, and the unit of time may be an OFDM symbol unit or a unit of time.

An example of a GPSD matrix using 2 x 2 and 4 x 4 Walsh codes as a unit matrix for obtaining a time-varying GPSD can be summarized as follows.

Example 4 Generalized Cyclic Delay Diversity

Since the phase shift based precoding and generalized phase shift diversity of Embodiments 1 to 3 are used in the frequency domain as shown in FIG. 5, the phase shift based precoding matrix or the generalized phase shift diversity for every subcarrier, unit resource, or frequency band. Since the matrix needs to be multiplied, the design of the sender tends to be complicated. In addition, since the receiving side also estimates the multi-antenna channel and calculates an equivalent channel by calculating the matrices each time according to the delay sample, the receiver has a complex structure. Accordingly, the present embodiment simplifies the design of the transceiver by implementing the phase shift based precoding and generalized phase shift diversity of the first to third embodiments in the time domain. This technique is called Generalized Cyclic Delay Diversity (GCDD).

10 conceptually illustrates a structure of a transceiver supporting GCDD.

As shown in FIG. 10, a signal that has undergone spatial processing has an inverse discrete Fourier transform (IFFT) applied to each antenna, and then a complex weight is applied in the time domain before being transmitted through each antenna. Circulation delay is according to each cyclic delay sample value. In FIG. 10, a complex weight is represented by uij, and uij represents a complex weight multiplied by a j-th IFFT output signal transmitted through an i-th antenna. Fig. 10 shows GCDD corresponding to the time-varying GPSD of Embodiment 3 in particular.

As can be seen with reference to FIG. 10, each IFFT output signal can be transmitted through all antennas by multiplying the complex weights independently of each antenna. In other words, in the time domain, each transmission stream may be multiplied by a different complex weight for each antenna to transmit through all antennas. The above-described unit matrix may be used to apply a complex weight to the IFFT output signal. In this case, the effect that the power of the signal transmitted through each transmission antenna can be evenly distributed can be expected. For example, when the number of transmitting antennas is 4, using the aforementioned 4x4 Walsh code as the precoding matrix, the complex weight for each antenna may be 1 or -1.

Example 5 Modification of Generalized Cyclic Delay Diversity

11 conceptually illustrates a structure of a transceiver supporting a modification scheme of GCDD.

는 특정 시간 t에서 N_t개의 행과 K개의 열을 가지는 임의의 프리코딩 행렬을 나타낸다. 바람직하게는 단위 행렬의 특징을 갖는 프리코딩 행렬이 될 것이다. 여기서, N_t는 송신 안테나 수에 상응하는 값이고 K는 프리코딩 행렬이 곱해지는 즉, 프리코더에 입력되는 OFDM 심볼 수에 상응하는 값이다. In the GCDD of the fourth embodiment, complex weights and cyclic delays are applied after the IFFT. However, as shown in FIG. It is applied to the time domain after IFFT using diversity technique. Here, the cyclic delay value may change over time. In addition, in FIG.

Denotes an arbitrary precoding matrix having N _t rows and K columns at a particular time t. Preferably it will be a precoding matrix having the characteristics of an identity matrix. Here, N _t is a value corresponding to the number of transmit antennas and K is a value corresponding to the number of OFDM symbols input to the precoder, that is, the precoding matrix is multiplied.

Example 6 Combination of GPSD and GCDD

This embodiment combines the GCDD in the time domain and the GPSD in the frequency domain to reduce the complexity of the transceiver and to combine it with the multiple antenna scheme of any structure according to the situation. In addition, when a plurality of users having different channels are allocated different frequency resources, an additional multi-antenna technique or a different cyclic delay sample value may be applied according to the user's channel to obtain the most suitable delay sample and multi-antenna technique. Make it applicable.

12 conceptually illustrates a structure of a transceiver to which a GPSD and a GCDD are combined and applied.

In FIG. 12, PS_i (j) represents a phase sequence sequence multiplied by an i-th OFDM symbol transmitted through the j-1 th antenna, and uij represents a j-th IFFT output signal transmitted through the i-th antenna. Τ _i (t) denotes a cyclic delay value applied at time t for a signal transmitted through the i-th antenna. A cyclic delay in the frequency domain can be realized through the phase shift sequence of PS_i (j), and a cyclic delay in the time domain can be implemented through the cyclic delay value of τ _i (t).

Example 7 Modification of Combination of GPSD and GCDD

In the sixth embodiment, the combination of GPSD and GCDD can be simplified by applying only the cyclic delay in the time domain and applying the process excluding the cyclic delay in the frequency domain as precoding.

FIG. 13 conceptually illustrates a transceiver structure in a case where a combination of GPSD and GCDD is modified according to the present embodiment.

As shown in FIG. 13, GPSD or precoding is applied to all users before the IFFT, and the precoder for precoding may be fixed or may have received feedback from the receiving end. In addition, the precoder, each phase value for the phase shift, and the delay sample value may change over time.

According to the present embodiment, it can be applied to all multi-antenna techniques having a precoder structure, and GPSD can be selectively used according to different frequency bands for each user. In addition, GPSD and precoding may be applied exclusively to each other, or both may be applied simultaneously.

In the sixth embodiment and the seventh embodiment, the combination of the GPSD and the GCDD in this way allows the gain of the cyclic delay applied in the frequency domain and the gain of the cyclic delay applied in the time domain. In addition, more effective resource utilization is possible considering the difference in the gains that can be obtained according to the delay size of the cyclic delay.

For example, as described above, a frequency diversity gain can be obtained when a large delay value is applied, and a frequency scheduling gain can be obtained when a small delay value is applied. Therefore, a large delay value is applied to the frequency domain. However, the frequency utilization rate can be increased by selectively applying the frequency or frequency group. In addition, frequency scheduling may be performed on the transmission signal by applying a small delay value in the time domain.

In other words, if different frequency domains are allocated according to users, a small delay sample is applied to all user frequency bands using a basic GCDD, and a delay for a specific user in a specific frequency domain using a GPSD or a precoder. The value may be applied to simultaneously acquire the frequency scheduling gain and the frequency diversity gain.

Example 8 Application Example 1 of Pilot Symbol

Various pilot symbols for channel estimation may be applied to the technique according to the embodiments described above. FIG. 14 illustrates a case in which pilot symbols are applied together with cyclic delay diversity by applying pilot symbols before the IFFT in Embodiment 7, and FIG. 15 illustrates an example in which only a cyclic delay is applied in the time domain described with reference to FIG. As shown in FIG. 14, the pilot symbols are processed before the IFFT to apply the cyclic delay diversity to the pilot symbols. However, the embodiments related to the pilot symbols described below including the eighth embodiment are described. It is not necessarily limited to the seventh embodiment, but may be applied to the first to seventh embodiments and any technique obviously deformable therefrom.

Here, since the pilot symbols are affected by the OFDM symbols and the cyclic delay diversity, the receiving end only needs to have the channel estimation and the equivalent channel for the GPSD without separately providing a channel estimation circuit for the pilot symbols. Therefore, there is an advantage that the complexity of the receiving end is reduced. The pilot transmitted in this manner is called a dedicated pilot. [Embodiment 9]-Application Example 2 of Pilot Symbol

FIG. 16 illustrates a case in which a pilot symbol is applied after cyclic delay diversity in the seventh embodiment, and FIG. 17 illustrates a pilot symbol similar to FIG. 16 for an example in which only a cyclic delay is applied in the time domain described with reference to FIG. 13. The case of application after the cyclic delay diversity is illustrated. In this case, since a pilot symbol is received without cyclic delay diversity, a channel estimation circuit for the pilot symbol must be separately provided, so that the complexity increases somewhat compared to the eighth embodiment. Since channel estimation is performed on a real channel without being influenced by, the performance of channel estimation is improved. The pilot transmitted in this manner is called a common pilot.

<Example 10>-Application Example 3 of Pilot Symbol

FIG. 18 illustrates a case where the pilot symbol application methods of the eighth embodiment and the ninth embodiment are used at the same time. That is, it is possible to use both pilot symbols to which cyclic delay diversity is applied in the time domain and pilot symbols to which cyclic delay diversity is not applied in the time domain. For example, assume that a large delay cyclic delay diversity scheme is applied in the frequency domain and a small delay cyclic delay diversity scheme is applied in the time domain. In this case, the pilot terminal applied before the IFFT can be used to obtain an equivalent channel with a cyclic delay diversity for the cyclic delay diversity technique with a small delay, and a pilot symbol applied after the cyclic delay diversity is used. Therefore, it is possible to expect the effect of reducing the complexity of the receiver without degrading the performance of channel estimation by enabling the receiver to acquire the actual channel.

In FIG. 18, an example of transmitting pilot symbols separately for whether cyclic delay diversity is applied in the time domain is shown, but the same pilot symbol application method may be used for cyclic delay diversity in the frequency domain. That is, it means that the pilot symbol may be applied before the cyclic delay diversity in the frequency domain is performed. In this case, the pilot symbol will apply not only diversity in frequency domain but also diversity in time domain. As described above, if a large delay cyclic delay is used for cyclic delay diversity in the frequency domain, a large delay cyclic delay diversity through a pilot symbol to which both the frequency and time domain diversity are applied at the receiving end. It is possible to obtain an equivalent channel to which is applied.

By transmitting the dedicated pilot and the common pilot at the same time, the following effects can be obtained.

First, when the information amount of the dedicated pilot is larger than the common pilot, when receiving feedback on a specific channel, the receiver may estimate that a certain transmission delay value should be used in that channel for better performance. Therefore, the receiver may increase the transmission efficiency by estimating the transmission delay value for optimal performance and feeding it back to the transmitter.

Second, when the information amount of the common pilot is larger than the dedicated pilot, the receiver may estimate the channel as the common pilot and then estimate the transmission delay between the transmitter and the receiver by measuring a transmission delay with the dedicated pilot. Therefore, the transmitting end does not need to inform the receiving end of the transmission delay value between the transmitting end and the receiving end during data transmission, thereby increasing transmission efficiency within limited resources.

Link performance is compared with the GCDD system of the above-described embodiment 4 and a conventional system such as Per-Antenna Rate Control (PARC) or Virtual Antenna Permutation (VAP). The performance of the system proposed by the present invention shown through FIGS. 19 and 20 shows the results of experiments for the case in which the system parameters are shown in Table 1 below.

Parameter Assumption OFDM parameters 5 MHz (300 + 1 subcarriers) Subframe length 0.5 ms Resource block size 75 subcarriers * 5 OFDM symbol Channel models ITU Pedestrian A, Typical Urban (6-ray) Mobile Speed (km / h) 3 Modulation schemes and channel coding rates QPSK (R = 1/3, 1/2, 3/4) 16-QAM (R = 1/2, 5/8, 3/4) 64-QAM (R = 3/5, 2/3, 3 / 4, 5/6) Channel Code Turbo code Component decoder: max-log-MAP MIMO mode SU-MIMO Resource allocation Localized mode Codeword MCW Antenna configuration 2 x 2 Antenna selection option 2 antenna groups (1bit ASI) Spatial correlation (Tx, Rx) (50%, 50%) MIMO receiver MMSE receiver CQI update period 3 TTIs CQI option Full CQI Channel Estimation Perfect channel estimation H-ARQ Bit-level chase combining # of Maximum Retransmission: 3 # of Retransmission delay: 6 TTIs

FIG. 19 shows the results of the simulation performed on the GCDD system and the conventional system in the ITU pedestrian-A channel, and FIG. 20 shows the results of the simulation performed in a typical urban (6-ray) environment. .

19 and 20, it can be seen that a high performance gain can be obtained in a MIMO-OFDM system using GCDD.

Example 11 Precoding Matrix for Transmission Power Allocation

After the spatial processing is performed on the OFDM symbol or the data stream, the transmission power transmitted from each transmission antenna may be adjusted by multiplying a precoding matrix for transmission power allocation before or after IFFT is performed for each antenna signal. .

21 is a diagram of an exemplary transceiver structure applying a precoding matrix for transmission power allocation according to an embodiment of the present invention.

Referring to FIG. 21, it can be seen that an example of applying a transmission power allocation precoding matrix when cyclic delay is applied in the time domain. After spatial processing is performed on the OFDM symbol or the data stream, precoding matrix processing for transmission power allocation is performed. After the transmission power allocation precoding matrix is multiplied, signal processing for IFFT and cyclic delay is performed for each transmit antenna signal and transmitted to the receiver through the corresponding transmit antenna.

More specifically, the case where a unit matrix of N _t × N _t is used as the precoding matrix for transmission power allocation will be described. As described above with respect to the unit matrix, the unit matrix must satisfy a power constraint that the size of each column constituting the matrix of the unit matrix is one. As a characteristic of such a power constraint, the transmission power transmitted from each transmission antenna may be equalized.

22 is a diagram of an exemplary transceiver structure applying a precoding matrix for transmission power allocation according to an embodiment of the present invention.

Referring to FIG. 22, it can be seen that an example of applying a transmission power allocation precoding matrix when cyclic delay diversity is applied in a frequency domain. In the present embodiment, it may be described as applying a phase shift based precoding matrix.

After spatial processing is performed on an OFDM symbol or data stream, a phase shift matrix for cyclic delay diversity and a precoding matrix for transmission power allocation are performed. After the transmission power allocation precoding matrix is multiplied, signal processing for the IFFT is performed for each transmit antenna signal and transmitted to the receiver through the corresponding transmit antenna. Although the phase shift matrix can be used in various embodiments, the phase shift matrix proposed as a component of the GPSD matrix can be used. Equation 20 shows a phase shift matrix proposed as one component of the GPSD matrix.

는 시간(t)에 대해 가변적 으로 사용할 수도 있고 고정하여 사용할 수도 있다. 수학식 20에 나타난 대각 행렬을 송신단에서 곱하여줌으로써 상술한 바와 같은 주파수 영역에서 송신 안테나 별로 순환지연 적용될 수 있다.Here, k may be not only a subcarrier index but also index information allocated for each unit resource according to a situation or index information allocated for each frequency band including one or more subcarriers. Also,

Can be used variably or fixedly for time (t). By multiplying the diagonal matrix shown in Equation 20 at the transmitter, cyclic delay may be applied to each transmit antenna in the frequency domain as described above.

The form of combining the phase shift matrix of Equation 20 and the aforementioned precoding matrix for transmission power allocation is shown in Equation 21 below.

In the case of Equation 21, it can be seen that it has a form similar to the above-described GPSD matrix. However, in the present embodiment, it is assumed that the precoding matrix of Equation 21 is multiplied with the signal on which the spatial processing is performed, and thus, the above-described GPSD matrix and the unit matrix have a difference in matrix size. That is, in the above-described GPSD matrix, a unit matrix of N _t × R is used, but in this embodiment, a unit matrix of N _t × N _t is used.

In other words, the present embodiment shows that any spatial processing method may be used to allocate transmission power by a unit matrix independently of spatial processing, and in this case, phase shift or cyclic delay may be applied together.

와

는 시간(t)에 따라 가변적으로 사용할 수도 있고 고정하여 사용할 수도 있음은 앞서 설명한 바와 같다.here

Wow

May be used variably or fixedly according to time t, as described above.

를 얻기 위하여 공간 처리부의 출력 값은 공간 처리 기법에 따라 다양한 형태를 가질 수 있다. 공간 처리부의 출력 값이

길이를 가지는 벡터

라고 하면 송신신호

는 수학식 22와 같이 나타낼 수 있다.Transmission signal using the determinant mentioned in Equation 21 above

In order to obtain the output value of the spatial processing unit may have various forms according to the spatial processing technique. The output value of the spatial processor

Vector with length

Speaking of transmission signal

May be expressed as in Equation 22.

및

는 시간(t)에 따라 가변적으로 사용할 수도 있고 고정하여 사용할 수도 있다.Equation 22

And

May be used variably or fixed depending on the time (t).

In Equations 20 to 22, the phase angles θ _i (t), i = 1, ..., N _t are given by the following _{equations according} to the delay values τ _i (t), i = 1, ..., N _t . It can be obtained as shown in Equation 23.

Here, N _fft represents the number of subcarriers of the OFDM signal.

As shown in Equation 20 to Equation 23, the time delay sample value and the unit matrix may change with time, and the unit of time may be an OFDM symbol unit or a unit of time.

Through the transmission power allocation precoding matrix according to the present embodiment, the transmission power transmitted from each transmission antenna is equalized, and thus the transmission power of each power amplifier of each antenna of the transmitter can be balanced. In addition, when used in conjunction with a phase shift or time delay diversity scheme, an effect of solving a problem in which a transmission signal is transmitted only in a specific direction is also expected.

23 is an exemplary transceiver structure diagram applying a precoding matrix for transmission power allocation according to an embodiment of the present invention.

Referring to FIG. 23, it can be seen that an example of applying a transmission power allocation precoding matrix when cyclic delay diversity is applied in the time domain and the frequency domain. That is, the embodiments described with reference to FIGS. 21 and 22 may be combined.

As described above, when the cyclic delay diversity is applied in the time domain and the frequency domain, the gain of the cyclic delay applied in the frequency domain and the gain of the cyclic delay applied in the time domain are combined as in the embodiment of the combination of GPSD and GCDD described above. Can be taken In addition, more effective resource utilization is possible considering the different gains that can be obtained according to the delay size of the cyclic delay, such as applying a large cyclic delay in the frequency domain and a small cyclic delay in the time domain.

Hereinafter, examples of applying a pilot symbol in the transceiver according to the above-described embodiments will be described.

FIG. 24 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 21 or FIG. 25 is an exemplary transceiver structure diagram of applying a pilot symbol to the embodiment shown in FIG. 22.

As can be seen with reference to Figures 24 and 25, the pilot symbol is applied before the IFFT is performed. Therefore, the cyclic delay diversity performed for each antenna after the IFFT process can also be used for the pilot symbol. In this case, since the pilot symbols are affected by the OFDM symbols and the cyclic delay diversity, the receiver need only include the channel estimation and the equivalent channel for the GPSD without separately providing a channel estimation circuit for the pilot symbols. Therefore, there is an advantage that the complexity of the receiving end is reduced.

In other words, by using the same phase shift or time delay in the pilot symbol, it is possible to solve the problem of increasing additional complexity in the receiver. However, depending on the situation, the pilot symbols do not use phase shift or time delay so that all receivers can estimate a channel to which no phase shift or time delay diversity scheme is applied.

FIG. 26 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 21 or FIG.

As can be seen with reference to FIG. 26, a pilot symbol is applied after IFFT is performed and cyclic delay diversity is also performed. Therefore, the cyclic delay diversity performed for each antenna after the separate IFFT processing is not applied to the pilot symbols.

In this case, since a pilot symbol is received without cyclic delay diversity, a channel estimation circuit for a pilot symbol must be separately provided, so that complexity increases somewhat compared to the embodiment in which cyclic delay diversity is applied to a pilot symbol. However, since the pilot symbols are channel estimation for actual channels without being affected by cyclic delay diversity, that is, phase shift, there is an advantage in that channel estimation performance is improved.

FIG. 27 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 23.

As can be seen with reference to Figure 27, the pilot symbol is applied before the cyclic delay, that is, the phase shift in the frequency domain. Therefore, not only the cyclic delay diversity in the time domain but also the cyclic delay diversity in the frequency domain are applied to the pilot symbols after the IFFT processing.

FIG. 28 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 21 or FIG.

As can be seen with reference to FIG. 28, a pilot symbol for each antenna is applied two or more times. That is, an example in which pilot symbols are applied two or more times, such as a pilot symbol applied before the IFFT is performed and a pilot symbol applied after the cycle delay diversity is performed in the time domain, is performed.

Thus, by transmitting both the pilot symbol to which cyclic delay diversity is applied and the pilot symbol to which cyclic delay diversity is not applied to the receiver, not only an equivalent channel to which cyclic delay diversity is applied but also actual channel information to which cyclic delay diversity is not applied. Can be acquired.

Although not illustrated in the drawings, the pilot symbols to which both cyclic delay diversity is applied in the frequency domain and the time domain may also be applied separately or together with the pilot symbol applications described above.

The present invention described above is capable of various substitutions, modifications, and changes without departing from the spirit of the present invention for those skilled in the art to which the present invention pertains. It is not limited by the drawings.

1 is a block diagram of an orthogonal frequency division multiplexing system having multiple transmit / receive antennas.

2 is a block diagram of a transmitter of a multiple antenna system using a conventional cyclic delay diversity scheme.

3 is a block diagram of a transmitter of a multi-antenna system using a conventional phase shift diversity scheme.

4 is a block diagram of a transmitting and receiving end of a multiple antenna system using a conventional precoding technique.

FIG. 5 is a block diagram illustrating a main configuration of a transceiver for performing phase shift based precoding.

FIG. 6 illustrates a case where a spatial multiplexing technique and cyclic delay diversity are applied to a multi-antenna system having four transmit antennas and a spatial multiplexing ratio of 2. FIG.

FIG. 7 illustrates a case where a phase shift based precoding matrix is applied to the multi-antenna system of FIG. 6.

8 illustrates a reconstruction method of a phase shift based precoding matrix.

9 graphically illustrates two applications of phase shift based precoding or phase shift diversity.

10 conceptually illustrates a structure of a transceiver supporting GCDD.

11 conceptually illustrates a structure of a transceiver that supports a modification scheme of GCDD.

12 conceptually illustrates a structure of a transceiver to which a GPSD and a GCDD are combined and applied.

FIG. 13 conceptually illustrates a transceiver structure when a combination of GPSD and GCDD is modified.

14 illustrates a case where a pilot symbol is also applied to a GPSD performed before the IFFT.

FIG. 15 is a mathematical expression representing a GPSD part in FIG. 14.

FIG. 16 illustrates a case in which pilot symbols are applied after cyclic delay diversity.

FIG. 17 is a mathematical expression representing a GPSD part in FIG. 16.

FIG. 18 illustrates a case where the pilot symbol application methods of FIGS. 14 and 16 are used simultaneously.

FIG. 19 shows the results of the simulation performed on the GCDD system and the conventional system in the ITU pedestrian-A channel.

20 shows the results of the simulation in a typical urban (6-ray) environment.

21 is a diagram of an exemplary transceiver structure applying a precoding matrix for transmission power allocation according to an embodiment of the present invention.

22 is an exemplary transceiver structure diagram applying a precoding matrix for transmission power allocation according to an embodiment of the present invention.

23 is an exemplary transceiver structure diagram applying a precoding matrix for transmission power allocation according to an embodiment of the present invention.

FIG. 24 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 21 or FIG.

FIG. 25 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 22.

FIG. 26 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 21 or FIG.

FIG. 27 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 23.

FIG. 28 is an exemplary transceiver structure diagram of applying pilot symbols to the embodiment shown in FIG. 21 or FIG.

Claims (16)

In the data transmission method in a multi-antenna system using a plurality of subcarriers, Spatially processing a transmission symbol corresponding to each subcarrier in a frequency domain; Converting the spatially processed transmission signal into a time domain; Multiplying the entirety of the transmission signal converted into the time domain by a predetermined weight; And Applying a predetermined cyclic delay to a transmission signal for each antenna Data transmission method using a cyclic delay comprising a. The method of claim 1, And adding a first pilot symbol corresponding to each antenna to the spatially processed transmission signal. The method according to claim 1 or 2, And adding a second pilot symbol transformed into a time domain for each antenna to the cyclic delayed transmission signal. A data transmission method in a multiple antenna system using a plurality of subcarriers, Performing precoding in a frequency domain on transmission symbols corresponding to each subcarrier; Converting the precoded transmission signal into a time domain; Applying a predetermined cyclic delay to a transmission signal for each antenna Data transmission method using a cyclic delay comprising a. The method of claim 4, wherein And adding a first pilot symbol corresponding to each antenna to the precoded transmission signal. The method according to claim 4 or 5, And adding a second pilot symbol transformed into a time domain for each antenna to the cyclic delayed transmission signal. In the data transmission method in a multi-antenna system using a plurality of subcarriers, Determining a phase shift based precoding matrix by multiplying a first matrix for phase shift and a second matrix for unitizing the first matrix; Multiplying the determined phase shift based precoding matrix by the transmission symbols corresponding to each subcarrier; Converting the transmission signal on which the phase shift based precoding is performed into a time domain; And Applying a predetermined cyclic delay to a transmission signal for each antenna Data transmission method using a cyclic delay comprising a. The method of claim 7, wherein And multiplying all of the transmission symbols converted into the time domain by a predetermined weight. The method of claim 8, And adding a first pilot symbol corresponding to each antenna to the phase shift based precoded transmission signal. The method according to any one of claims 7 to 9, And adding a second pilot symbol transformed into a time domain for each antenna to the cyclic delayed transmission signal. The method according to any one of claims 7 to 9, The result of multiplying the first matrix and the second matrix,

The phase angles θ _i (t), i = 1, ..., N _t and the second matrix of the first matrix are changed by a predetermined time unit. In the data transmission method in a multi-antenna system, Performing spatial processing associated with the multiple antennas on each of the transmission streams; Performing precoding for transmit power allocation so that each signal generated by the spatial processing is transmitted through at least one of the multiple antennas; Converting each signal generated by the precoding into a signal in a time domain for each antenna; And Transmitting the converted time domain signal via one or more of the multiple antennas Data transmission method comprising a. The method of claim 12, The data transmission method Applying phase shift diversity to each signal generated by the spatial processing; And Applying cyclic delay diversity to each of the transformed time domain signals And at least one of the data transmission method. The method of claim 13, And a large cyclic delay value is applied for the phase shift diversity and a small cyclic delay value is applied for the cyclic delay diversity. The method according to any one of claims 12 and 13, The data transmission method Adding a first pilot symbol for each antenna for each signal generated by the spatial processing; Adding a second pilot symbol for each antenna for each signal generated by the precoding; And And adding at least one third pilot symbol converted into a time domain for each signal to which the cyclic delay diversity is applied, for each antenna. The method of claim 12, The transmission power allocation precoding is performed by multiplying a unit matrix of N _t × N _t , where N _t is the number of the multiple antennas.

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