KR102449737B1 - Method and apparatus for tuning finite impulse response filter in in-band full duplex transceiver - Google Patents

Abstract

A method and apparatus for tuning an FIR filter in a co-band full-duplex transceiver are disclosed. The tuning method includes the steps of estimating input information of the FIR filter after setting the degree of attenuation of the FIR filter to a first value, estimating a delta response using the estimated input information of the FIR filter, and the estimated delta response and updating the degree of attenuation of the FIR filter to a second value using .

Description

METHOD AND APPARATUS FOR TUNING FINITE IMPULSE RESPONSE FILTER IN IN-BAND FULL DUPLEX TRANSCEIVER

The present invention relates to a method and apparatus for tuning an FIR filter in a co-band full-duplex transceiver.

Currently, most of the wireless communication systems employ a half-duplex method. Since the half-duplex method transmits or receives by distributing time or frequency, orthogonality may be maintained between transmission and reception. However, this half-duplex method wastes resources (time or frequency).

The in-band full duplex (IFD) method is proposed as a solution to solve the inefficiency of the half-duplex method. The same-band full-duplex method is a technology that enables simultaneous transmission and reception in the same band. Since the same-band full-duplex method can theoretically increase the link capacity up to 2 times, it is an indispensable technology to achieve 1000 times the traffic capacity required for 5G mobile communication.

However, the same-band full-duplex (IFD) method has a disadvantage in that the self-transmission signal is introduced into the receiver, and the self-transmission signal is much stronger than the effective reception signal, which causes the self-transmission signal to act as a self-interference signal. For self-interference cancellation (SIC), there is an antenna area SIC technology that physically separates a transmit antenna and a receive antenna significantly. A technology for lowering the self-interference level through the antenna domain SIC technology and removing the remaining self-interference in the digital domain is in the commercialization stage. However, this technique is not easy to apply to small devices due to the physical separation between the transmitting and receiving antennas.

In order to apply the same-band full-duplex (IFD) method to a small device, a technique for eliminating magnetic interference in the analog circuit area without the aid of physical separation between antennas is required. SIC technology in analog circuit area is largely divided into passive SIC technology and active SIC technology. Passive SIC technology can easily obtain SIC gain using passive elements, but there is a limit to the gain. On the other hand, active SIC technology uses an adaptive analog FIR (Finite Impulse Response) filter circuit to achieve high SIC technology gains compared to passive SIC technology. However, the existing active SIC technology has a problem in that it cannot continuously maintain a high SIC gain while rapidly adapting to changes in the surrounding environment over a wide band.

An object of the present invention is to provide a method and apparatus for tuning an FIR filter so that a same-band full-duplex transceiver has a high SIC gain.

According to an embodiment of the present invention, there is provided a method of tuning the FIR filter in a co-band full-duplex transceiver including a finite impulse response (FIR) filter that receives a transmission signal and removes a self-transmission interference signal. The method includes the steps of setting the attenuation degree of the FIR filter to a predetermined first value, estimating input information of the FIR filter, estimating a delta response using the input information of the FIR filter, the The method may include updating the degree of attenuation to a second value using a delta response, and applying the second value to the FIR filter.

The delta response may correspond to a value obtained by subtracting a baseband equivalent time impulse response for the FIR filter set to the first value from a baseband equivalent time impulse response for a radio channel.

The updating may include calculating a third value that is an attenuation degree of the FIR filter to be updated using the delta response, and calculating the second value by adding the third value to the first value. may include steps.

The step of estimating the input information of the FIR filter includes, in a first period, setting the degree of attenuation to the first value, and then generating and transmitting a first training signal, in the first period, the self-transmission Measuring a first signal that is a baseband equivalent signal for a signal obtained by subtracting the output signal of the FIR filter from the interference signal, generating and transmitting a second training signal in a second period, in the second period, the The method may include measuring a second signal that is a baseband equivalent signal with respect to the self-transmission interference signal, and estimating input information of the FIR filter using the first signal and the second signal.

The estimating of the delta response may include estimating the delta response using the input information of the FIR filter and the first signal.

The step of estimating the input information of the FIR filter includes subtracting the first signal from the second signal to generate a third signal that is a baseband equivalent signal with respect to the output signal of the FIR filter, and using the third signal Thus, it may include calculating the input information of the FIR filter.

The measuring of the second signal may include measuring the second signal after suppressing generation of an output signal of the FIR filter.

The first training signal and the second training signal may be the same signal.

The input information of the FIR filter may include a linear component and a nonlinear component.

The first period and the second period may be included in a field for transmitting the first training signal and the second training signal in a time domain frame.

According to another embodiment of the present invention, a co-band full-duplex transceiver is provided. The same-band full-duplex transceiver includes a transmitter that generates and transmits a transmit signal, a receiver that converts a received signal into a baseband signal and demodulates, a finite impulse response (FIR) filter that receives the transmit signal and removes a self-transmission interference signal; and an estimator for estimating input information of the FIR filter using the output of the analog-to-digital converter included in the receiver, and setting an attenuation degree of the FIR filter using the input information of the FIR filter.

The estimator sets the degree of attenuation to a predetermined first value, estimates input information of the FIR filter, estimates a delta response using input information of the FIR filter, and attenuates the attenuation using the delta response The degree may be updated to a second value, and the second value may be applied to the FIR filter.

The delta response may correspond to a value obtained by subtracting a baseband equivalent time impulse response for the FIR filter set to the first value from a baseband equivalent time impulse response for a radio channel.

The transmitter may generate and transmit a first training signal in a first period and generate and transmit a second training signal in a second period, and the estimator may set the degree of attenuation as the first value in the first period. After setting, the first signal may be measured using the output signal of the analog-to-digital converter, and in the second period, the second signal, which is a baseband equivalent signal for the self-transmission interference signal, may be measured.

The estimator subtracts the first signal from the second signal to generate a third signal that is a baseband equivalent signal to the output signal of the FIR filter, and calculates input information of the FIR filter using the third signal. can

The estimator may set an attenuation degree of the FIR filter in the second period to suppress an output signal of the FIR filter.

The co-band full-duplex transceiver may further include an antenna for transmitting the transmission signal and receiving the received signal, and a splitter for transmitting the transmission signal to the antenna and transmitting the received signal to the receiver.

The transmission signal may be an output signal of a power amplifier included in the transmitter.

The first training signal and the second training signal may be the same signal.

The input information of the FIR filter may include a linear component and a nonlinear component.

According to an embodiment of the present invention, by setting the degree of attenuation of the FIR filter by using the baseband equivalent signal for the input signal of the FIR filter, it is possible to obtain a high SIC gain while quickly adapting to a change in the surrounding environment over a wide band.

1 is a diagram illustrating a same-band full-duplex transceiver according to an embodiment of the present invention.
2 is a diagram illustrating a same-band full-duplex transceiver according to another embodiment of the present invention.
3 is a diagram illustrating a time domain frame structure according to an embodiment of the present invention.
4 is a flowchart illustrating a method of estimating input information of an FIR filter according to an embodiment of the present invention.
5 is a flowchart illustrating a method for an estimator to tune an FIR filter according to an embodiment of the present invention.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings so that those of ordinary skill in the art to which the present invention pertains can easily implement them. However, the present invention may be embodied in several different forms and is not limited to the embodiments described herein. And in order to clearly explain the present invention in the drawings, parts irrelevant to the description are omitted, and similar reference numerals are attached to similar parts throughout the specification.

Throughout the specification, a terminal is a mobile terminal (MT), a mobile station (MS), an advanced mobile station (AMS), a high reliability mobile station (HR-MS) ), a subscriber station (subscriber station, SS), a portable subscriber station (PSS), an access terminal (AT), user equipment (UE), and the like, and may refer to a terminal, MT, It may include all or some functions of AMS, HR-MS, SS, PSS, AT, UE, and the like.

In addition, the base station (base station, BS) is an advanced base station (advanced base station, ABS), a high reliability base station (high reliability base station, HR-BS), a Node B (node B), an advanced node B (evolved node B, eNodeB), an access point (AP), a radio access station (RAS), a base transceiver station (BTS), a mobile multihop relay (MMR)-BS, a relay serving as a base station station, RS), may refer to a high reliability relay station (HR-RS) that serves as a base station, etc., and may refer to ABS, NodeB, eNodeB, AP, RAS, BTS, MMR-BS, RS, HR It may include all or part of functions such as -RS.

Throughout the specification, a transceiver is a terminal, a mobile terminal (MT), a mobile station (MS), an advanced mobile station (AMS), a high reliability mobile station , HR-MS), a subscriber station (subscriber station, SS), a portable subscriber station (PSS), an access terminal (AT), may refer to a user equipment (UE), etc., It may include all or part of functions such as terminal, MT, AMS, HR-MS, SS, PSS, AT, and UE.

In addition, the transceiver (transceiver) is a base station (base station, BS), an advanced base station (advanced base station, ABS), high reliability base station (high reliability base station, HR-BS), Node B (node B), advanced Node B (evolved node B, eNodeB), access point (AP), radio access station (RAS), base transceiver station (BTS), MMR (mobile multihop relay)-BS, base station role It may also refer to a relay station (RS), a high reliability relay station (HR-RS), etc. that perform the role of a base station, ABS, NodeB, eNodeB, AP, RAS, BTS, MMR- It may include all or some functions of BS, RS, HR-RS, and the like.

1 is a diagram illustrating a same-band full-duplex transceiver 1000 according to an embodiment of the present invention.

1, the same-band full-duplex transceiver 1000 according to an embodiment of the present invention includes a baseband digital domain transceiver 100, an analog circuit domain transceiver 200, and an antenna unit 300. include

The baseband digital domain transceiver 100 includes a transmission data generator 110 and an estimator 120 . Although not shown in FIG. 1 , the baseband digital domain transceiver 100 may further include a reception data demodulation generator. The transmission data generator 110 generates a digital signal corresponding to data to be transmitted by performing encoding and modulation. In FIG. 1, an output signal of the transmission data generator 110, that is, a baseband time domain signal, is represented by x[m]. The received data demodulation generator (not shown in FIG. 1) performs demodulating and decoding on the received data. The estimator 120 sets the attenuation degree for each tap of the finite impulse response (FIR) filter 240 . The estimator 120 according to an embodiment of the present invention performs a function of estimating input information of the FIR filter 240 in order to set the attenuation degree for each tap of the FIR filter 240 . In addition, the estimator 120 according to the embodiment of the present invention may remove the SIC by setting the attenuation degree for each tap of the FIR filter 240 using the estimated input information of the FIR filter 240 .

The analog circuit area transceiver 200 includes a digital analog converter (DAC) 210 , a mixer 220 , a power amplifier (PA) 230 , an FIR filter 240 , and a signal combiner 250 . ), a low noise amplifier (LNA) 260 , an integrator 270 and an analog digital converter (ADC) 280 . The transmission data generator 110 , the DAC 210 , the mixer 220 , and the PA 230 form a transmitter. In addition, the signal combiner 250 , the LNA 260 , the integrator 270 , the ADC 280 and the reception data demodulation generator form a receiver.

The DAC 210 converts a digital signal into an analog signal, and the mixer 220 converts a baseband frequency signal into a radio frequency (RF) band signal using the carrier frequency f _C . The PA 230 amplifies and outputs the RF band signal. In FIG. 1 , the output signal of the DAC 210 is represented by x _b (t), and the output signal of the PA 230 , that is, the final transmission signal, is represented by s (t). The final transmission signal s(t) contains nonlinear components (harmonics component and phase noise component) due to the mixer 220 and the PA 230, and as described below, an estimator according to an embodiment of the present invention (120) estimates baseband equivalent information for s(t) including nonlinear components.

The antenna unit 300 includes a transmit antenna 310 and a receive antenna 320 . The transmit antenna 310 transmits a transmit signal s(t) to the outside, and the receive antenna 320 receives the receive signal. Since the transmission antenna 310 and the reception antenna 320 are physically close to each other, the transmission signal s(t) flows in through the reception antenna 320 and acts as an interference signal, and this interference signal is a self-transmission interference signal. This is because the transceiver 1000 according to the embodiment of the present invention operates in the same-band full-duplex mode. The signal received through the reception antenna 320 includes not only a normal reception signal but also a self-transmission interference signal. In FIG. 1, the self-transmission interference signal is represented by y(t). As will be described below, in the embodiment of the present invention, the self-transmission interference signal y(t) may be removed by using the FIR filter 240 .

The FIR filter 240 receives the transmission signal s(t) and generates and outputs a signal that minimizes the self-transmission interference signal y(t). As shown in Figure 1, the FIR filter 240 according to the embodiment of the present invention is a plurality of delays (d ₀ ~ d _N-1 ), a plurality of attenuators (attenuator) (a ₀ ~ a _N-1 ) and a signal A coupler 241 is included.

Each of the plurality of delays d ₀ to d _N-1 has a fixed delay. The delay intervals between each delayer d _i (i=0,1,...,N-1) may be the same or all different, and may be divided into a plurality of groups having the same delay interval. Here, N denotes the total number of taps, and d _i (i=0,1,2,...,N-1) denotes a delay applied to the taps.

A plurality of attenuators (a ₀ ~ a _N-1 ) is respectively connected to the plurality of delay (d ₀ ~ d _N-1 ) attenuates the signal. The attenuation degree of each attenuator a _i (i=0,1,2...,N-1) is variable, and the attenuation degree is set by the estimator 120 . The value of the attenuation degree of each attenuator a _i (i=0,1,2...,N-1) may have one of magnitude, magnitude and phase, real value or complex value, and in the following description, attenuation for convenience The precision is explained assuming that it has a magnitude or a real value.

The signal combiner 241 combines the output signals of the plurality of attenuators a ₀ to a _N-1 . The signal combiner 241 adds and combines all of the signals output from the plurality of attenuators a ₀ to a _N-1 . In FIG. 1 , the output signal of the signal combiner 241 , that is, the output signal of the FIR filter 240 is represented by z(t).

The signal combiner 250 combines the received signal with the output signal of the FIR filter 240 and outputs it to the LNA 260 . The signal combiner 250 subtracts the signal output from the FIR filter 240 from the received signal and then combines the two signals. At this time, since the FIR filter 240 outputs a signal that minimizes the self-transmission interference signal y(t), the signal combiner 250 removes the self-transmission interference signal y(t) from the received signal. (260) is output. In FIG. 1, the output signal of the signal combiner 250 is represented by v(t).

The LNA 260 receives a received signal from which the self-transmission interference signal y(t) is removed from the signal combiner 250 , and removes noise from the input signal and amplifies the received signal. The integrator 270 converts the RF band signal into a baseband signal using the carrier frequency f _C . The ADC 260 converts an analog baseband signal into a digital signal.

2 is a diagram illustrating a same-band full-duplex transceiver 1000' according to another embodiment of the present invention.

2, the same-band full-duplex transceiver 1000' according to another embodiment of the present invention includes a baseband digital domain transceiver 100, an analog circuit domain transceiver 200', and an antenna unit ( 300'). Since the antenna unit 300 ′ is the same as that of FIG. 1 except that the antenna unit 300 ′ is composed of one transmitting and receiving antenna and the analog circuit area transmitting and receiving unit 200 ′ further includes a divider 290 , the description overlapping with FIG. 1 will be omitted. .

The antenna unit 300' includes a transmission/reception antenna 330 that simultaneously performs a reception function as well as a transmission function for the same-band full-duplex method. That is, a transmission signal is transmitted and a reception signal is received through the antenna 330 .

The analog circuit area transceiver 200 ′ further includes a divider 290 . The splitter 290 is connected to the antenna 330 and sends a transmit signal s(t) to the antenna 330 . And the splitter 290 transmits the received signal received from the antenna 330 to the receiving module (signal combiner 250, LNA 260, etc. in FIG. 1). That is, the splitter 290 according to an embodiment of the present invention serves to transmit a transmission signal to the antenna 330 and a reception signal to a reception module. The distributor 290 may be implemented as a circulator or an EBD (Electrical Balance Duplex). Circulators and EBDs are known to those of ordinary skill in the art to which the present invention pertains, and detailed descriptions thereof will be omitted.

First, a method for the estimator 120 to estimate input information of the FIR filter 240 will be described. The input information z(t) of the FIR filter 240 includes a nonlinear component, and the estimator 120 according to an embodiment of the present invention includes baseband equivalent information of s(t) including the nonlinear component. to estimate A method of estimating baseband equivalent information of z(t) described below will be described with reference to FIG. 2 , but may be equally applied to FIG. 1 .

In order to explain how the estimator 120 estimates the input information of the FIR filter 240 , various signals of the same-band full-duplex transceiver 1000 ′ are mathematically modeled.

x[m] denotes a baseband time-domain signal oversampled by d times. If d=1, x[m] means a baseband sampled time domain signal. When the sampled signal passes through the DAC 210, a baseband analog signal x _b (t) as shown in Equation 1 below is generated.

In Equation 1, x[m] is x _b (m/W), and sinc(t) is defined as in Equation 2 below.

)의 선형 결합(Linear Combination)으로 표현할 수 있다는 샘플링 이론이 수학식 1에 적용된다. 수학식 1에서, W는 대역폭을 의미하며, 기저대역의 시스템 대폭일 수도 있고 d배 오버샘플링(oversampling)된 대역폭일 수 있다. 아래에서는 설명의 편의상 W를 1배(d=1) 오버샘플링된 대역폭으로 가정하여 설명하였으나, d의 값은 다른 값으로 설정될 수 있다. The expression of Equation 1 is that the sampling theory is applied. That is, all baseband waveforms band-limited to W/2 have a coefficient (i.e., x[m]) given by the samples and an orthogonal basis (

), the sampling theory that can be expressed as a linear combination of ) is applied to Equation 1. In Equation 1, W means a bandwidth, and may be a system wide baseband or a bandwidth oversampled d times. In the following description, it is assumed that W is a bandwidth oversampled by one time (d=1) for convenience of description, but the value of d may be set to another value.

When x _b (t) passes through the mixer 220 , it becomes an RF signal, and when this RF signal passes through the PA 230 , it becomes a time domain input signal s(t) of the FIR filter 240 . If this s(t) is expressed mathematically, it becomes as in Equation 3 below.

는 x_b(t)가 PA(230)를 통과하면서 발생되는 비선형성분을 포함하는 RF 신호에 대한 등가 기저대역 신호를 의미한다. P는 PA(230)에 의해 증폭되는 송신전력을 의미하며, f_c는 믹서(220)의 반송 주파수를 의미한다. 수학식 3를 통해, s(t)는 선형성분과 비선형성분을 모두 포함하고 있음을 알 수 있다. in Equation 3

denotes an equivalent baseband signal with respect to an RF signal including a nonlinear component generated while x _b (t) passes through the PA 230 . P denotes transmit power amplified by the PA 230 , and f _c denotes a carrier frequency of the mixer 220 . From Equation 3, it can be seen that s(t) includes both a linear component and a nonlinear component.

The RF signal s(t) as shown in Equation 3 passes through the FIR filter 240, and the time domain expression of the passed RF signal becomes as shown in Equation 4 below.

In Equation 4, h _fir (t) means the RF domain time impulse response of the FIR filter 240 . The baseband equivalent time impulse response for h _fir (t) is expressed in Equation 5 below.

In Equation 5, d _i (i=0, 1, …, N-1) and a _i (i=0, 1, …, N-1) are time delays for each tap of the FIR filter 240, respectively. ) and path attenuation. And l means the index of the filter tab. The digital (baseband) domain equivalent signal z[m] for z(t), which is the output of the FIR filter 240, is obtained by the following Equation 6 when Equation 5 is applied.

은 수학식 3에 의해

이 되며, M은 시간영역의 최대 샘플수를 의미한다. In Equation 6,

is by Equation 3

and M means the maximum number of samples in the time domain.

Using the expressions of Equations 4 to 6, the digital (baseband) domain equivalent time domain expression (y[m]) for the self-transmission interference signal y(t) is obtained by the following Equations 7 and become together

은 분배기(290)와 안테나(330)의 특성에 의해 생성되는 채널에 대한 기저대역(디지털영역) 등가 시간임펄스 응답을 의미한다. 그리고,

와

는 각각 분배기(290)와 안테나(330)의 특성에 의해 생성되는 채널에 대한 다중 경로 i별 경로감쇄(attenuation)와 시간지연(time delay)를 의한다. N'는 채널에 대한 총 다중 경로 수를 의미한다. In Equation 7,

denotes a baseband (digital domain) equivalent time impulse response for a channel generated by the characteristics of the divider 290 and the antenna 330 . and,

Wow

, respectively, are due to path attenuation and time delay for each multi-path i for the channel generated by the characteristics of the splitter 290 and the antenna 330 . N' means the total number of multi-paths for the channel.

A method for estimating input information of the FIR filter 240 by the estimator 120 according to an embodiment of the present invention will be described with reference to FIGS. 3 and 4 . In the following description, it is assumed that the LNA 260, the integrator 270, and the ADC 280 have no hardware damage in each of Equations 5 to 7 above.

3 is a diagram illustrating a time domain frame structure according to an embodiment of the present invention.

As shown in FIG. 3 , the time domain frame according to the embodiment of the present invention includes an In-band Full Duplex (IFD) training field 410 and an IFD communication field 420 . The IFD training field 410 is a separate field for transmitting a training signal to estimate input information of the FIR filter 240 . In the IFD training field 410 , a first training signal 411 and a second training signal 412 are transmitted. The first training signal 411 and the second training signal 412 are separately generated signals for estimating input information of the FIR filter 240, and the two signals may be allocated in the frequency domain or may be allocated in the time domain. have. Meanwhile, the first training signal 411 and the second training signal 412 are the same signal. In addition, the IFD communication field 420 is a section in which data for same-band full-duplex transmission/reception is transmitted.

4 is a flowchart illustrating a method of estimating input information of the FIR filter 240 according to an embodiment of the present invention.

First, the estimator 120 measures v[m] after self-interference cancellation (SIC) is performed using the first training signal 411 (S410). The estimator 120 sets the degree of attenuation (a _i (i=0,1,2...,N-1)) of the FIR filter 240 as an arbitrary initial setting value (corresponding to a (u) of Equation 22). set And the transmission data generation unit 110 generates a first training signal 411, the estimator 120 is the output information (v[m]) of the signal combiner 250 for the first training signal 411 measure Here, v[m] is a signal measured using the output of the ADC 280 with respect to the first training signal 411 , and the estimator 120 receives the output of the ADC 280 and calculates v[m]. measure Meanwhile, using Equations 6 and 7, the baseband equivalent signal v[m] for the output signal v(t) of the signal combiner 250 becomes as Equation 8 below.

The estimator 120 measures the baseband equivalent signal y[m] of the self-interference signal using the second training signal 412 ( S420 ). That is, the estimator 120 measures Equation 7, which is the baseband equivalent signal y[m] with respect to the self-transmission interference signal y(t).

In order for the estimator 120 to measure y[m], only the self-transmission interference signal y(t) needs to flow into the estimator 120 . As a method in which the self-transmission interference signal y(t) flows into the estimator 120 , the estimator 120 determines the degree of attenuation (a _i (i=0,1,2...,N) of the FIR filter 240 . -1)) can be set to a very high value. Through this, it is possible to physically suppress the generation of the output signal of the FIR filter 240, and only the self-transmission interference signal y(t) is passed through the LNA 260, the integrator 270 and the ADC 280 to the estimator 120 ) is entered in At this time, the estimator 120 measures the signal input from the ADC 280, and the measured signal corresponds to y[m].

Next, the estimator 120 estimates the input information of the FIR filter 240 (S430). If Equation 8 is subtracted from Equation 7, the baseband equivalent signal z[m] of the output signal z(t) of the FIR filter 240 expressed as Equation 6 is obtained. That is, the estimator 120 may estimate z[m] by subtracting v[m] measured in step S410 from y[m] measured in step S420 . In order to estimate the input information of the FIR filter 240 from the thus estimated z[m], the matrix expression for Equation 6 is obtained, as shown in Equation 9 below.

는

의 토프리즈(Toeplitz) 행렬이며, 아래의 수학식 10과 같이 표현된다. In Equation 9 above,

Is

It is a Toeplitz matrix of , and is expressed as Equation 10 below.

In Equation 10, c denotes the number of non-causal elements (ie, samples).

은 수학식 5에 대한 벡터 표현식으로, 아래의 수학식 11과 같이 표현된다. In Equation 9,

is a vector expression for Equation 5, and is expressed as Equation 11 below.

In Equation 11, the symbol T means transpose of a vector. As described in step S410, the time delay value (d ₀ ~ d _N-1 ) and the degree of attenuation (a _i (i=0,1,2...,N-1) for each tap of the FIR filter 240 are arbitrary initial values. Since it is set to a value, the baseband time impulse response vector as shown in Equation 11 can be calculated and obtained, that is, the estimator 120 knows the baseband time impulse response vector as shown in Equation 11 in advance.

Using the vector z of Equation 9 estimated as described above and Equation 11 obtained in advance, the baseband equivalent matrix for the input information of the FIR filter 240 is estimated as Equation 12 below.

는

의 의사역행렬(pseudo-inverse matrix)을 의미하고,

는

의 허미션(Hermitian)을 의미한다. In Equation 12,

Is

means a pseudo-inverse matrix of

Is

means Hermitian.

을 추출한다. 이와 같이 추출된

이 추정부(120)가 구하고자 하는 FIR 필터(240)의 입력 신호에 대한 기저대역 등가신호이다. 그리고 상기에서 설명한 바와 같이,

에는 선형성분에 대한 정보 뿐만 아니라 비선형선헝분에 대한 정보도 포함되어 있다. Finally, the estimator 120 determines the equivalent baseband component of the input signal of the FIR filter 240 in the matrix estimated in Equation (12).

to extract extracted in this way

This is a baseband equivalent signal to the input signal of the FIR filter 240 that the estimator 120 wants to obtain. And as explained above,

contains information on nonlinear linear components as well as information on linear components.

을 이용하여 자기간섭신호를 최소화 시키기 위해, FIR 필터(240)의 감쇄 정도(a_i(i=0,1,2…,N-1))를 재설정한다. 즉, 추정부(120)는 S410 단계에서 초기 값으로 설정한 감쇄 정도를 자기간섭신호를 최소화 시키기 위한 값으로 튜닝한다. 아래에서는 도 5를 참조하여, 본 발명의 실시예에 따른 추정부(120)가 FIR 필터(240)를 튜닝하는 방법에 대해서 설명한다. On the other hand, the estimator 120

In order to minimize the self-interference signal using , the degree of attenuation (a _i (i=0,1,2...,N-1)) of the FIR filter 240 is reset. That is, the estimator 120 tunes the degree of attenuation set as the initial value in step S410 to a value for minimizing the self-interference signal. Hereinafter, a method for the estimator 120 to tune the FIR filter 240 according to an embodiment of the present invention will be described with reference to FIG. 5 .

5 is a flowchart illustrating a method for the estimator 120 to tune the FIR filter 240 according to an embodiment of the present invention.

First, the estimator 120 according to an embodiment of the present invention estimates a delta response (S510). v[m] of Equation 8 can be expressed as Equation 13 below.

이 추정부(120)가 추정하고자 하는 델타 응답이다.

은

이며 상기 수학식 5를 참조하면 아래의 수학식 14와 같이 표현된다.

는 무선 채널에 대한 기저대역 등가 시간임펄스응답을 의미하며,

는 상기 수학식 5와 동일하다. In Equation 13,

This is the delta response that the estimator 120 wants to estimate.

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and, referring to Equation 5, it is expressed as Equation 14 below.

is the baseband equivalent time impulse response for the radio channel,

is the same as in Equation 5 above.

는 i번째 감쇄기의 업데이트 감쇄정도를 의미한다. 상기 수학식 13에 대한 행렬 표현식을 구하면, 아래의 수학식 15와 같다. In Equation 14 above,

is the update attenuation degree of the i-th attenuator. The matrix expression for Equation 13 is obtained as Equation 15 below.

는 상기 수학식 10과 동일하며, 도 4에서 이미 구한 값이다. 즉,

는 상기 수학식 12에서 같이 추정된 행렬에 해당한다.

은 FIR 필터(240)의 시간임펄스응답 특성을 업데이트 하기 위해, 현재의 기저대역 등가 시간임펄스응답에 더해지는 델타응답 벡터를 의미한다.

는 아래의 수학식 16과 같이 표현된다. In Equation 15,

is the same as in Equation 10, and is a value already obtained in FIG. 4 . in other words,

corresponds to the matrix estimated as in Equation 12 above.

is a delta response vector added to the current baseband equivalent time impulse response in order to update the time impulse response characteristic of the FIR filter 240 .

is expressed as in Equation 16 below.

or

에 대한 의사 역행렬을 취하면,

는 아래의 수학식 17과 같이 추정된다. In Equation 15 above,

Taking the pseudo-inverse for

is estimated as in Equation 17 below.

는

의 의사역행렬을 의미한다.

는 상기 수학식 12에서와 같이 이미 추정해서 알고 있는 행렬이므로,

는 계산에 의해서 획득될 수 있다. 그리고 v도 이미 S410 단계에서 제1 트레이닝 신호를 통해서 측정하여 구해진 값이다. 따라서, 추정하고자 하는 델타 응답(

)는 상기 수학식 17을 통해 추정된다. In Equation 17 above,

Is

means the pseudo-inverse of

is a matrix that is already estimated and known as in Equation 12,

can be obtained by calculation. And v is already a value obtained by measuring through the first training signal in step S410. Therefore, the delta response (

) is estimated through Equation 17 above.

)를 계산한다(S520). 상기 수학식 14의 델타응답 표현식으로부터 FIR 필터(240)의 업데이트 감쇄 정도(

)를 계산하기 위해, 상기 수학식 17은 아래의 수학식 18과 같이 표현될 수 있다. Next, the estimator 120 according to the embodiment of the present invention updates the degree of attenuation (

) is calculated (S520). From the delta response expression of Equation 14, the update attenuation degree of the FIR filter 240 (

), Equation 17 may be expressed as Equation 18 below.

In Equation 18, s is expressed as Equation 19 below.

or

In s as in Equation 19, fc is a known value as a carrier frequency, d ₀ to d _N-1 are preset values as a fixed delay value, and W is a known value as a bandwidth. Accordingly, s can be obtained by calculating through Equation 19 above.

는 아래의 수학식 20과 같이 표현된다. And in Equation 18,

is expressed as in Equation 20 below.

는 S510단계에서 추정해서 구한 벡터이고, s도 상기 수학식 19에 의해 계산된 값으로서 이미 알고 있으므로,

는 아래의 수학식 21과 같이 구해진다. In Equation 18 above

is a vector obtained by estimating in step S510, and since s is already known as a value calculated by Equation 19,

is obtained as in Equation 21 below.

or

를 이용하여 FIR 필터(240)의 감쇄 정도를 튜닝한다(S530). 상기 수학식 21과 같이 구한

를 이용하여, 추정부(120)는 업데이트할 감쇄 정도(a(u+1))를 아래의 수학식 22와 같이 결정한다. Finally, the estimator 120 according to the embodiment of the present invention

to tune the degree of attenuation of the FIR filter 240 ( S530 ). obtained as in Equation 21

Using , the estimator 120 determines the degree of attenuation to be updated ( a (u+1)) as shown in Equation 22 below.

In Equation 22, a (u) denotes a vector of a current attenuation degree and corresponds to an arbitrary initial setting value described in step S410 of FIG. 4 . And a (u+1) denotes a vector of the degree of attenuation to be finally updated.

The FIR filter 240 applies the attenuation degree to be updated (( a (u+1)) determined as in Equation 22 to the attenuator of the FIR filter 240. Through this, the estimator ( 120) can minimize the self-interference signal.

According to this embodiment of the present invention, by setting the degree of attenuation of the FIR filter 240 using a baseband equivalent signal for the input signal of the FIR filter 240 including a nonlinear component as well as a linear component, a high SIC gain can be obtained.

Although the embodiments of the present invention have been described in detail above, the scope of the present invention is not limited thereto. is within the scope of the right.

Claims (20)

A method of tuning the FIR filter in the same-band full-duplex transceiver including a finite impulse response (FIR) filter that receives a transmission signal and removes a self-transmission interference signal, the method comprising:
After setting the degree of attenuation of the FIR filter to a predetermined first value, estimating input information of the FIR filter;
estimating a delta response using the input information of the FIR filter;
using the delta response, updating the degree of attenuation to a second value; and
applying the second value to the FIR filter;
The step of estimating the input information of the FIR filter,
In a first period, after setting the degree of attenuation to the first value, generating and transmitting a first training signal;
measuring, in the first period, a first signal that is a baseband equivalent signal for a signal obtained by subtracting an output signal of the FIR filter from the self-transmission interference signal;
in a second period, generating and transmitting a second training signal;
measuring, in the second period, a second signal that is a baseband equivalent signal for the self-transmission interference signal; and
and estimating input information of the FIR filter by using the first signal and the second signal. According to claim 1,
The delta response corresponds to a value obtained by subtracting a baseband equivalent time impulse response for the FIR filter set to the first value from a baseband equivalent time impulse response for a radio channel. According to claim 1,
The updating step is
calculating a third value that is an attenuation degree of the FIR filter to be updated using the delta response, and
calculating the second value by adding the third value to the first value. delete According to claim 1,
The estimating of the delta response includes estimating the delta response using the input information of the FIR filter and the first signal. According to claim 1,
The step of estimating the input information of the FIR filter,
generating a third signal that is a baseband equivalent signal for an output signal of the FIR filter by subtracting the first signal from the second signal; and
and calculating input information of the FIR filter by using the third signal. According to claim 1,
The measuring of the second signal includes measuring the second signal after suppressing generation of an output signal of the FIR filter. According to claim 1,
wherein the first training signal and the second training signal are the same signal. According to claim 1,
The input information of the FIR filter includes a linear component and a non-linear component. According to claim 1,
wherein the first period and the second period are included in a field for transmitting the first training signal and the second training signal in a time domain frame. a transmitter that generates and transmits a transmission signal;
a receiver that converts and demodulates the received signal into a baseband signal;
A Finite Impulse Response (FIR) filter that receives the transmission signal and removes a self-transmission interference signal, and
and an estimator for estimating input information of the FIR filter using the output of the analog-to-digital converter included in the receiver, and setting an attenuation degree of the FIR filter using the input information of the FIR filter,
The estimator sets the degree of attenuation to a predetermined first value, estimates input information of the FIR filter, estimates a delta response using input information of the FIR filter, and attenuates the attenuation using the delta response update the degree to a second value, and apply the second value to the FIR filter,
The transmitter generates and transmits a first training signal in a first period and generates and transmits a second training signal in a second period,
The estimator is
In the first period, after setting the degree of attenuation to the first value, a first signal is measured using the output signal of the analog-to-digital converter,
In the second period, the same-band full-duplex transceiver measures a second signal that is a baseband equivalent signal with respect to the self-transmission interference signal. delete 12. The method of claim 11,
The delta response corresponds to a value obtained by subtracting a baseband equivalent time impulse response for the FIR filter set to the first value from a baseband equivalent time impulse response for a radio channel. delete 12. The method of claim 11,
The estimator generates a third signal that is a baseband equivalent signal with respect to an output signal of the FIR filter by subtracting the first signal from the second signal, and calculates input information of the FIR filter using the third signal same-band full-duplex transceiver. 12. The method of claim 11,
The estimator sets an attenuation degree of the FIR filter in the second period to suppress an output signal of the FIR filter. 12. The method of claim 11,
An antenna for transmitting the transmission signal and receiving the received signal, and
The same-band full-duplex transceiver further comprising a splitter for sending the transmitted signal to the antenna and sending the received signal to the receiver. a transmitter that generates and transmits a transmission signal;
a receiver that converts and demodulates the received signal into a baseband signal;
A Finite Impulse Response (FIR) filter that receives the transmission signal and removes a self-transmission interference signal, and
and an estimator for estimating input information of the FIR filter using the output of the analog-to-digital converter included in the receiver, and setting an attenuation degree of the FIR filter using the input information of the FIR filter,
The transmission signal is a co-band full-duplex transceiver that is an output signal of a power amplifier included in the transmitter. 12. The method of claim 11,
The same-band full-duplex transceiver, wherein the first training signal and the second training signal are the same signal. 12. The method of claim 11,
The input information of the FIR filter includes a linear component and a nonlinear component.

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