KR102270836B1 - Low loss phase shifter using tunable transmission line - Google Patents

Abstract

It relates to a phase shifter, the phase shifter comprising: a pair of reflective loads spaced apart from each other; and a hybrid coupler disposed to connect the pair of reflective loads and input/output ports, wherein the reflective load may be formed by a series connection between a varactor diode and a transmission line.

Description

Low-loss phase shifter using tunable transmission line {LOW LOSS PHASE SHIFTER USING TUNABLE TRANSMISSION LINE}

The present application relates to a phase shifter. In particular, the present application relates to a low-loss phase shifter using a varactor and a series reflective load of a transmission line.

)를 조정하여 신호의 위상을 변화시킨다.A phase shifter is a device that delays an input signal in a given frequency band and outputs a signal by shifting the phase. The reflection type analog phase shifter mainly uses a hybrid coupler, and the reflection coefficient (

) to change the phase of the signal.

The phase shifter is an essential component for realizing a Massive MIMO (Massive Multi Input Multi Output,) system, and is used in beamforming and beam steering technology. Such a phase shifter is being used in a wide range, such as a base station of mobile communication, a transmitter of wireless power transmission, and a high-frequency system that operates a phased array antenna as well as a radar navigation device. In addition, the phase shifter may be used to manufacture an impedance converter for impedance matching that reduces reflection loss of an RF system.

In particular, the analog phase shifter has a continuous phase change compared to the digital phase shifter, and power consumption and manufacturing cost can be reduced, so that its application is high. For this application, the amount of phase shift should be 360° or more, and simplified fabrication and low loss characteristics are required.

In a reflective analog phase shifter, how to correct the reflection load determines the phase shift characteristics.

As an example, conventional document 1 [Chien-San Lin, Sheng-Fuh Chang and Wen-Chun Hsiao, "A Full-360 Reflection-Type Phase Shifter With Constant Insertion Loss", IEEE Trans. In Microwave Theory Tech, vol.18, No.2, pp.106-108, February, 2008], a reflective load was constructed by connecting a parallel resonant circuit in a cascade. In addition, prior document 2 [Artem R. Vilenski, Mikhail N.Makurin, Ekaterina I. Poshisholina, and Chongmin Lee, "Design Technique for Varactor Analog Phase Shifters with Equalized Losses", Progress In Electromagnetics Research C, Vol.86, pp. 1-16, July 2018] reduces the ripple of the insertion loss by connecting several varactors to the reflective load and controlling them individually.

In addition, Korean Patent No. 10-0366299 discloses a configuration in which a hybrid coupler and a reflective load including a compensating resistor are connected. Here, the reflective load is configured by connecting a series connection circuit of a varactor diode and an inductor in a cascade and then connecting a 1/4 transmission line therebetween.

In addition, Korean Patent Publication No. 10-1497018 proposes a method of manufacturing a compact phase shifter by configuring a hybrid coupler and a π-type reflective load. In addition, in U.S. Patent No. 4288763, a reflective load was configured using a varactor diode in a hybrid coupler to change the phase of 60°, and then connected to 6 stages to realize a phase of 360°. . In addition, in U.S. Patent No. 7969359, a reflective load is configured by connecting a varactor and an inductor in parallel to a hybrid coupler.

The reflective type (reflection type) analog phase shifter including the above-mentioned conventional documents commonly uses the resonance of the reflective load. The reflection type analog phase shifter uses a method that increases the phase range by causing LC resonance to change the reactance to infinity. However, this method has a problem in that current is concentrated as a resistive component and transmit power is inhibited due to resonance loss. That is, the conventional literature has a limit in minimizing the loss due to resonance in the reflective load.

Accordingly, it is required to develop a phase shifter (particularly, a reflective analog phase shifter) capable of adjusting a phase of 360° while minimizing a loss due to resonance in a reflective load.

The present application aims to solve the problems of the prior art described above, and to provide a phase shifter (in particular, a reflective low-loss phase shifter) capable of adjusting a phase of 360° while minimizing a loss due to resonance in a reflective load. do it with

However, the technical problems to be achieved by the embodiments of the present application are not limited to the technical problems as described above, and other technical problems may exist.

As a technical means for achieving the above technical problem, the phase shifter according to the first aspect of the present application, a pair of reflective loads spaced apart from each other; and a hybrid coupler disposed to connect the pair of reflective loads and input/output ports, wherein the reflective load may be formed by a series connection between a varactor diode and a transmission line.

In addition, the pair of reflective loads are arranged to be respectively connected to the remaining ports except for the ports connected to the input/output ports among the ports of the hybrid coupler, and by changing the DC voltage applied to the varactor diode, the varactor The capacitance of the diode may be changed so that a phase angle shift with respect to the output signal may be made.

In addition, the length of the transmission line may be set to a length such that a phase change of a reflection coefficient of the reflective load shifts a phase angle of 180° or more.

In addition, the length of the transmission line is set to λ/8, where λ may be a preset value according to a frequency.

As a technical means for achieving the above technical problem, the phase shifter according to the second aspect of the present application is a 360° phase shift capable of 360° phase shift by serially connecting the phase shifter according to the first aspect of the present application in two stages. it can be a gimmick

In addition, the 360° phase shifter may operate to reduce insertion loss through electronic control.

As a technical means for achieving the above technical problem, the phase shift method according to the third aspect of the present application is a phase shift method by a phase shifter according to the first aspect of the present application, (a) the hybrid in the phase shifter receiving, by a coupler, an input signal applied from an input port; and (b) outputting, by the hybrid coupler, an output signal whose phase is shifted from the input signal toward an output port connected to the hybrid coupler in response to the received applied signal, wherein the output signal is A signal generated by synthesizing the reflected signals reflected from each of the pair of reflective loads in the phase shifter in response to the applied input signal, but affected by the series connection between the varactor diode and the transmission line in the reflective load It could be a signal.

In this case, the phase shifter may include: a pair of reflective loads spaced apart from each other; and a hybrid coupler disposed to connect the pair of reflective loads and input/output ports, wherein the reflective load may be formed by a series connection between a varactor diode and a transmission line.

In addition, the pair of reflective loads are arranged to be respectively connected to the remaining ports except for the ports connected to the input/output ports among the ports of the hybrid coupler, and by changing the DC voltage applied to the varactor diode, the varactor The capacitance of the diode may be changed so that a phase angle shift with respect to the output signal may be made.

In addition, the length of the transmission line may be set to a length such that a phase change of a reflection coefficient of the reflective load shifts a phase angle of 180° or more.

In addition, the length of the transmission line is set to λ/8, where λ may be a preset value according to a frequency.

As a technical means for achieving the above technical problem, the computer program according to the fourth aspect of the present application may be stored in a recording medium to execute the phase shift method according to the third aspect of the present application.

The above-described problem solving means are merely exemplary, and should not be construed as limiting the present application. In addition to the exemplary embodiments described above, additional embodiments may exist in the drawings and detailed description.

According to the above-described problem solving means of the present application, by configuring the reflective load as a series connection between the varactor diode and the transmission line, the problem of the conventional reflective analog phase shifter (that is, the current is concentrated due to the resistance component, and the resonance loss It is possible to reduce the resonance loss caused by the resistor by solving the problem that transmit power is impaired).

According to the above-described problem solving means of the present application, by setting the length of the transmission line to λ/8, it is possible to control the phase with a transition angle of 180° or more with linearity.

According to the above-described problem solving means of the present application, it is possible to provide a phase shifter capable of 360° phase shift by serially connecting two phase shifters including a pair of reflective loads and a hybrid coupler.

However, the effects obtainable herein are not limited to the above-described effects, and other effects may exist.

1 is a block diagram showing a schematic configuration of a phase shifter according to an embodiment of the present application.
2 is a diagram illustrating a schematic configuration of a phase shifter according to an embodiment of the present application.
3 is a diagram illustrating an example of a layout photograph of a phase shifter according to an embodiment of the present application actually manufactured according to an experimental example of the present application.
4 is a diagram illustrating an example of a phase change according to a change in reactance of a reflective load in a phase shifter according to an embodiment of the present application.
5 is a diagram schematically showing a circuit diagram of a phase shifter according to another embodiment of the present application.
6 is a view showing an example of a layout photograph of a phase shifter according to another embodiment of the present application actually manufactured according to an experimental example of the present application.
7 is a view showing a change in power loss according to a DC voltage applied to a varactor diode in a phase shifter according to an embodiment of the present application for each frequency as a result according to an experimental example of the present application.
8 is a view showing a change in a phase of a reflection coefficient according to a DC voltage applied to a varactor diode in a phase shifter according to another embodiment of the present application for each frequency as a result according to an experimental example of the present application.
9 is a view showing an example of the result of improving the insertion loss of the phase shifter according to another embodiment of the present application through electronic control as a result according to an experimental example of the present application.
10 shows an example of a circuit diagram of a reflective load included in the phase shifter according to an embodiment of the present application.
11 is a view showing an insertion loss according to a change in a DC voltage applied to a varactor diode of a phase shifter according to an embodiment of the present application as a result according to an experimental example of the present application.
12 is a diagram illustrating a phase change according to a change in a DC voltage applied to a varactor diode of a phase shifter according to an embodiment of the present application as a result according to an experimental example of the present application.
13 is a diagram illustrating performance comparison between a phase shifter according to an embodiment of the present application and a conventional phase shifter as a result according to an experimental example of the present application.
14 is a diagram illustrating a change in input/output magnitude according to frequency (a) and an input/output phase change according to frequency (b) in the phase shifter according to an embodiment of the present application as a result according to an experimental example of the present application.
15 is a diagram illustrating performance comparison between a phase shifter according to another embodiment of the present application and a conventional phase shifter as a result according to an experimental example of the present application.
16 is a diagram illustrating a performance comparison between a phase shifter proposed in the present application and an ideal phase shifter as a result according to an experimental example of the present application.
17 is a view showing a Smith chart of a phase shifter manufactured according to the technology proposed in the present application as a result according to an experimental example of the present application.
18 is a diagram schematically illustrating a configuration of a phase shifter according to an embodiment of the present application.
19 shows an example of an equivalent circuit for the definitions of voltage and current (a) and the incremental length of a transmission line.
20 is a diagram for explaining the reason for connecting a varactor diode and a transmission line in a reflective load in a series structure in the phase shifter according to an embodiment of the present application.
21 is a diagram illustrating an example of an actual production photograph of a 360° phase shifter, which is a phase shifter according to another embodiment of the present application.
22 is a diagram illustrating an example of a circuit diagram of a 360° phase shifter that is a phase shifter according to another embodiment of the present application.
23 is a diagram illustrating an example of a control algorithm performed by a 360° phase shifter that is a phase shifter according to another embodiment of the present application.
24 is a diagram illustrating another example of a control algorithm performed by a 360° phase shifter that is a phase shifter according to another embodiment of the present application.
25 is an operation flowchart of a phase shift method according to an embodiment of the present application.

Hereinafter, embodiments of the present application will be described in detail with reference to the accompanying drawings so that those of ordinary skill in the art to which the present application pertains can easily implement them. However, the present application may be embodied in several different forms and is not limited to the embodiments described herein. And in order to clearly explain the present application in the drawings, parts irrelevant to the description are omitted, and similar reference numerals are attached to similar parts throughout the specification.

Throughout this specification, when a part is "connected" with another part, it is not only "directly connected" but also "electrically connected" or "indirectly connected" with another element interposed therebetween. "Including cases where

Throughout this specification, when it is said that a member is positioned "on", "on", "on", "under", "under", or "under" another member, this means that a member is positioned on the other member. It includes not only the case where they are in contact, but also the case where another member exists between two members.

Throughout this specification, when a part "includes" a component, it means that other components may be further included, rather than excluding other components, unless otherwise stated.

The present application relates to a phase shifter (particularly, a low-loss reflective analog phase shifter) having a reflection load in which a varactor diode and a transmission line are connected in series. Prior to the detailed description of the present application, a description of some terms mentioned herein is as follows.

Electronically controlled may refer to a component, device, or circuit that changes a state or function when an electrical control signal is applied or changed. This signal (electronic control signal) may include voltage, current, frequency, phase, and other electrical characteristics. An electronic control device or element may be either "analog" or "digital".

Digital may refer to a component, device, or circuit having multiple states separated by possible integers.

Analog may refer to a component, device, or circuit that can exist in any one possible state among a continuous range of possible states.

Reactance refers to an electrical characteristic. Reactance, along with resistance, helps define the circuit performance of a component. Reactance can be either capacitive or inductive. For example, capacitor components typically have large capacitive reactance and low resistance.

A varactor (or varactor diode) refers to a component having a variable reactance. Accordingly, the electronically controlled varactor means a component having a capacitance or inductance value that can be changed by application or change of an electronic control signal.

A phase means an electrical characteristic. Phase, along with amplitude, helps define the state of an electrical signal. Phase can be used to describe time in a time-varying periodic electrical signal.

A phase shift refers to a change in a phase state with respect to a periodic electrical signal.

A phase shifter may mean an electrical circuit including one or more phase shift elements, and may also incorporate other engineered electrical functions. Disclosed herein are various arrangements for a phase shifter.

Here, the phase shift element refers to a circuit element that causes a phase shift of an electrical signal. Also, a circuit element or element may refer to a small electrical circuit including one or more electronic components that perform an engineered electrical function.

1 is a block diagram showing a schematic configuration of a phase shifter 100 according to an embodiment of the present application. 2 is a view showing a schematic configuration of a phase shifter 100 according to an embodiment of the present application. 3 is a view showing an example of a layout photograph of the phase shifter 100 according to an embodiment of the present application actually manufactured according to an experimental example of the present application.

Hereinafter, the phase shifter 100 according to an embodiment of the present application will be referred to as the phase shifter 100 for convenience of description.

1 to 3 , the present phase shifter 100 is a phase shifter with reduced loss (particularly, loss due to resonance) compared to a conventional phase shifter (particularly, a reflective analog phase shifter), and a low loss It may be expressed differently as a reflective analog phase shifter or the like.

The phase shifter 100 may include a pair of reflective loads 10 and 20 and a hybrid coupler 30 . The reflection load may include a varactor diode and a transmission line. The reflective load may be formed by a series connection between the varactor diode and the transmission line.

The varactor diode may be expressed differently as a varactor, a varactor diode device, a variable capacitance diode, an electronically controlled varactor, and the like.

The phase shifter 100 (this phase shifter) according to an embodiment of the present application may be expressed differently as a phase shifter consisting of one stage. On the other hand, the phase shifter 200 according to another embodiment of the present application to be described later is a phase shifter in which two main phase shifters 100 are connected in series, unlike a phase shifter consisting of two stages. can be expressed

The pair of reflective loads 10 and 20 may include a first reflective load 10 and a second reflective load 20 . A pair of loads 10 and 20 may be spaced apart from each other.

The varactor diode and the transmission line included in the first reflective load 10 may be referred to as a first varactor diode 11 and a first transmission line 12 , respectively. Also, the varactor diode and the transmission line included in the second reflective load 20 may be referred to as the second varactor diode 21 and the second transmission line 22 , respectively.

The first reflective load 10 may be formed of a series connection between the first varactor diode 11 and the first transmission line 12 . The second reflective load 20 may be formed of a series connection between the second varactor diode 21 and the second transmission line 22 .

The hybrid coupler 30 may be arranged to connect the pair of reflective loads 10 and 20 and the input/output ports 1 and 2 . In particular, the hybrid coupler 30 is disposed between the pair of reflective loads 10 and 20 and the input/output ports 1 and 2, and between the pair of reflective loads 10 and 20 and the input/output ports 1 and 2 can be connected

The hybrid coupler 30 may be a four-port hybrid coupler having four ports. That is, the hybrid coupler 30 may include a first port 31 , a second port 32 , a third port 33 , and a fourth port 34 as four ports.

The hybrid coupler 30 includes a pair of first transmission lines 35 and 36 spaced apart in a first direction, and a pair of second transmission lines spaced apart in a second direction perpendicular to the first direction. It may include furnaces 37 and 38 . Here, based on the drawing of FIG. 2 , the first direction may mean a horizontal direction (9 o'clock-3 o'clock direction), and the second direction may mean a vertical direction (12 o'clock-6 o'clock direction).

The pair of first transmission lines 35 and 36 include one side (or input side) connected to the input port 1 , and the other side (or output side) connected to the output port 2 , the first transmission line 35 . It may include a furnace 36 . In addition, the pair of second transmission lines 37 and 38 include a second transmission line 37 and a pair of first transmission lines connecting one end of the pair of first transmission lines 35 and 36 between one end. It may include a second second transmission line 38 for connecting between the other end of the (35, 36).

A pair of reflective loads (10, 20) is a port (31, 32, 33, 34) of the hybrid coupler 30 except for the ports (31, 32) connected to the input/output ports (1, 2) ( 33 and 34) may be arranged to be connected to each other.

Illustratively, based on the drawing of FIG. 2 , the input/output ports 1 and 2 are connected to the ports 31 and 32 located on the left side of the hybrid coupler 30 , and the pair of reflective loads 10 and 20 are It may be arranged to be connected to the ports 33 and 34 located on the right side of the hybrid coupler 30 . In other words, the pair of reflective loads 10 and 20 may be disposed (positioned) on the shoulder of the hybrid coupler 30 .

Among the ports (31, 32, 33, 34) of the hybrid coupler 30, an input port (1, Input port) is connected to a first port 31, and an output port (2. Output) is connected to the second port 32 port) can be connected. In addition, the first reflective load 10 is connected to the third port 33 of the ports 31 , 32 , 33 , and 34 of the hybrid coupler 30 , and the second reflective load 10 is connected to the fourth port 34 . 20) can be connected.

In other words, the input port 1 , the first transmission line 35 and the first reflective load 10 on one side of the hybrid coupler 30 may be arranged in series connection with each other. In addition, the output port 2, the other side of the first transmission line 36 and the second reflective load 10 in the hybrid coupler 30 may be arranged in series connection with each other.

The hybrid coupler 30 can theoretically be implemented by arranging two types of transmission lines having a λ/4 length having a characteristic impedance of 50Ω and 35.35Ω. In other words, the hybrid coupler 30 may have a pair of first transmission lines 35 and 36 having a characteristic impedance of 35.35Ω and a length of λ/4. Also, the hybrid coupler 30 may have a pair of second transmission lines 37 and 38 having a specific impedance of 50Ω and a length of λ/4.

이고, 전기적 길이가 λ/4 일 수 있다. 또한, 하이브리드 커플러(30)의 전송선로 중 세로로 배치되는 제2 전송선로(37, 38)는 특성 임피던스가 Z₀ (=50Ω) 이고, 전기적 길이가 λ/4 일 수 있다.In other words, the first transmission lines 35 and 36 arranged horizontally among the transmission lines of the hybrid coupler 30 have a specific impedance.

, and the electrical length may be λ/4. In addition, among the transmission lines of the hybrid coupler 30 , the second transmission lines 37 and 38 disposed vertically may have a characteristic impedance of Z ₀ (=50Ω) and an electrical length of λ/4.

The input signal can flow to the reflective load after dividing the power at the contact point of the transmission line with characteristic impedance of 50Ω and 35.35Ω. Thereafter, the signal reflected from the reflective load may be synthesized at the output port.

In other words, the signal (input signal) input to the hybrid coupler 30 through the input port 1 is the second transmission line 37 and 38 having a characteristic impedance of 50 Ω and a first transmission line having a specific impedance of 35.35 Ω ( After dividing the power at the contact points of 35 and 36 , it can flow to a pair of reflective loads 10 and 20 . Thereafter, the signal (reflected signal) reflected from the pair of reflective loads 10 and 20 may be synthesized and outputted through the output port 2 . That is, the output signal output through the output port 2 may be a synthesized reflected signal generated by synthesizing each of the two reflected signals reflected from the pair of reflective loads 10 and 20 . At this time, in order to prevent loss due to signal synthesis (synthesis of two reflected signals), the two reflective loads must be identical.

Accordingly, in this phase shifter 100, in order to prevent loss due to signal synthesis, the same two reflective loads as a pair of reflective loads 10 and 20, a first reflective load 10 and a second reflective load ( 20 ) may be connected to the hybrid coupler 30 . That is, the first reflective load 10 and the second reflective load 20 have the same characteristics and may have the same structure.

Since the hybrid coupler 30 is a configuration well known to those of ordinary skill in the art to which the present application pertains, the following is a description of the hybrid coupler 30 itself (eg, structure, characteristics, operating principle, etc.) Rather than a description, the example in which the hybrid coupler 30 is applied to the present phase shifter 100 will be mainly described.

Each of the pair of reflective loads 10 and 20 in the phase shifter 100 may be formed of one varactor diode and one grounded transmission line on a circuit. That is, the first transmission line 12 included in the first reflective load 10 is expressed differently as the grounded first transmission line 12 , and the second transmission line 22 included in the second reflective load 20 . ) may be expressed differently as the grounded second transmission line 22 .

The reflective load considered in the phase shifter 100 may be expressed as Equation 1 below. That is, each of the first reflective load 10 and the second reflective load 20 may be expressed as in Equation 1 below.

[Equation 1]

,

은 전송선로(12, 22)의 길이를 나타낸다. 이때, 전송선로의 길이는 전기적 길이(Electric Length)를 의미할 수 있다. 전기적 길이라 함은 높은 주파수일수록 짧은 파장이 됨으로, 그 파장에 근거한 길이를 의미할 수 있다. 즉, 전기적 길이는 실제 물리적 길이 L을 전파의 파장 λ으로 나눈 길이인 L/λ를 의미할 수 있다.Here, x is the reactance of the reflective loads (10, 20),

,

denotes the length of the transmission lines 12 and 22 . In this case, the length of the transmission line may mean an electric length. The electric length may mean a length based on the wavelength because the higher the frequency, the shorter the wavelength. That is, the electrical length may mean L/λ, which is the length obtained by dividing the actual physical length L by the wavelength λ of the radio wave.

는 2π × 공급주파수[GHz], C 는 버랙터 다이오드(11, 21)의 커패시턴스[pF]를 나타낸다. 여기서, 공급주파수는 입력포트(1)를 통해 제공되는 입력신호에 대응하는 주파수를 의미할 수 있다. Also, in Equation 1,

is 2π × supply frequency [GHz], C represents the capacitance [pF] of the varactor diodes (11, 21). Here, the supply frequency may mean a frequency corresponding to an input signal provided through the input port 1 .

)는 반사계수 정의를 이용하여 하기 수학식 2와 같이 표현될 수 있다.Reflection coefficient of reflective load (10, 20) (

) can be expressed as in Equation 2 below using the reflection coefficient definition.

[Equation 2]

)는 하기 수학식 4와 같이 달리 표현될 수 있다. 또한, 상기 수학식 2에 도시된 반사계수의 위상(반사계수 위상,

)는 하기 수학식 5와 같이 표현될 수 있다.Here, Z _RL may be expressed as in Equation 3 below. In addition, the reflection coefficient (

) can be expressed differently as in Equation 4 below. In addition, the phase of the reflection coefficient shown in Equation 2 (reflection coefficient phase,

) can be expressed as in Equation 5 below.

[Equation 3]

where, Z _RL RL in is the abbreviation for Reflection Load, and Z _RL is the impedance of the reflective load.

[Equation 4]

[Equation 5]

The reflective loads 10 and 20 of the phase shifter 100 have no theoretical loss, and the phase change according to the change in x , which is the reactance of the reflective loads 10 and 20, may be as shown in FIG. 4 .

4 is a diagram illustrating an example of a phase change according to a change in reactance of a reflective load in the phase shifter 100 according to an embodiment of the present application. In other words, FIG. 4 is a view showing a change in the reflection coefficient phase according to a change in the reactance x of the reflective load as a tangent function. Here, the tangent function may mean a function expressed as in FIG. 5 described above.

Referring to FIG. 4 , the phase shifter 100 may output a changed phase as shown in FIG. 4 as the reactance of the reflective loads 10 and 20 changes.

In this phase shifter 100, the capacitance of the varactor diodes 11 and 21 is changed as the direct current (DC) voltage applied to the varactor diodes 11 and 21 is changed (changed, controlled) A phase angle may be shifted with respect to the output signal of the shifter 100 .

That is, the DC voltage provided from the DC voltage port 3 may be applied to the varactor diodes 11 and 21 . As the DC voltage applied to the varactor diodes 11 and 21 is changed (changed), the capacitance C of the varactor diodes 11 and 21 may be changed. The phase of the phase shifter 100 may be shifted (changed) by a change in capacitance of the varactor diodes 11 and 21 .

In other words, as the DC voltage applied to the varactor diodes 11 and 21 is controlled, the capacitance values of the varactor diodes 11 and 21 are changed so that the transition angle can be adjusted. The varactor diodes 11 and 21 in the phase shifter 100 may be electronically controlled through a DC voltage.

The phase of the signal (input signal) input to the phase shifter 100 is adjusted according to the DC voltage applied to the varactor diodes 11 and 21, and the phase-adjusted signal is an output signal as an output port (2) can be output toward

Meanwhile, in the phase shifter 100, the length of the transmission lines 12 and 22 included in the pair of reflective loads 10 and 20 (that is, the length of the first transmission line and the length of the second transmission line, respectively) may be set to a length such that the phase change of the reflection coefficient of the reflective loads 10 and 20 shifts a phase angle of 180° or more. The length of the transmission lines 12 and 22 may be preferably set to λ/8, where λ may be a preset value according to a frequency (supply frequency). A more detailed description of the length setting of the transmission lines 12 and 22 is as follows.

Specifically, in the phase shifter 100, the transmission lines 12 and 22 included in the pair of reflective loads 10 and 20 may have different characteristics of the reflective loads according to their lengths. That is, each of the first transmission line 12 connected in series with the first varactor diode 11 and the second transmission line 22 connected in series with the second varactor diode 21, according to the length The characteristics of the reflective load may vary.

As an example, when the supply frequency is 5.8 GHz, the phase change calculated by combining [Equation 1] and [Equation 2] may be as shown in Table 1 below. In other words, Table 1 below shows the transmission lines 12 and 22 obtained according to an experiment of the present application for setting (determining) the lengths of the optimal transmission lines 12 and 22 applied to the phase shifter 100. A table comparing the phase change according to the length of In this case, the Length in Table 1 is the length of the transmission line and may mean an Electric Length.

[Table 1]

In an experiment of the present application for setting (determining) the length of the optimal transmission lines 12 and 22, it is assumed, for example, that the varactor diodes 11 and 21 operate in the range of 0.1 to 1.2 pF. In addition, the characteristics of the device used in an experiment of the present application may be as shown in Table 2 below.

[Table 2]

At this time, based on the drawing of FIG. 2 , in Table 2, L _A is the vertical length of the hybrid coupler 30 , in particular, the length of a pair of second transmission lines 37 and 38 included in the hybrid coupler 30 . can mean In addition, L _B may indicate the length of the first transmission lines of the pair are included in, in particular, hybrid coupler 30 as the width of the hybrid couplers 30, 35 and 36.

In addition, L ₁ may mean the length of a transmission line at the front end of the reflective loads 10 and 20, and L ₂ may mean a transmission line at the rear end of the reflective loads 10 and 20. Here, the front end transmission line may refer to a transmission line connected to one side (hybrid coupler side) with respect to the varactor diodes 11 and 21 based on the drawing of FIG. 2 . In addition, the rear-end transmission line may refer to a transmission line connected to the other side (ground side) with respect to the varactor diodes 11 and 21 based on the drawing of FIG. 2 . The rear-end transmission line may refer to the above-described first transmission line 12 and the second transmission line 22 . Also, L ₁ and L ₂ may mean actual physical lengths.

Accordingly, the first reflective load 10 may include a front end transmission line corresponding to the length L ₁ , the first varactor diode 11 , and the first transmission line 12 as a rear end transmission line. Also, the second reflective load 20 may include a front end transmission line corresponding to the length L ₁ , a second varactor diode 21 , and a second transmission line 22 as a rear end transmission line.

According to an experiment of the present application, the transition angle tends to be wider as the length of the transmission lines 12 and 22 increases at 5.8 GHz, but the phase changes rapidly according to the change in the capacitance of the varactor diodes 11 and 21, It was confirmed that a value that cannot be expressed occurs depending on the applied direct current (DC) voltage.

In other words, according to an experiment of the present application, the greater the degree of change in reactance within the limited capacitance range of the varactor diode, the greater the transition angle, but accordingly, the phase operates nonlinearly with respect to the capacitance, making it difficult to control.

In order to solve this problem, the present application secures a phase of 180° or more, and a transmission line of an appropriate length (a transmission line having a length of λ/8) in which the phase is linearly shifted is connected in series with a varactor diode (or capacitor). A pair of reflective loads 10 and 20 may be configured.

Accordingly, in the present application, the length of the transmission lines 12 and 22 included in the present phase shifter 100 is set to λ/8, which shows the best linearity among the length values of the transmission lines for shifting the phase by 180° or more. You can (select and apply). That is, in the present phase shifter 100, the length of the transmission lines 12 and 22 may be preferably set to λ/8 as an optimal value. The transmission lines 12 and 22 may have transmission line characteristics that reduce resonance loss.

In the present application, the varactor diodes 11 and 21 are difficult to directly connect to the hybrid coupler 30 in the process of manufacturing the actual substrate in relation to the phase shifter 100. For example, having a length of λ/24. It may be arranged to be indirectly connected with a transmission line interposed therebetween.

As the transmission lines 12 and 22 in the phase shifter 100 are set to a length of λ/8, the phase shifter 100 may shift a phase angle of 180° or more.

According to this, the phase shifter 100 provides a reflective load in a series connection between the varactor diode and the transmission line, thereby causing problems with the conventional reflective analog phase shifter (that is, current is concentrated due to the resistance component, and the resonance loss is reduced. Accordingly, the problem that transmit power is impaired) can be solved. That is, the phase shifter 100 provides the first reflective load 10 through a series connection between the first varactor diode 11 and the first transmission line 12 , and generates the second reflective load 20 . By providing a series connection between the two varactor diodes 21 and the second transmission line 22, it is possible to reduce (reduce) the resonance loss due to the resistance. This is because, when the varactor diode and the transmission line are connected in series, the magnitude of the resistance component among the impedance of the reflective load becomes smaller than the magnitude of the reactance, thereby reducing the effect on the reflection loss.

In other words, the phase shifter 100 arranges the varactor diode and the transmission line in the reflective loads 10 and 20 in a series connection, so that the size of the resistance component of the impedance of the reflective load becomes smaller than the size of the reactance. It is possible to reduce the effect on the reflection loss, thereby reducing the resonance loss caused by the resistor to achieve a small power loss (ie, the power loss can be effectively reduced).

5 is a diagram schematically showing a circuit diagram of a phase shifter 200 according to another embodiment of the present application. 6 is a view showing an example of a layout photograph of the phase shifter 200 according to another embodiment of the present application actually manufactured according to an experimental example of the present application. For example, in FIG. 5 , DC blocks 1 and 2 (DC block_1 and DC block_2) may be set to 35 pF, but is not limited thereto.

Hereinafter, the phase shifter 200 according to another embodiment of the present application will be referred to as the 360° phase shifter 200 for convenience of description.

5 and 6 , the 360° phase shifter 200 may include two phase shifters 100 . Here, the two phase shifters 100 may mean that the above-described phase shifters 100 are two. Therefore, even if the description of the 360° phase shifter 200 is omitted below, the contents described with respect to the above-described phase shifter 100 are included in the 360° phase shifter 200 . The same may be applied to the description of each of the two phase shifters 100 .

The present 360° phase shifter 200 may be prepared by connecting two main phase shifters 100 in series in two stages. In other words, as the phase shifter 100 is connected in series in two stages, the present 360° phase shifter 200 capable of a 360° phase shift may be provided. The main phase shifter 200 is a phase shifter in which two main phase shifters 100 are connected in series, and may be expressed differently as a two-stage phase shifter.

The 360° phase shifter 200 may extend the range of the shift phase angle compared to the single bone phase shifter 100 by connecting the phase shifters 100 in series in two stages. That is, the present 360° phase shifter 200 may increase the range of shiftable phase angles compared to one main phase shifter 100 .

The 360° phase shifter 200 may operate to reduce insertion loss through electronically controlled. That is, the present 360° phase shifter 200 can reduce the insertion loss by electronically controlling the two main phase shifters 100 connected in two stages.

The 360° phase shifter 200 may reduce loss by electronically controlling the varactor diodes included in each of the two main phase shifters 100 through a DC voltage.

The two main phase shifters 100 included in the 360° phase shifter 200 may be individually electronically controlled, or may be electronically controlled in an integrated manner. In this case, electronically controlling the phase shifter 100 may mean, for example, controlling the DC voltage applied to the phase shifter 100 to be changed, but is not limited thereto.

For example, the electronic control of any one of the two main phase shifters 100 is referred to as a first electronic control DC Control_1, and the other one of the main phase shifters 100 is referred to as a first electronic control (DC Control_1). ) may be referred to as a second electronic control DC Control_2. The 360° phase shifter 200 transmits the signal of the first electronic control (DC Control_1) and the signal of the second electronic control (DC Control_2) to two DC voltages connected to each of the two main phase shifters 100 , respectively. It may be provided from each of the ports (3).

In addition, a processor unit (not shown) may be connected to the two DC voltage ports 3 for electronic control of the 360° phase shifter 200 , and electronic control may be performed by the processor unit. Here, the processor unit (not shown) may mean a circuit element or device capable of generating and providing various control signals for electronic control, but is not limited thereto. In addition, a memory unit (not shown) for storing various signals (input signals, output signals, control signals, etc.) related to the operation of the 360° phase shifter 200 may be connected to the processor unit (not shown). have.

In the 360° phase shifter 200 , the phase shift angle may be variously adjusted by control (control) of the applied DC voltage.

Accordingly, the present application may provide the present phase shifter 100 and the present 360° phase shifter 200 . Hereinafter, for convenience of description, the present phase shifter 100 and the present 360° phase shifter 200 proposed herein will be referred to as the proposed phase shifters 100 and 200 .

The phase shifter 100 includes a hybrid coupler 30 and a pair of reflective loads 10, having a 4-port structure for connecting the input/output ports 1 and 2 and the pair of reflective loads 10 and 20. 20) may be included. Here, the reflective loads 10 and 20 may be formed of a series connection between a varactor diode and a transmission line. This phase shifter 100 may be otherwise referred to as a low-loss reflective analog phase shifter.

A pair of reflective loads having the same reflective loads 10 and 20 may be connected to the two ports 33 and 34 of the hybrid coupler 30 in the phase shifter 100 . The phase shifter 100 may have a transition angle of at least 180° or more by changing the impedance viewed from the input side of the reflective loads 10 and 20 according to the voltage.

The phase shifter 100 is configured such that the phase change of the reflection coefficient viewed from the input side of the reflective loads 10 and 20 shifts a phase angle of 180° or more in consideration of the characteristics of the varactor diodes 11 and 21 connected in series. It may have transmission lines 12 and 22 whose length is determined.

By connecting the phase shifter 100 in series in two stages, the present 360° phase shifter 200 capable of shifting a phase of 360° can be provided. That is, the 360° phase shifter 200 is a hybrid coupler 30 and the phase shifter 100 having a pair of reflective loads 10 and 20 connected in series in two stages, so that the phase of 360° is It is possible to provide a phase shifter 200 for shifting .

The 360° phase shifter 200 may reduce loss by electronically controlling the varactor diodes included in each of the two main phase shifters 100 through a DC voltage.

The present disclosure may provide a phase shifter capable of 360° shifting through the present 360° phase shifter 200 . This 360° phase shifter 200 may be otherwise referred to as a low-loss analog phase shifter capable of 360° shifting. The 360° phase shifter 200 can reduce the resonance loss and shift the phase of 360°.

The present disclosure may provide the phase shifters 100 and 200 with reduced insertion loss and reduced complexity through simplified fabrication. The proposed phase shifters 100 and 200 may adjust the phase of an input signal (input signal) according to a DC voltage applied to the varactor diode, and provide the phase-adjusted signal as an output signal. The present application can shift the phase of 360° through the 360° phase shifter 200, and can lower power loss through electronic control. Through this, the proposed phase shifters 100 and 200 can increase the reception power of the terminal during downlink and supply power required for a high data rate.

The present disclosure may provide the phase shifters 100 and 200 capable of increasing power efficiency between transceivers during downlink in mobile communication. The present application can provide the proposed phase shifters 100 and 200 as a reflection type low-loss phase shifter operating at 5 to 6 GHz.

The present disclosure can provide the phase shifters 100 and 200 that can solve the problems of the conventional phase shifters (that is, the problem of increased power loss accompanying increasing the shift amount of the analog phase shifters). In particular, the present application reduces the loss due to resonance of the analog phase shifter by configuring the reflective load as a series connection between the varactor diode and the transmission line, and at the same time, it is possible to form a transition angle of at least 180° or a transition angle of 360°. It is possible to provide the phase shifters (100, 200).

The present invention can provide a phase shifter capable of shifting in a 360° range while reducing an insertion loss between input and output signals through the proposed phase shifters 100 and 200 . The proposed phase shifters 100 and 200 can achieve low power loss.

In the present application, the reflected load is simplified through the proposed phase shifters 100 and 200, and through this, the power burden is reduced to configure the base station to increase the reception power of the terminal during downlink of mobile communication. can be raised

In addition, the present application can efficiently supply power required for a rapidly increasing data rate through the proposed phase shifters 100 and 200 . The proposed phase shifters 100 and 200 can output higher power through an amplifier, thereby effectively reducing the time required for wireless charging.

In one experiment of the present application, which will be described later, in order to measure the performance of the proposed phase shifters 100 and 200, the proposed phase shifters 100 and 200 as an example were fabricated on an RF35 substrate of 0.3 mm as an example.

7 is a view showing a change in power loss according to a DC voltage applied to a varactor diode in the phase shifter 100 according to an embodiment of the present application for each frequency as a result according to an experimental example of the present application.

Referring to FIG. 7 , with respect to the present phase shifter 100 having a 1-stage structure, a change in loss according to a change in a DC voltage applied to the varactor diode can be confirmed. According to this, since the hybrid coupler 30 is designed to operate at, for example, 5.8 GHz, it can be confirmed that the loss is relatively large at other frequencies.

8 is a view showing a change in the phase of the reflection coefficient according to the DC voltage applied to the varactor diode in the phase shifter 200 according to another embodiment of the present application for each frequency as a result according to an experimental example of the present application.

In FIG. 8 , ΔDC Voltage1 and ΔDC_Voltage2 mean a change in the magnitude of a DC voltage applied to each of the two main phase shifters 100 included in the 360° phase shifter 200 .

Referring to FIG. 8 , this 360° phase shifter 200 is a 360° phase shifter 200 in which two phase shifters 100 capable of shifting a phase of 180° or more are serially connected in a 2-stage structure at 5.6 to 6.0 GHz. ° It can be confirmed that the input signal can be changed. That is, it can be confirmed that a phase shift of 360° is possible with respect to the input signal through the 360° phase shifter 200 . In addition, it can be seen that the 360° phase shifter 200 can perform a phase shift of 340 degrees at 5.0 GHz, 340 degrees at 5.2 GHz, and 354 degrees at 5.4 GHz.

9 is a result according to an experimental example of the present application, showing an example of the result of improved insertion loss of the phase shifter (200, that is, a phase shifter composed of two stages) according to another embodiment of the present application through electronic control. the drawing shown.

Referring to FIG. 9 , in the case of the 360° phase shifter 200, it can be confirmed that the insertion loss is improved after the electronic control is performed (After Process) compared to before the electronic control is performed (Before Process). have.

Meanwhile, in the above description, the reflective loads 10 and 20 in which the varactor diode and the transmission line are connected in series may be referred to as tunable transmission lines 10 and 20 differently. That is, the first reflective load 10 may be referred to as a first tunable transmission line 10 , and the second reflective load 20 may be referred to as a second tunable transmission line 20 .

The present disclosure may provide the low-loss phase shifters 100 and 200 using the tunable transmission lines 10 and 20 . In general, a tunable transmission line may refer to a transmission line whose impedance is adjusted by a change in the thickness or length of the transmission line. Herein, the impedance may be adjusted according to a change in the capacitance of the varactor diode. Accordingly, the tunable transmission lines 10 and 20 considered herein may mean a series reflective load between a varactor connected to a phase shifter and a transmission line.

With the introduction of 5G mobile communication and MIMO technology, the importance of beamforming and beam steering technology is increasing, and the design of a low-loss phase shifter determines its performance. In the conventional phase shifter using resonance, there is a problem in that the radiation characteristic of the antenna is deteriorated due to an increase in power loss due to resonance. In other words, a phase shifter having a wide shift angle and low insertion loss is essential for implementing MIMO and beamforming technologies. The conventional phase shifter has a problem in that it has a high power loss due to resonance of a reflective load.

Accordingly, the present application proposes analog phase shifters 100 and 200 to which the tunable transmission line 10 and 20 technique is applied to improve insertion loss. According to the present disclosure, loss compared to a conventional phase shifter can be improved and a design area can be reduced by providing the phase shifters 100 and 200 using the tunable transmission lines 10 and 20 . In addition, the present disclosure may provide a phase shifter 200 capable of changing a phase of 360° by connecting two phase shifters 100 in series in a two-stage manner.

The technique proposed in the present application is based on a change in the reflection coefficient according to the reactance of the reflective load. The phase shifter 100 (this phase shifter) proposed herein may include a hybrid coupler 30 and two reflective loads 10 and 20 as shown in FIGS. 2 and 10 .

10 shows an example of a circuit diagram of a reflective load included in the phase shifter 100 according to an embodiment of the present disclosure.

Referring to FIG. 10 , each of the first reflective load 10 (a first tunable transmission line) and the second reflective load 20 (the second tunable transmission line) may be formed of a circuit diagram as shown in FIG. 10 . In this case, the first varactor diode 11 included in the first reflective load 10 may be, for example, the first capacitor 11 , and the second varactor diode 21 may be, for example, the second capacitor 21 . can be

In the present application, after the reflective loads 10 and 20 are connected in series with the varactor diodes 11 and 21 and the transmission lines 12 and 22, the transmission lines 12 and 22 are powered according to resonance within the capacitance change range. The impact of losses can be effectively reduced.

The capacitors 11 and 21 and the grounded transmission lines 12 and 22 can be replaced with the grounded transmission lines 12 and 22 having the same reactance as in Equation 6 below, and by changing the capacitance, an equivalent transmission line may vary in length. When connected to a reflective load, it may be shifted by twice the phase of the equivalent transmission line according to Equation 7 below.

In other words, the capacitance of the reflective loads 10 and 20 and the transmission lines 12 and 22 can be replaced with the grounded equivalent transmission lines through Equation 6 below. In addition, it can be seen that the phase of the input signal is shifted by twice the length of the equivalent transmission line through Equation 7 below.

[Equation 6]

[Equation 7]

In the case of the conventional phase shifter, in order to express a phase of 360°, resonance in which reactance is infinitely changed has to occur, which causes current to be concentrated in the resistance component, thereby increasing the loss of the load.

Accordingly, the present application configures the reflective load so that resonance does not occur (that is, configures the reflective load so that the varactor diode and the transmission line form a series connection with each other) to lower the loss, and in addition, 360 through the 2-stage series connection It is possible to provide the phase shifters 100 and 200 for adjusting the entire phase of °.

11 is a view showing an insertion loss according to a change in a DC voltage applied to a varactor diode of the phase shifter 100 according to an embodiment of the present application as a result according to an experimental example of the present application. 11, 'Conventional' means a case of a conventional phase shifter (conventional resonant phase shifter), and 'Simulation of Proposed' means a simulation result of the present phase shifter 100 proposed herein, , 'Experiment of Proposed' is a diagram showing an example of an experimental result of the present phase shifter 100 proposed in the present application.

12 is a view showing a phase change according to a change in a DC voltage applied to a varactor diode of the phase shifter 100 according to an embodiment of the present application as a result according to an experimental example of the present application.

13 is a diagram illustrating performance comparison between the phase shifter 100 according to an embodiment of the present application and a conventional phase shifter as a result according to an experimental example of the present application. In FIG. 13 , 'Conventional' indicates a result by the conventional phase shifter, and 'Proposed' indicates a result by the present phase shifter 100 proposed herein. In addition, 'IL' represents the size of an insertion loss between the input and output of FIG. 2 . At this time, among the values described in relation to 'Proposed', the values in parentheses mean the simulation result values.

Hereinafter, in describing the experimental results of the present application, the conventional phase shifter (Conventional), which is a performance comparison target with the present phase shifter 100, is exemplarily referred to as the above-mentioned conventional document 1 [Chien-San Lin, Sheng-Fuh Chang and Wen-Chun Hsiao, “A Full-360Reflection-Type Phase Shifter With Constant Insertion Loss”, IEEE Trans. Microwave Theory Tech, vol.18, No.2, pp.106-108, February, 2008] may mean a phase shifter manufactured.

11 to 13 , the phase shifter 100 (this phase shifter) proposed herein implements a tunable transmission line (reflected load) 10, 20 in which a varactor diode and a transmission line are connected in series. , it is possible to improve the insertion loss by about 22.5% p compared to the conventional phase shifter. In addition, according to the phase shifter 100 proposed herein, the area required for designing the phase shifter 100 is reduced by about 5 minutes compared to the conventional phase shifter 100 by simplifying the reflective loads 10 and 20 . can be reduced to 1. The phase shifter 100 proposed herein can contribute to reducing the complexity and power burden of a sub-6 GHz RF system of 5G, which can be confirmed through experimental results.

In addition, the capacitance value C1 of the first varactor diode 11 and the capacitance value C2 of the second varactor diode 21 included in the phase shifter 100 may have, for example, the same value. . For example, the C1 value and the C2 value may be set to satisfy the condition of '0.1 ≤ C1,2 ≤ 1.2 [pF]'.

14 is a result according to an experimental example of the present application, showing a change in the input/output size according to frequency (a) and a change in input/output phase according to frequency (b) in the phase shifter 100 according to an embodiment of the present application It is a drawing.

14, in the case of the phase shifter 100 as seen in simulation, when the capacitance is 0.1pF, it has a bandwidth of 2.2 GHz within -3 dB, 2.0 GHz within -2 dB, and 1.2 GHz within -1 dB. It can be seen that the characteristics of operation are shown.

15 is a diagram illustrating performance comparison between a phase shifter 200 (this 360° phase shifter) according to another embodiment of the present application and a conventional phase shifter as a result according to an experimental example of the present application. In FIG. 15 , 'Conventional' indicates a result obtained by the conventional phase shifter, and 'Proposed' indicates an experimental result by the present 360° phase shifter 200 . At this time, among the values described in relation to 'Proposed', the values in parentheses mean the simulation result values.

Referring to FIG. 15 , it can be confirmed that the power efficiency (η) of the 360° phase shifter 200 is improved by 26.79% p compared to the conventional phase shifter as a result of simulation. In addition, it can be confirmed that the 360° phase shifter 200 has improved power efficiency of up to 8.78% p as a result of the experiment.

16 is a view showing a comparison of performance (insertion loss) between the phase shifters 100 and 200 proposed in the present application and an ideal phase shifter as a result according to an experimental example of the present application.

In FIG. 16 , Experiment indicates a result according to an experiment of the present application, EM Simulation indicates an example of a simulation result according to an embodiment of the present application, and Ideal indicates a result of an ideal phase shifter.

Referring to FIG. 16 , the phase shifters 100 and 200 proposed herein apply the tunable transmission line 10 and 20 technique (that is, a technique for implementing a varactor diode and a transmission line in a serial connection form). It can be confirmed that by reducing the insertion loss through , it is possible to design close to the ideal phase shifter.

As such, the present application implements a 360° low-loss phase shifter (this 360° phase shifter, 200) used for beamforming by connecting the phase shifters including the tunable transmission lines 10 and 20 in series in 2-stage. can

In addition, the present application implements the phase shifter 100 by connecting the reflective loads 10 and 20 to the capacitors 11 and 21 and a transmission line of λ/8 in series, thereby exhibiting linearity with a transition angle of 180° or more. You can control the phase.

In addition, the present 360° phase shifter 200 can improve p- loss (insertion loss) by 8.78% compared to the conventional phase shifter. In addition, the 360° phase shifter 200 can reduce an area required for design to three-eighths compared to that of a conventional phase shifter design.

The phase shifters 100 and 200 proposed herein can be applied to a 5G Sub-6GHz base station, a radar, a wireless power transmission transmitter, etc. to effectively improve the complexity and power burden of the RF system.

17 is a view showing a Smith chart of the phase shifters 100 and 200 manufactured according to the technology proposed in the present application as a result according to an experimental example of the present application. In other words, FIG. 17 is a Smith chart showing a change in a reflection coefficient according to a change in a DC voltage for each frequency in the phase shifters 100 and 200 according to an embodiment of the present application as a result according to an experimental example of the present application. Smith chart).

In particular, (a) to (f) in FIG. 17 is a view showing the change of the reflection coefficient according to the change of the DC voltage at 5.8 GHz, 5.0 GHz, 5.2 GHz, 5.4 GHz, 5.6 GHz, and 6.0 GHz, respectively, as a Smith chart. to be.

Referring to FIG. 17 , a reflection coefficient of a reflective load included in the proposed phase shifters 100 and 200 may be changed according to a change in a DC voltage in a frequency band between 5 and 6 GHz. In addition, as the capacitance of the varactor diode included in the reflective load is changed according to the change in the DC voltage, the phase angles of the proposed phase shifters 100 and 200 may be shifted accordingly.

In other words, FIG. 17 is a diagram that visually expresses the phase change of the reflection coefficient, which is a core principle of manufacturing the phase shifter, and the magnitude of the loss according to the phase change through the Smith chart. Since the phase shifter used in one experiment of the present application was premised to be used in the ISM band centered at 5.8 GHz, a Smith chart is shown in a circular shape at 5.8 GHz, showing a relatively constant power loss, while at other frequencies, it has an elliptical shape. It can be seen that the Smith chart is shown. In addition, the Smith chart means that the magnitude of the power loss increases toward the center of the circle, and the larger the radius of the concentric circle, the smaller the magnitude of the power loss.

Hereinafter, in relation to the operation method of the phase shifters 100 and 200 proposed in the present application, a phase change method, a loss minimization algorithm application technology, and a linearization algorithm application technology will be described in detail.

First, the description of the phase change method may be more easily understood with reference to FIG. 18 .

18 is a diagram schematically illustrating a configuration of a phase shifter 100 (this phase shifter) according to an embodiment of the present application. That is, FIG. 18 is an example in which the configuration of FIG. 2 is expressed differently.

Referring to FIG. 18 , the phase shifter 100 may be configured by connecting one hybrid coupler 30 and two reflection loads 10 and 20 . The reflective loads 10 and 20 may be configured by a series connection between the varactor diodes 11 and 21 and the transmission lines 12 and 22 . Each of the transmission lines 12 and 22 is a grounded transmission line and may operate as an inductor according to Equation 8 below.

[Equation 8]

이고, λ=c/f 일 수 있다. λ는 파장의 길이, c는 광속, f는 주파수를 의미할 수 있다.

은 전송선로(12, 22)의 물리적 길이를 의미하고,

는 전송선로(12, 22)의 전기적 길이(deg)를 의미할 수 있다.

는 2πf이고,

은 인덕턴스를 의미할 수 있다.Here, Z _L is the impedance of the transmission lines 12 and 22 , and Z ₀ is a characteristic impedance and may be, for example, 50 Ω. Also,

and λ=c/f. λ may indicate the length of a wavelength, c may indicate a speed of light, and f may indicate a frequency.

means the physical length of the transmission line (12, 22),

may mean the electrical length (deg) of the transmission lines (12, 22).

is 2πf,

may mean inductance.

The phase shifter 100 may change the capacitance of the varactor diodes 11 and 21 by applying a DC voltage, thereby changing the phase of the input signal and outputting it. The impedance of the reflective loads 10 and 20 may be expressed as in Equation 9 below. Also, the magnitude of the phase changed by the phase shifter 100 may be expressed as shown in FIG. 4 according to Equation 10 below.

[Equation 9]

[Equation 10]

는 반사계수의 위상을 나타낸다.Here, Z _in is the impedance of the reflective loads (10, 20), x is the reactance of the reflective loads (10, 20),

represents the phase of the reflection coefficient.

Strictly speaking, the inverse function of the tangent function is defined only in [-90°, 90°], but in FIG. 4 , other ranges are also expressed in order to intuitively grasp the phase change.

For example, when the reactance (x) of the phase shifters 100 and 200 proposed herein is -100 to 50, the phase may be changed by 220 deg. In addition, when the reactance of the phase shifters 100 and 200 proposed herein is -∞ to ∞, the phase may be changed by 360 deg.

The phase shifters 100 and 200 proposed herein adjust the capacitance C value of the varactor diodes 11 and 21 through the DC voltage provided from the DC voltage port 3, so that the reflective loads 10 and 20 You can change the reactance, which can cause the phase to shift.

In the phase shifters 100 and 200 , a matching terminal for impedance matching may be additionally connected between the hybrid coupler 30 and the reflective loads 10 and 20 . Such a matching terminal may be connected between the hybrid coupler 30 and the reflective loads 10 and 20 in an open stub method (ie, a method of connecting the transmission line in parallel without grounding), and at both matching terminals can be connected in the same way.

A more detailed description of the characteristic impedance of the transmission lines 35 , 36 , 37 , and 38 in the hybrid coupler 30 is as follows.

Since Traveling Voltage and Current Waves in a high-frequency circuit depend on the travel distance variable of an electric signal, they can be expressed as Equations 11 and 12 by the wave equation.

[Equation 11]

[Equation 12]

19 shows an example of an equivalent circuit for the definitions of voltage and current (a) and the incremental length of a transmission line.

The following Equation 13 may be established according to the characteristics of the transmission line at high frequency according to FIG. 19 .

[Equation 13]

In this case, if Equation 12 and Equation 13 are combined, it can be expressed as Equation 14 below.

[Equation 14]

Using Equation 14, the following Equation 15 may be experimentally derived in consideration of the fringing effect of the microstrip transmission line among the transmission lines. Here, the fringing effect refers to a phenomenon in which magnetism is not uniformly formed in space, unlike an ideal assumption.

[Equation 15]

는 유효 유전율(Effective dielectric constant)을 의미한다.

는 마이크로스트립 전송선로의 높이,

는 마이크로스트립 전송선로의 두께를 의미한다.

는 마이크로스트립 전송선로를 제작하는데 사용된 유전체의 유전율을 의미한다.here,

denotes an effective dielectric constant.

is the height of the microstrip transmission line,

is the thickness of the microstrip transmission line.

is the dielectric constant of the dielectric used to fabricate the microstrip transmission line.

이 필요하게 되었고, 이를

로 표현하기도 한다.In general, the characteristic impedance is Z ₀ =50 Ω, but due to the characteristics of the hybrid coupler 30, the characteristic impedance according to Equation 15 is

was needed, and

It is also expressed as

Meanwhile, in the present application, the reason for connecting the reflective loads 10 and 20 in a series structure between the varactor diode and the transmission line will be described as follows.

20 is a diagram for explaining the reason for connecting a varactor diode and a transmission line in a reflective load in a series structure in the phase shifter according to an embodiment of the present application.

In FIG. 20, R1 and R2 mean parasitic resistances of the varactor diode, which may have a magnitude of 0 in an ideal case. An inductor of nH, a capacitor of pH, and a resistance of 1 to 4 Ω can be used, for example, to operate at 5.8 Hz.

로 표현되는 전력 손실과 관련될 수 있다. 병렬 구조의 경우는 하기 수학식 16과 같이 표현되고, 직렬 구조의 경우는 하기 수학식 17과 같이 표현될 수 있다.In general, it is irrelevant, but when LC resonance occurs, the size of the resistance component and the impedance real part of the parallel structure and the series structure will be different, which

It can be related to the power loss expressed as . The parallel structure may be expressed as Equation 16 below, and the serial structure may be expressed as Equation 17 below.

[Equation 16]

[Equation 17]

Accordingly, in the present phase shifter 100, a low loss can be achieved by using a series structure according to the equation of power loss proportional to the real part of the impedance. According to the structure of the reflective load proposed in the present application (ie, if the reflective load is configured through the technique proposed herein), the present application can reduce the magnitude of power loss occurring in the resistor due to resonance. This can be proved using Equations 16 and 17 described above.

Hereinafter, the loss minimization algorithm technology applied to the phase shifters 100 and 200 proposed in the present application will be described as follows.

21 is a view showing an example of an actual production photograph of a 360° phase shifter 200, which is a phase shifter according to another embodiment of the present application, and FIG. 22 is a 360° phase shifter according to another embodiment of the present application. It is a diagram showing an example of a circuit diagram of the shifter 200 .

21 and 22 , the 360° phase shifter 200 may include a first phase shifter 101 and a second phase shifter 102 as two phase shifters 101 and 102 . can Here, each of the first phase shifter 101 and the second phase shifter 102 may be a phase shifter having the same structure and function as the aforementioned phase shifter 100 . Accordingly, even if omitted below, the descriptions of the phase shifter 100 may be equally applied to the descriptions of the first phase shifter 101 and the second phase shifter 102 , respectively. .

The present 360° phase shifter 200 can change the phase of 360° by connecting two of the present phase shifters 100 in series. That is, the 360° phase shifter 200 may be provided by connecting the first phase shifter 101 and the second phase shifter 102 in series.

The 360 ° phase shifter 200 may be provided by connecting the input (Input ₂₎ of the output (Output ₁₎ and the second phase shifter 102 of the first phase shifter is 101.

In this case, the electronic control of the first phase shifter 101 is referred to as a first electronic control (DC Control_1), and the electronic control of the second phase shifter 102 is referred to as a second electronic control (DC Control_2). can be That is, the 360° phase shifter 200 applies a DC voltage to the first DC voltage port 3 ′ for applying the DC voltage to the first phase shifter 101 and the second phase shifter 102 . A second DC voltage port 3 ″ may be provided.

At this time, the description of the above-described DC voltage port 3 is the same in the description of each of the first DC voltage port 3' and the second DC voltage port 3'', even if the contents are omitted below. can be applied.

The 360° phase shifter 200 may change the phase by individually controlling the two DC voltage ports 3' and 3'' to which the DC voltage is applied. The power loss of the present 360° phase shifter 200 in which the two phase shifters 101 and 102 are connected in two stages can be mitigated by applying a control algorithm that minimizes the resonance loss. In other words, the 360° phase shifter 200 may reduce power loss by applying a control algorithm that minimizes resonance loss. That is, the 360° phase shifter 200 may perform a control algorithm.

Here, the control algorithm adjusts the magnitude of the DC voltage applied to the varactor diode (varactor diode element) included in each of the two phase shifters 101 and 102, and finds a combination that minimizes the measured resonance loss. can be designed to

For the implementation of the above-described control algorithm, the 360° phase shifter 200 is a discretized analog to digital converter (ADC) (not shown) and a memory unit 210 as shown in FIG. 22 . , a processor unit 202 , and a digital-to-analog converter (DAC, not shown).

The ADC can discretize the loss measured at the input port and the output port. The memory unit 210 may store a digital signal or a discrete digital signal. The processor unit 220 may compare the magnitude (size) with the power loss of the previous trial. The processor unit 220 may be referred to as a control unit differently. The DAC may output the updated digital code as DC Control_1 through the first DC voltage port 3' and DC Control_2 through the second DC voltage port 3'' as a DC voltage. Based on these components, the present 360° phase shifter 200 may perform a control algorithm as shown in FIG. 22 .

23 is a diagram illustrating an example of a control algorithm performed by the 360° phase shifter 200 which is a phase shifter according to another embodiment of the present application. In particular, the control algorithm shown in FIG. 23 represents an example of a loss minimization algorithm.

In FIG. 23 , V ₁ is a control signal applied through the first DC voltage port 3 ′ for controlling the first phase shifter 101 , and may mean a first electronic control (DC Control_1) signal. . V ₂ is a control signal applied through the second DC voltage port 3 ″ for controlling the second phase shifter 102 , and may mean a second electronic control (DC Control_2) signal. i ₁ may mean a current applied to the first phase shifter 101 , and i ₂ may mean a current applied to the second phase shifter 102 .

Referring to Figure 23, the 360 ° phase shifter 200 phase value defined based obtain the phase measured by the loss and V ₁ and V ₂ as measured by V ₁ and V _2, and the user of the acquired values, and It can be compared to the minimum loss. The parameter value may be changed based on the comparison result value, and this 360° phase shifter 200 may control and change the phase.

In the 360° phase shifter 200 , a combination of DC voltages (capacitance) having the same phase shift amount may be formed based on a preset value. Referring to FIG. 9 , it can be seen that the power loss is reduced (reduced) by the 360° phase shifter 200 operating in the same manner as the above-described control algorithm. That is, in the case of the present 360° phase shifter 200, by electronically controlling the two phase shifters 101 and 102 individually, before electronic control is performed (Before Process) compared to after electronic control is performed (After Process) ), it can be seen that the insertion loss is improved.

Hereinafter, the linearization algorithm technology applied to the phase shifters 100 and 200 proposed in the present application will be described as follows.

The linearization algorithm may be implemented in the same manner as the aforementioned loss minimization algorithm, as shown in FIG. 21 .

In order to output a phase difference desired by a user through the 360° phase shifter 200 , the above-described control algorithm may be applied to the 360° phase shifter 200 . Specifically, the 360° phase shifter 200 may perform a control algorithm to derive the closest phase to the following Equation (18).

[Equation 18]

For linearization control, the 360° phase shifter 200 measures the phase difference between the input port and the output port and outputs the phase closest to the phase derived through Equation 18, similar to the loss minimization control described above, The value of the first electronic control signal DC Control_1 and the value of the second electronic control signal DC Control_2 may be sequentially increased. Accordingly, the 360° phase shifter 200 is controlled to operate in the same manner as the above-described control algorithm, thereby reducing the time required for control and the amount of storage in controlling the phase desired by the user.

24 is a diagram illustrating another example of a control algorithm performed by the 360° phase shifter 200 which is a phase shifter according to another embodiment of the present application. In particular, the control algorithm shown in Fig. 24 represents an example of a linearization algorithm.

Referring to FIG. 24 , in the processor unit 220 of the 360° phase shifter 200, the value of the first electronic control (DC Control_1) signal and the value of the second electronic control (DC Control_2) signal are respectively V ₁ It may be designed to calculate a difference from a theoretically derived value by measuring the phase difference at _{V 2} and V 2 .

The processor unit 220 of the 360° phase shifter 200 may store these result values in the memory unit 210 . Thereafter, the processor unit 220 may compare the newly derived value by calling the value stored in the memory unit 210 through a command during the next execution, and store a smaller value in the memory unit 210 , at this time V ₁ and V ₂ may also be stored in the memory unit 210 together. _{The processor unit 220 may store V 1} and V ₂ after the final execution in the memory unit 210, and the 360° phase shifter 200 viewed from this may improve the linearity of the phase difference according to the DC voltage. can

Referring again to FIG. 22 , the present 360° phase shifter 200 is an optimal solution having the result of reducing the average return loss of the reflective load (reflective load), the ripple of the power loss, and the nonlinearity of the phase change with respect to the power change. Resonance loss can be reduced by selecting a control algorithm of That is, the 360° phase shifter 200 selects a combination of magnitude values of the DC voltage applied to the varactor diodes provided in each of the first phase shifter 101 and the second phase shifter 102 (DC By controlling the magnitude values of the voltage), an optimal algorithm can be performed.

To this end, the amplitude and phase of the S-parameters of the input port and the output port may be input to the processor unit 220 . Also, the processor 220 may output a combination of voltage values of DC_Control 1 and DC_Control_2, and operate the control algorithm based on the combination of the voltage values of DC_Control_1 and DC_Control_2.

In the step of manufacturing the 360° phase shifter 200, an optimal control algorithm (or control program) applied to the 360° phase shifter 200 may be determined, which is ) may be called (loaded) into the memory unit 210 of the .

In addition, the control algorithm (or control program) applied to the 360° phase shifter 200 is, for example, a phase shift (0 degrees to 360 degrees) and two DC voltages applied by two control channels of the phase shifter. The operation may be performed based on a table in which a relationship between the values (ie, a signal value of DC Control_1 and a signal value of DC Control_2) is defined in advance.

According to an embodiment of the present application, the processor unit 220 may receive a request for setting the output of the 360° phase shifter 200 as a given value for the output phase. The processor unit 220 may identify (confirm) two control voltages corresponding to the phase closest to the required output phase by using the above table stored in the memory unit 210 . Thereafter, the processor unit 220 transmits appropriate digital codes to a Digital-Analog Converter (DAC) connected to the two DC voltage ports 3' and 3'' provided in the 360° phase shifter 200. By doing so, it is possible to set the two control voltages to the previously identified (verified) values.

As such, the processor unit 220 individually electronically controls the two phase shifters 101 and 102 to effectively improve the insertion loss after the individual electronic control compared to before the electronic control is performed.

The memory unit 210 may store information on a relationship between a plurality of independent voltages and a phase change amount of an RF signal. The processor unit 220 checks voltage values corresponding to the amount of phase change based on the information stored in the memory unit 210 , and applies the checked voltage values to reflective loads included in each of the two phase shifters 101 and 102 . can be approved for

A value individually applied to the present 360° phase shifter 200 connected in two stages may be selected from a plurality of candidates for applicable voltage values. Here, the multiple candidates may include combinations of voltage values. Combinations of voltage values included in the plurality of candidates may be defined as including a combination of voltage values causing internal resonance of reflective loads.

The value individually applied to the 360° phase shifter 200 may be selected from, for example, a combination of voltage values representing the same phase change stored in the memory unit 210 . The processor unit 220 may compare combinations showing less insertion loss and output a result value. A DC voltage may be applied to the 360° phase shifter 200 by combining the voltage values derived by this process.

This application solves the problem that current is concentrated due to the resistance component in the conventional phase shifter and the transmission power is inhibited according to the resonance loss, thereby effectively reducing the resonance loss due to the resistor.

Hereinafter, an operation flow of the present application will be briefly reviewed based on the details described above.

25 is an operation flowchart of a phase shift method according to an embodiment of the present application.

The phase shifting method shown in FIG. 25 may be performed by the present phase shifter 100 or the present 360° phase shifter 200 described above. Therefore, even if omitted below, the descriptions of the present phase shifter 100 or the present 360° phase shifter 200 may be equally applied to the description of the phase shifting method.

Referring to FIG. 25 , in step S11, the hybrid coupler in the phase shifter may receive an input signal applied from an input port. Here, the phase shifter may be the present phase shifter 100 or the present 360° phase shifter 200 as described above.

Next, in step S12, in response to the applied signal received in step S11, the hybrid coupler may output an output signal whose phase is shifted relative to the input signal toward the output port connected to the hybrid coupler.

At this time, the output signal is a signal generated by synthesizing the reflected signals reflected from each of a pair of reflective loads in the phase shifter in response to the applied input signal, and is a serial connection between the varactor diode in the reflective load and the transmission line. It could be an affected signal. That is, the output signal may be a signal in which the insertion loss between the input and output signals is reduced by the serial connection between the varactor diode and the transmission line in the reflective load.

In the above description, steps S11 and S12 may be further divided into additional steps or combined into fewer steps, according to an embodiment of the present application. In addition, some steps may be omitted as necessary, and the order between steps may be changed.

The phase shift method according to an embodiment of the present application may be implemented in the form of program instructions that can be executed through various computer means and recorded in a computer-readable medium. The computer-readable medium may include program instructions, data files, data structures, etc. alone or in combination. The program instructions recorded on the medium may be specially designed and configured for the present invention, or may be known and available to those skilled in the art of computer software. Examples of the computer-readable recording medium include magnetic media such as hard disks, floppy disks and magnetic tapes, optical media such as CD-ROMs and DVDs, and magnetic such as floppy disks. - includes magneto-optical media, and hardware devices specially configured to store and carry out program instructions, such as ROM, RAM, flash memory, and the like. Examples of program instructions include not only machine language codes such as those generated by a compiler, but also high-level language codes that can be executed by a computer using an interpreter or the like. The hardware devices described above may be configured to operate as one or more software modules to perform the operations of the present invention, and vice versa.

In addition, the above-described phase shift method may be implemented in the form of a computer program or application executed by a computer stored in a recording medium.

The foregoing description of the present application is for illustration, and those of ordinary skill in the art to which the present application pertains will understand that it can be easily modified into other specific forms without changing the technical spirit or essential features of the present application. Therefore, it should be understood that the embodiments described above are illustrative in all respects and not restrictive. For example, each component described as a single type may be implemented in a dispersed form, and likewise components described as distributed may be implemented in a combined form.

The scope of the present application is indicated by the following claims rather than the above detailed description, and all changes or modifications derived from the meaning and scope of the claims and their equivalents should be construed as being included in the scope of the present application.

100, 200: phase shifter
10: first reflective load
11: first varactor diode
12: first transmission line
20: second reflective load
21: second varactor diode
22: second transmission line
30: hybrid coupler

Claims (12)

As a phase shifter,
A pair of reflective loads spaced apart from each other; and
A hybrid coupler disposed to connect the pair of reflective loads and input/output ports,
The reflective load consists only of a varactor diode connected to the hybrid coupler and a grounded transmission line that is a transmission line directly connected in series with the varactor diode,
The phase shifter,
The hybrid coupler in the phase shifter receives an input signal applied from an input port, and in response to the received applied signal, the hybrid coupler moves toward an output port connected to the hybrid coupler in phase with respect to the input signal output an output signal,
The output signal is
A signal generated by synthesizing the reflected signals reflected from each of the pair of reflective loads in the phase shifter in response to the applied input signal, but affecting the series connection between the varactor diode in the reflective load and the transmission line received signal,
The length of the transmission line is set to λ/8 as a length for linearly shifting the phase, and is set to a length such that the phase change of the reflection coefficient of the reflective load shifts a phase angle of 180° or more. cloth. According to claim 1,
The pair of reflective loads are arranged to be respectively connected to the remaining ports except for the ports connected to the input/output ports among the ports of the hybrid coupler,
As the DC voltage applied to the varactor diode is changed, the capacitance of the varactor diode is changed so that a phase angle shift with respect to the output signal is made, a phase shifter. delete According to claim 1,
The λ is a preset value according to the frequency, the phase shifter. A 360° phase shifter capable of 360° phase shift by connecting the phase shifter of claim 1 in series in two stages. 6. The method of claim 5,
wherein the 360° phase shifter operates to reduce insertion loss through electronic control. A phase shifting method by the phase shifter of claim 1, comprising:
(a) receiving, by the hybrid coupler in the phase shifter, an input signal applied from an input port; and
(b) in response to the received applied signal, the hybrid coupler outputting an output signal whose phase is shifted relative to the input signal toward an output port connected to the hybrid coupler,
The output signal is
A signal generated by synthesizing the reflected signals reflected from each of the pair of reflective loads in the phase shifter in response to the applied input signal, but affecting the series connection between the varactor diode in the reflective load and the transmission line received signal,
The phase shifter,
A pair of reflective loads spaced apart from each other; and
A hybrid coupler disposed to connect the pair of reflective loads and input/output ports,
The reflective load consists only of a varactor diode connected to the hybrid coupler and a grounded transmission line that is a transmission line directly connected in series with the varactor diode,
The length of the transmission line is set to λ/8 as a length for linearly shifting the phase, and is set to a length such that the phase change of the reflection coefficient of the reflective load shifts a phase angle of 180° or more. transition method. delete 8. The method of claim 7,
The pair of reflective loads are arranged to be respectively connected to the remaining ports except for the ports connected to the input/output ports among the ports of the hybrid coupler,
As the DC voltage applied to the varactor diode is changed, the capacitance of the varactor diode is changed so that the phase angle shift with respect to the output signal is made. delete 8. The method of claim 7,
The λ is a value preset according to the frequency, the phase shift method. A computer-readable recording medium recording a program for executing the method of any one of claims 7, 9 and 11 on a computer.

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