KR102156325B1 - unipolar and synchronous isolation-type class D audio amplifier by use of center-tap transformer - Google Patents

Abstract

The present invention relates to an isolated Class D audio amplifier for use in a digital audio amplifier system in general. Especially, the present invention relates to the isolated Class D audio amplifier that high efficiency can be achieved by implementing DC voltage adjustment, electrical isolation, and sound source signal amplification at the same time with one isolated Class D audio amplifier without the need for a separate SMPS stage deviated from a conventional digital audio amplifier system consisting of an isolated SMPS stage and a non-isolated Class D amplifier stage, a size of the system and manufacturing cost can be reduced by eliminating a capacitor bank, which was essential in the conventional digital audio amplifier system, a center-tap transformer is adopted, and the isolated Class D audio amplifier is operated by an unipolar synchronous method. According to the present invention, the DC voltage adjustment, electrical isolation, and sound source signal amplification are implemented at the same time with one isolated Class D audio amplifier without the need for the separate SMPS stage, thereby achieving high efficiency. In addition, the present invention has an advantage of reducing the size of the system and manufacturing cost and improving life and reliability of the system by eliminating the capacitor bank. Also, since it is possible to significantly increase a switching frequency of a signal applied to a transformer element, a small transformer can be applied so that the amplifier system can be reduced in size and weight.

Description

Unipolar and synchronous isolation-type class D audio amplifier by use of center-tap transformer}

The present invention relates generally to an isolated Class D audio amplifier for use in a digital audio amplifier system.

In particular, the present invention deviates from the conventional digital audio amplifier system consisting of an isolated SMPS stage and a non-isolated Class D amplifier stage, and without the need for a separate SMPS stage, DC voltage regulation, electrical isolation, sound source signal amplification with one isolated Class D audio amplifier. Simultaneous implementation of the system enables high efficiency to be achieved, and by eliminating the capacitor bank, which was essential in the conventional digital audio amplifier system, reduces the system size and manufacturing cost, adopts a center tap transformer, and operates in a unipolar synchronous method. It relates to a type Class D audio amplifier.

Audio amplifiers that amplify sound source signals (small signals) in an audio system to drive speakers are largely classified into analog amplifiers and digital amplifiers. While the analog amplifier has excellent sound quality, it is not only low in efficiency due to high power semiconductor losses, but also has a disadvantage in that the size and weight of the amplifier system are very large because it requires a large heat dissipation mechanism to reduce heat generation of elements. On the other hand, digital amplifiers have significantly lower loss and heat generation of power semiconductors than analog amplifiers, and most of the recent large-capacity audio amplifiers use Class D amplifiers, which are digital amplifiers.

1 is a block diagram showing a general configuration of a digital audio amplifier system according to the prior art. Referring to [Fig. 1], a digital audio amplifier system generally includes a rectification stage 11 that receives AC commercial power (Vac) as an input, an isolated SMPS (Switch Mode Power Supply) stage 12, and a Class D amplifier stage ( 13). As such, the conventional digital audio amplifier system consists of a two-stage power stage structure in which the isolated SMPS stage (12) and the Class D amplifier stage (13) are cascaded. Due to this 2-stage power stage structure, the power of the amplifier system The efficiency is low and the number of parts increases.

This digital audio amplifier system is characterized by an isolation design, which is to prevent electric shock. If the insulation design is not provided, when an electric shock occurs due to the user touching the speaker with a wet hand, the current flows directly from the commercial power supply (Vac) to the human body, which increases the risk of personal accident. To prevent such an accident from occurring, an insulation transformer 12a is installed at the SMPS terminal 12 between the AC input commercial power Vac and the speaker to electrically separate the commercial power Vac and the speaker. Even in the event of an electric shock, the current from the commercial power supply (Vac) does not flow directly to the human body, thus significantly reducing the risk of personal accidents.

However, in this structure, the amount of current supplied to the Class D amplifier stage 13 is limited by the current capacity of the insulated SMPS stage 12. With this amount of current, there is a problem that the woofer that amplifies the instantaneous large amplitude low-frequency voice signal cannot be properly driven, so the capacitor bank 14 between the isolated SMPS stage 12 and the Class D amplifier stage 13 Required. The capacitor bank 14 normally receives current from the insulated SMPS stage 12 to store energy, but the Class D amplifier stage 13 uses a large current that exceeds the capacity of the insulated SMPS stage 12 When this occurs, the energy stored in the capacitor bank 14 is supplied. The capacitor bank 14 is generally composed of a large number of large electrolytic capacitors and is quite large in size. As the load (speaker capacity and number) increases, the capacity and number of capacitors increase proportionally. The high-capacity digital audio system has a problem in that the system size is increased due to the capacitor bank 14, and there is a problem that the system life and reliability are weakened due to the electrolytic capacitor.

A general circuit configuration for implementing the digital audio amplifier system according to the prior art is shown in [Fig. 2]. [Fig. 2] shows a circuit diagram of the conventional most common digital audio amplifier system. The rectification stage 11, the isolated SMPS stage 12, and the Class D amplifier stage 13, which input AC commercial power (Vac). Consists of The isolated SMPS stage 12 is equipped with an insulation transformer 12a, a capacitor bank 14, and a converter control circuit 12b, and the class D amplifier stage 13 has a switching circuit 13a and a low-pass filter 13b. ) And an amplifier control circuit 13c.

In such a digital audio amplifier system, a general circuit configuration of the rectifying stage 11 is shown in FIG. 3. The rectifying stage 11 serves as a kind of DC voltage source in a digital audio amplifier system. First, when power factor improvement is not required, diodes are used as shown in (a) and (b) of [Fig. 3]. After rectifying the AC commercial voltage through the device, a large smoothing capacitor is used to output a DC voltage source with a small fluctuation rate. [Fig. 3] (a) is called a full-bridge rectifier, which is used when the commercial power supply (Vac) is at a high voltage of about 220 V, and (b) of [Fig. 3] is called a half-bridge rectifier or back-pressure circuit rectifier. Vac) is used when the voltage is as low as 110 V. In addition, when power factor improvement and harmonic regulation avoidance are required, a rectifier of a method using a separate power factor improvement converter as shown in (c) of FIG. 3 is adopted.

In such a digital audio amplifier system, the general circuit configuration of the Class D amplifier stage 13 is shown in [Fig. 4]. Class D amplifier stage 13 synthesizes the analog input signal (S1) to be amplified with a high frequency carrier signal (S2) and modulates it by a PWM (Pulse Width Modulation) method, and then converts the switching transistor with this synthesized PWM signal (S3). A method of amplifying the signal by operating and passing the amplified PWM signal through an LC low pass filter, resulting in amplification of the small signal analog input signal (S1) into a large-signal analog signal (S4) having a large amplitude Adopted.

Referring to [Fig. 4], the amplifier control circuit 13c is generally an input signal (audio signal) (S1) in an audible frequency band (eg, 20 Hz to 20 kHz) and a high frequency (eg 200 kHz to 550 kHz). A phosphorus carrier signal S2 is provided. These input signals S1 and carrier signals S2 are small signals (for example, 3.3 V). At this time, the carrier signal is also called a sampling signal, and a triangle wave, sawtooth wave, or the like may be used. The amplifier control circuit 13c synthesizes the input signal S1 and the carrier signal S2 and performs PWM modulation to generate a high-frequency (for example, 200 kHz to 550 kHz) synthesized PWM signal S3. In this process, the signal amplitude value of the input signal S1 is reflected in the duty ratio value of the synthesized PWM signal S3. That is, when the input signal S1 increases, the duty ratio of the synthesized PWM signal S3 increases, and when the input signal S1 decreases, the duty ratio of the synthesized PWM signal S3 decreases.

The switching circuit 13a3 switches and controls the upper switching transistor 13a1 and the lower switching transistor 13a2 in opposite directions according to the synthesized PWM signal S3. In general, MOSFET power transistors are used as these switching transistors 13a1 and 13a2. By controlling the switching of these transistors 13a1 and 13a2, a PWM signal amplified with a high voltage (±HV) (eg, ±70 V) is obtained, and the amplified PWM signal is passed through the LC low-pass filters 13b1 and 13b2. The output signal S4 is obtained. This output signal S4 is transmitted to the speaker to drive the speaker unit. At this time, the upper and lower switching transistors 13a1 and 13a2 constituting the Class D amplifier stage 13 only come and go in the cut-off region and the saturation region and do not operate in the linear region. The loss is definitely smaller than that of an amplifier.

[Fig. 5] and [Fig. 6] are diagrams showing another implementation method of a digital audio amplifier system according to the prior art.

First, [Fig. 5] is a diagram showing an implementation method to which the line transformer 22 is applied. Referring to [Fig. 5], a line transformer 22 is installed in the commercial power supply (Vac) for electrical insulation between the commercial power supply (Vac) and the amplifier output. In this case, by using the turn ratio of the transformer 22 as well as electrical insulation, the AC commercial voltage (Vac) is boosted or lowered to immediately obtain a predetermined DC voltage required by the Class D amplifier stage 23. It is possible, and therefore, there is no need for a separate SMPS, so the overall circuit is very simple and easy to implement. On the other hand, a transformer element operating at a frequency of 60 Hz of a commercial power supply (Vac) is mainly applied to small capacity because its size is large and its weight is heavy.

In addition, [Fig. 6] is a diagram showing an implementation method in which the impedance matching transformer 32 is applied. Referring to FIG. 6, an impedance matching transformer 32 was installed between the Class D amplifier stage 33 and the speaker for electrical insulation between the commercial power supply (Vac) and the amplifier output. In this case, the equivalent impedance of the speaker can be converted by using the turn ratio of the transformer 32 as well as electrical insulation. This method also has the advantage that the circuit is very simple and easy to implement as a separate SMPS is not required. However, this method is also mainly applied to small capacity because the frequency of the sound source signal is still very low, about 20Hz to 20kHz, which is an audible frequency band, and the impedance matching transformer 32 is large in size and weight compared to capacity.

In the above, the implementation form of the digital audio amplifier system according to the prior art has been described. As shown in [Fig. 2], the method in which the insulation transformer 12a is built into the SMPS has the advantage that it can be used for general purposes ranging from small capacity to large capacity, and is suitable in size and weight, whereas the insulated SMPS stage 12 and the non-insulated type The class D amplifier stage (13) is a cascade-connected two-stage power stage structure, so the efficiency is low, and the converter control circuit (12b) and the class D amplifier stage (12b) to control the output voltage of the isolated SMPS stage (12) The amplifier control circuit 13c is required to control the output audio signal of 13), and has a disadvantage in that the overall manufacturing cost is quite high.

In addition, in order to prevent distortion such as clipping of the output signal due to the current capacity limit of the isolated SMPS stage 12 when amplifying an instantaneous large amplitude low-frequency audio signal, a large number of large electrolytic capacitors are used. There is also a disadvantage that a capacitor bank 14 is required. As such, system efficiency is low, the number of parts is large, and due to the large number of electrolytic capacitors, not only is it weak in terms of system life and reliability, but also has a high manufacturing cost and a problem that the size of the entire system is increased.

In addition, there existed an implementation method using a line transformer 22 or an implementation method using an impedance matching transformer 32, and these have the advantage that the overall circuit is very simple and easy to implement because a separate SMPS is not required. Since the operating frequency of the voltage and signal applied to the voltages 22 and 32 is very low, the transformers 22 and 32 have a problem in that they have a large size compared to their capacity and have a heavy weight, so they are used only in a small capacity application environment.

Accordingly, there is a need for a technique capable of solving the problems of the prior art for a digital audio amplifier system.

It is an object of the present invention to provide an isolated Class D audio amplifier for use in a digital audio amplifier system in general.

In particular, the object of the present invention is to deviate from the conventional digital audio amplifier system consisting of an isolated SMPS stage and a non-insulated Class D amplifier stage, and without the need for a separate SMPS stage, DC voltage adjustment, electrical isolation, sound source with one isolated type Class D audio amplifier. High efficiency can be achieved by implementing signal amplification at the same time, and it is possible to reduce the system size and manufacturing cost by eliminating the capacitor bank, which was essential in the conventional digital audio amplifier system, and adopts a center tap transformer and operates in a unipolar synchronous method. It provides an isolated Class D audio amplifier.

On the other hand, the problem to be solved of the present invention is not limited to these matters, and other problems may be understood from the description of the present specification.

In order to achieve the above object, the unipolar synchronous isolation type Class D audio amplifier of the center tap transformer according to the present invention includes any one of a full bridge circuit, a half bridge circuit, and a push-pull circuit, thereby providing a DC input power source (Vin ) To generate a high-frequency pulse signal (Vpg) by switching at a high speed (210, 310, 410); The secondary side electrically insulates the amplifier output from DC input power (Vin) through an internal center-tap insulating transformer element composed of a center-tapped winding, and DC input power by the turn ratio of the center-tap insulating transformer element from a high-frequency pulse signal (Vpg). High-frequency transforming units 220, 320, and 420 for generating a high-frequency transforming pulse signal Vpm obtained by adjusting the step-up or step-down of the pulse amplitude of the (Vin) level; A high-frequency transforming pulse signal by switching on and off according to a switch driving signal (Vqsw) that switches and controls a plurality of internal switching transistors (Q1 to Q4) constituting the current flow path for the high-frequency transforming pulse signal (Vpm) in a unipolar synchronous method. Pulse width modulators 230, 330, and 430 for generating a high-frequency PWM switching signal Vpwm in which (Vpm) is pulse width-modulated in a unipolar synchronous manner in response to the switch driving signal Vqsw; A low-pass filter unit 240, 340, and 440 configured to generate an audio output signal Vout by low-pass filtering the high-frequency PWM switching signal Vpwm; Receives feedback on the audio output signal Vout output from the low-pass filter units 240, 340, and 440, generates a high frequency carrier signal Vcar whose frequency and phase are synchronized with the high frequency pulse signal Vpg, and inputs audio In response to the error between the signal Vs and the audio output signal Vout, the internal switching transistors Q1 to Q4 of the pulse width modulators 230, 330, and 430 are turned on from the carrier signal Vcar according to the unipolar synchronization method. And an amplifier control unit 150 that generates a switch driving signal Vqsw for off-switching and provides it to the pulse width modulators 230, 330, and 430.

At this time, the center tap insulating transformer element of the high frequency transformers 220 and 320 is configured to form a path for subtracting electric energy from the parasitic capacitors of the internal switching transistors M1 to M4 of the high frequency pulse generators 210 and 310, The high-frequency pulse generators 210 and 310 switch the DC input power supply Vin at a fixed frequency and 50% fixed duty at high speed to provide zero voltage to the internal switching transistors M1 to M4 of the high-frequency pulse generators 210 and 310. It can be configured to perform switching.

In addition, the high frequency pulse generator 410 is configured in a push-pull topology, the pulse width modulator 430 is driven in a unipolar manner, and the frequency of the high frequency pulse generator 410 and the pulse width modulator 430 The phases can be configured in synchronization with each other.

Further, the amplifier control unit 150 includes a scaler 151 for matching a voltage scale between the audio input signal Vs and the audio output signal Vout; A subtractor 152 that obtains an error value between the audio input signal Vs and the audio output signal Vout; In order to generate a high frequency carrier signal (Vcar) whose frequency and phase are synchronized with the high frequency pulse signal (Vpg), and increase or decrease the audio output signal (Vout) so that the error value converges to zero, the carrier signal (Vcar) ) From the internal switching transistors Q1 to Q4 of the pulse width modulators 230, 330, and 430 according to the unipolar method to generate a switch driving signal Vqsw for switching on and off the pulse width modulators 230, 330, It may be configured to include a; switching control unit 153 provided to 430.

In addition, the high frequency transformers 220, 320, 420 have their own resonance frequency matched to the switching frequency fpg of the high frequency pulse generators 210, 310, and 410 to achieve a resonance condition, and the center tap 1 of the insulating transformer element LC resonance tanks (Lr, Cr) configured to be connected in series with the secondary side or the secondary side; may be provided.

According to the present invention, DC voltage regulation, electrical isolation, and sound source signal amplification are simultaneously implemented with a single isolated Class D audio amplifier without the need for a separate SMPS stage, thereby achieving high efficiency.

In addition, according to the present invention, there is an advantage in that a capacitor bank, which is essential in a conventional digital audio amplifier system, is unnecessary, thereby reducing a system size and manufacturing cost, and improving system life and reliability.

In addition, according to the present invention, it is possible to significantly increase the switching frequency of a signal applied to the transformer element, so that it is possible to apply a transformer of small parts, thereby reducing the size and weight of the amplifier system.

[Fig. 1] is a block diagram showing a general configuration of a digital audio amplifier system according to the prior art.
[Fig. 2] is a diagram showing a general circuit configuration of a digital audio amplifier system according to the prior art.
[Fig. 3] is a diagram showing a general circuit configuration of a rectifying stage for a digital audio amplifier system.
[Fig. 4] is a diagram showing a general circuit configuration of a Class D amplifier stage for a digital audio amplifier system.
[Fig. 5] is a diagram showing another implementation method of a digital audio amplifier system according to the prior art.
[Fig. 6] is a diagram showing another implementation method of a digital audio amplifier system according to the prior art.
[Fig. 7] is a block diagram showing the basic concept of an isolated Class D audio amplifier according to the present invention.
[Fig. 8] is a diagram showing the circuit topology of a high frequency pulse generator in the present invention.
[Fig. 9] is a diagram showing the circuit topology of a high frequency transformer in the present invention.
[Fig. 10] is a diagram showing a simplified combined circuit topology of a pulse generator and a high-frequency transformer in the present invention.
[Fig. 11] is a diagram showing the circuit topology of a pulse width modulator and a low-pass filter unit in the present invention.
12 is a diagram showing a unipolar operation waveform of a pulse width modulator in the present invention.
13 is a diagram showing a bipolar operation waveform of a pulse width modulator in the present invention.
[Fig. 14] is a diagram schematically showing a driving method of a pulse width modulator in the present invention.
[Fig. 15] is a diagram showing main switching operation waveforms according to a driving method of a pulse width modulator in the present invention.
[Fig. 16] is a view showing a first embodiment of a unipolar synchronous insulation type Class D audio amplifier of a center tap transformer according to the present invention.
[Fig. 17] is a diagram showing a unipolar operation waveform when the audio output signal Vout> 0 in the first embodiment of the present invention.
[Fig. 18] is a diagram showing a conduction path for each time section of a unipolar operation when the audio output signal Vout> 0 in the first embodiment of the present invention.
[Fig. 19] is a diagram showing a unipolar operation waveform when the audio output signal Vout <0 in the first embodiment of the present invention.
[Fig. 20] is a diagram showing a conduction path for each time section of a unipolar operation when the audio output signal Vout <0 in the first embodiment of the present invention.
[Fig. 21] is a view showing a second embodiment of a unipolar synchronous insulation type Class D audio amplifier of a center tap transformer according to the present invention.
[Fig. 22] is a diagram showing a unipolar operation waveform when the audio output signal Vout> 0 in the second embodiment of the present invention.
[Fig. 23] is a diagram showing a unipolar operation waveform when the audio output signal Vout <0 in the second embodiment of the present invention.
[Fig. 24] is a diagram showing a third embodiment of a unipolar synchronous insulation type Class D audio amplifier of a center tap transformer according to the present invention.
[Fig. 25] is a diagram showing a unipolar operation waveform when the audio output signal Vout> 0 in the third embodiment of the present invention.
[Fig. 26] is a diagram showing a unipolar operation waveform when the audio output signal Vout <0 in the third embodiment of the present invention.

Hereinafter, the present invention will be described in detail with reference to the drawings.

The conventional digital audio amplifier system has a two-stage power stage structure in which the isolated SMPS stage 12 and the non-isolated Class D amplifier stage 13 are cascaded, so the power efficiency of the amplifier system is low and the number of parts is large. . The present invention is characterized in that it is possible to achieve high efficiency by simultaneously implementing DC voltage adjustment, electrical isolation, and sound source signal amplification even with the isolated Class D audio amplifier 100 alone. In particular, since the capacitor bank 14, which was essential in the conventional digital audio amplifier system, is not required, the system size and manufacturing cost can be reduced.

[Fig. 7] is a block diagram showing the basic concept of the isolated Class D audio amplifier 100 according to the present invention. Referring to [Fig. 7], the isolated Class D audio amplifier 100 according to the present invention includes a high frequency pulse generator 110, a high frequency transformer 120, a pulse width modulator 130, a low pass filter unit ( 140), it is configured to include the amplifier control unit 150. At this time, the rectification stage 11 may be implemented with the rectification circuit illustrated in [Fig. 3], and generates a direct current (DC) operating power (Vin) from an AC commercial power source (Vac) to generate an insulated Class D according to the present invention. This module is supplied to the audio amplifier 100.

Hereinafter, each component of the isolated Class D audio amplifier 100 according to the present invention will be described.

The high-frequency pulse generator 110 switches the DC input voltage Vin supplied from the rectifying terminal 11 at high speed, preferably at a fixed frequency and 50% fixed duty, so that a high-frequency square wave pulse signal, that is, a high-frequency pulse signal Vpg. Create

In the present invention, since the high frequency pulse generator 110 switches at a fixed frequency and a fixed duty, a separate control circuit is not required, and as switching at a fixed frequency and 50% fixed duty, the soft switching of the power semiconductor device is performed regardless of the load. As a result, switching losses can be significantly reduced, resulting in high efficiency and high-speed switching at very high frequencies. In this case, it is possible to significantly reduce the size of the components of the insulating transformer element embedded in the high-frequency transformer unit 120 to obtain the advantage of miniaturization and weight reduction of the amplifier system.

The high frequency transformer 120 electrically insulates the amplifier output terminal from the DC input power (Vin) through an internal insulation transformer element, and uses the turn ratio of the insulation transformer element from the high frequency pulse signal (Vpg) to the DC input power (Vin) level. A high-frequency transforming pulse signal (Vpm) obtained by adjusting the step-up or step-down according to the desired pulse amplitude of is generated. As a high-frequency pulse signal (Vpg) of hundreds of kHz to several MHz is applied to the primary side of the insulating transformer element, the size of the parts of the insulating transformer element can be significantly reduced, thereby miniaturizing and reducing the weight of the amplifier system.

The pulse width modulator 130 is provided by the amplifier control unit 150 with a plurality of internal switching transistors Q1 to Q4 constituting a current flow path for the high frequency transforming pulse signal Vpm output from the high frequency transforming unit 120 By switching on and off according to the switch driving signal Vqsw, a high-frequency PWM switching signal Vpwm is generated in which the high-frequency transforming pulse signal Vpm is pulse-width modulated in response to the switch driving signal Vqsw.

The low-pass filter unit 140 generates an audio output signal Vout by low-pass filtering the high-frequency PWM switching signal Vpwm output from the pulse width modulator 130. In general, the speaker is driven by the audio output signal Vout.

The amplifier control unit 150 receives feedback of the audio output signal Vout output from the low-pass filter unit 140, and corresponds to an error between the audio input signal Vs and the audio output signal Vout, and a switch driving signal Vqsw And provides it to the pulse width modulator 130. When the audio input signal (Vs) is larger than the audio output signal (Vout), a switch driving signal (Vqsw) is generated in the direction of increasing Vout by increasing the PWM duty ratio in the pulse width modulator 130 to generate a switch driving signal (Vqsw) between Vs and Vout. Let the error converge to zero. Conversely, when the audio input signal Vs is smaller than the audio output signal Vout, a switch driving signal Vqsw is generated in the direction of reducing Vout by reducing the PWM duty ratio in the pulse width modulator 130, Make the error between Vout converge to zero.

By compensating for an error between the audio input signal Vs and the audio output signal Vout through this action, the isolated Class D audio amplifier 100 according to the present invention outputs a precise audio output signal.

On the other hand, the amplifier control unit 150 operates as a high frequency carrier signal Vcar, from which the internal switching transistors Q1 to Q4 of the pulse width modulator 130 are transferred according to a unipolar or bipolar method. A switch driving signal Vqsw for on-off switching is generated. In this specification, the switching frequency of the carrier signal Vcar is denoted as fcar. As shown in [Fig. 7], the pulse width modulator 130 is provided with a high-frequency transforming pulse signal (Vpm) of a switching frequency (fpg) and a switch driving signal (Vqsw) of a switching frequency (fcar), and using these Thus, pulse width modulation is achieved.

The amplifier control unit 150 may include a scaler 151, a subtracter 152, and a switching control unit 153. The scaler 151 is a component for matching a voltage scale between an audio input signal Vs, which is a small signal, and an audio output signal Vout, which is a large signal. In addition, the subtractor 152 is a component that obtains an error value between the audio input signal Vs and the audio output signal Vout in which the scale matching is performed. In addition, the switching control unit 153 increases or decreases the audio output signal Vout so that the error value between the audio input signal Vs and the audio output signal Vout converges to zero. ) From the internal switching transistors Q1 to Q4 of the pulse width modulator 130 according to a unipolar or bipolar method as shown in [Fig. 12] and [Fig. 13] to generate a switch driving signal (Vqsw) It is a component provided to the pulse width modulator 130.

Comparing [Fig. 2] and [Fig. 7], the conventional digital audio amplifier system requires individual control circuits 12b and 13c for the isolated SMPS stage 12 and the non-isolated Class D amplifier stage 13 On the other hand, since the present invention is composed of only one isolated Class D audio amplifier 100, only one control circuit 150 is required, thus simplifying the amplifier system and reducing the manufacturing cost.

In addition, conventionally, a capacitor bank 14 composed of a plurality of electrolytic capacitors was required to prevent clipping distortion due to the current capacity limit of the isolated SMPS stage 12, and these electrolytic capacitors would be vulnerable in terms of system life and reliability. In addition, there was a problem that the size and manufacturing cost of the amplifier system increased. On the other hand, in the present invention, since the DC voltage source (Vin) generated in the rectifying stage 11 is pulse-width modulated and applied to the output load (speaker) directly, the capacitor bank 14 is not required, and the conventional problem caused by the electrolytic capacitor This is resolved.

First, a detailed circuit topology of the high frequency pulse generator 110 and the high frequency transformer 120 for the isolated Class D audio amplifier 100 according to the present invention with reference to FIGS. 8 to 10 Be described clearly.

[Fig. 8] is a diagram showing the circuit topology of the high frequency pulse generator 110 in the present invention. In the present invention, the high-frequency pulse generator 110 is a component that generates a high-frequency square-wave pulse signal by switching the DC input voltage Vin supplied from the rectifying terminal 11 at high speed. Three circuit topologies of the high-frequency pulse generator (PG: Pulse Generator) 110 for the present invention are presented in [Fig. 8], and these are high-frequency pulse signals (Vpg) at the A and B terminals (eg, a square wave of 200 kHz). Signal). At this time, (a) and (b) of [Fig. 8] are a full-bridge method and a half-bridge method, respectively, and (c) of [Fig. 8] is a push-pull method (push- As a pull) method, it is a structure suitable for use with a transformer having a tap on the high-frequency transformer unit 120.

In the present invention, the high frequency pulse generator 110 is preferably configured to switch at a fixed frequency and 50% fixed duty. In the case of switching at a fixed frequency and a fixed duty, a separate control circuit is not required, so system complexity and manufacturing cost can be reduced. In addition, when switching at a fixed frequency and a fixed 50% duty, soft switching of the power semiconductor devices M1 to M4 can always be guaranteed while cooperatively operating with the transformer element of the high frequency transformer 120. When soft switching is performed, the switching loss of the power semiconductor devices (M1 to M4) is significantly lowered to achieve high efficiency as well as to reduce heat generation, thereby increasing the switching frequency of the power semiconductor devices (M1 to M4) to a very high level, for example. It becomes possible to go up to hundreds of kHz to several MHz levels. As the frequency of the high-frequency pulse signal Vpg increases, the size of the transformer element of the high-frequency transformer 120 can be reduced, so that the size of the amplifier can be made smaller.

When the high-frequency pulse generator 110 switches to a fixed frequency and 50% fixed-duty, soft switching of the power semiconductor devices (M1 to M4), in other words, zero-voltage switching is performed by the size of the load. Regardless of whether you can always guarantee. In general, during the switching process, the parasitic capacitors of the switching transistors (M1 to M4) are filled with electrical energy and emptied repeatedly.At this time, the electrical energy that has not been emptied is converted into thermal energy during the switching process. Heat generation occurs in (M1 ~ M4). In the present invention, the insulating transformer element of the high frequency transformer unit 120 is configured to provide a path for subtracting electrical energy from the parasitic capacitors of the switching transistors M1 to M4 of the high frequency pulse generator 110, the high frequency pulse generator 110 ) Operates at 50% fixed duty, so the condition for emptying the electric energy stored in the parasitic capacitor of the transistor is the same for all switching transistors (M1 ~ M4), and switching at a fixed frequency is affected by the load condition. Since this is eliminated, the zero voltage switching once achieved is still guaranteed regardless of the change in the load amount.

On the other hand, as described above, in soft switching, since heat generation in the switching process is very small, the switching frequency of the switching transistors M1 to M4 can be raised to a very high level, for example, to a level of several MHz. As also described above, as the frequency of the high-frequency pulse signal Vpg generated by the high-frequency pulse generator 110 increases, the size of the transformer element of the high-frequency transformer 120 can be further reduced, according to the present invention. You can get the advantage of making the size smaller.

[Fig. 9] is a diagram showing a circuit topology of the high frequency transformer 120 in the present invention. In the present invention, the high-frequency transformer 120 electrically insulates the output terminal of the amplifier from the input power source Vin and uses the turn ratio of the transformer element to determine the pulse amplitude of the high-frequency pulse signal Vpg from the original DC input power source Vin level. It is a component that generates a high-frequency transforming pulse signal (Vpm) by boosting or stepping down as desired. The high-frequency transformer 120 is provided with a transformer element for electrical insulation. In the present invention, the high-frequency pulse generator 110 generates and provides a pulse signal Vpg having a high frequency of about several tens to several hundred kHz, or several MHz. The transformer element of the high frequency transformer 120 may be of a very small size. The high-frequency transformer 120 adjusts the pulse amplitude using the turn ratio of the transformer element, which corresponds to a function conventionally performed by the isolated SMPS stage 12. Therefore, the isolated Class D audio amplifier 100 according to the present invention may not have a separate SMPS module.

Eight circuit topologies of the high-frequency transformer (TR) 120 for the present invention are presented in FIG. 9. Depending on the requirements of the application field, you can select and use an appropriate circuit topology among them. For example, it is advantageous to adopt a circuit topology using a tapped transformer element for low voltage and high current requirements, and a circuit topology using a tapless transformer element for high voltage and low current requirements. It is advantageous.

In addition, in (e), (f), (g), and (h) of [Fig. 9], an LC resonance tank composed of Lr and Cr elements is connected in series with the primary or secondary side of the insulating transformer element. The LC resonance tanks Lr and Cr are set to achieve a resonance condition with the switching frequency fpg of the high frequency pulse generator 110. At this time, when the resonance frequencies of the resonance tanks Lr and Cr match the switching frequency fpg of the high frequency pulse generator 110, that is, the same or very close, the resonance condition of the LC resonance tank is established. When the resonance condition is satisfied, when the high frequency pulse generator 110 applies a square wave voltage to the high frequency transformer 120, a sine wave current flows, which is advantageous in situations in which EMI (Electro Magnetic Interference) noise reduction is required. In the LC resonance tank, Lr can be implemented as a separate inductor element inserted from the outside, but depending on implementation examples, it can also be implemented as a leakage inductor of a transformer element.

In addition, the switching transistors M1 to M4 provided in the high frequency pulse generator 110 are turned on by adjusting the magnitude of the magnetizing inductor of the transformer element provided in the high frequency transformer 120 and the magnetizing inductor current flowing therein. It is also possible to ensure zero voltage switching at the time. Zero-voltage switching can significantly reduce the switching loss of the high-frequency pulse generator 110, so that not only high efficiency can be obtained, but also the frequency of the high-frequency pulse signal Vpg can be increased, so that the size of the transformer element of the high-frequency transformer 120 It is advantageous in that it can be made small.

[FIG. 10] is a diagram showing a simplified combined circuit topology of the high frequency pulse generator 110 and the high frequency transformer 120 in the present invention. [Fig. 8] shows three circuit topologies (PG-1 to PG-3) of the high-frequency pulse generator 100, and [Fig. 9] shows eight circuit topologies (TR-) of the high-frequency transformer unit 120. 1 to TR-8) are presented. Among them, when TR-5 or TR-7 is coupled to the half-bridge PG-2, the input capacitors C1 and C2 of the high frequency pulse generator 110 are the capacitors Cr of the LC resonance tanks Lr and Cr. Since the redundancy or the role becomes weak, these can be omitted and a simplified circuit topology can be configured as shown in (a) and (b) of [Fig. 10].

Next, referring to [Figs. 11] to [Fig. 15], the circuit topology of the pulse width modulator 130 and the low pass filter unit 140 for the isolated Class D audio amplifier 100 according to the present invention It will be described in detail.

11 is a diagram showing the circuit topology of the pulse width modulator 130 and the low-pass filter unit 140 in the present invention. In the present invention, the pulse width modulator 130 modulates the output signal of the high frequency transformer 120, that is, the high frequency transform pulse signal Vpm, in response to the switch driving signal Vqsw of the amplifier controller 150. ) Is a component. The output signal of the high frequency transformer 120 is a signal obtained by passing a high frequency pulse signal Vpg through a transformer element, and thus corresponds to a fixed frequency and a fixed duty of 50%. The pulse width modulator 130 PWM modulates a high-frequency transforming pulse signal Vpm having a fixed frequency and 50% fixed duty in response to the switch driving signal Vqsw. As described above, since the amplifier control unit 150 generates the switch driving signal Vqsw based on the error between the audio input signal Vs and the audio output signal Vout, the PWM modulation degree of the pulse width modulator 130 It is made in response to an error between the audio input signal Vs and the audio output signal Vout.

Five circuit topologies of the pulse width modulator (PM) 130 for the present invention are presented in [Fig. 11]. Referring to the circuit topology of FIG. 11, the internal switching transistors Q1 to Q4 of the pulse width modulator 130 constitute a current flow path for the high-frequency transforming pulse signal Vpm. Depending on the circuit topology, one or two diode elements may be included in the current flow path. When switching transistors (Q1 to Q4) are switched on and off according to the switch driving signal (Vqsw), the high-frequency transforming pulse signal (Vpm) may or may not be transmitted to the output terminal (Vpwm), or to the output terminal (Vpwm). Edo can be delivered as positive (+) or negative (-) by adjusting the delivery path. The switch driving signal Vqsw performs PWM modulation by adjusting the turn-on time period and the turn-off time period of the internal switching transistors Q1 to Q4 of the pulse width modulator 130, and the resulting output The signal is referred to as a high frequency PWM switching signal (Vpwm) in this specification.

At this time, the high-frequency PWM switching signal Vpwm, which is an output signal of the pulse width modulator 130, flows through the internal switching transistors Q1 to Q4 through the high-frequency transforming pulse signal Vpm transmitted from the high-frequency transformer 120. It is a large signal. In addition, the high-frequency PWM switching signal Vpwm is a (+) high level time period, a zero level time period, and a (-) high level time period a switch driving signal for on-off control of the switching transistors Q1 to Q4 ( Vqsw).

[Fig. 11] (a) and (b) are a form applicable to the case of the pulse width modulation method of the synchronous unipolar driving method, and (c), (d), (e) of [Fig. 11] are all pulses. The width modulation method, that is, the synchronous unipolar driving method, the asynchronous unipolar driving method, the synchronous bipolar driving method, and the asynchronous bipolar driving method are all applicable. The snubber capacitor (Csb) inserted in (c) and (d) of FIG. 11 is a high-frequency pulse signal Vpg of the high-frequency pulse generator 110 and a pulse width modulator ( When switching occurs when the switch driving signals (Vqsw) of 130) deviate from each other, the pulse width to perform the function of absorbing the ringing of current and voltage that occurs during a short period of the switching transient period. It was provided in the modulator 130. The snubber capacitor (Csb) is sufficient for a very small capacity suitable for this purpose, for example a very small capacity on the order of several ㎋ to several ㎌.

In addition, the low-pass filter unit 140 is a component that generates an audio output signal Vout by low-pass filtering the high-frequency PWM switching signal Vpwm output from the pulse width modulator 130. . The low-pass filter unit 140 may be implemented as a general single-stage LC filter shown in FIG. 11, and other types of low-pass filter circuits, such as a multi-stage LC filter or LCL filter, etc. It can be implemented as a filter circuit.

[Fig. 12] is a diagram showing a unipolar operation waveform of the pulse width modulator 130 in the present invention, and [Fig. 13] is a view showing a bipolar operation waveform of the pulse width modulator 130 in the present invention. In [Fig. 12] and [Fig. 13], the audio output signal (Vout) means a signal to be output by the audio amplifier. The switching transistors Q1 to Q4 of the pulse width modulator 130 are switched on and off as shown in Figs. 12 and 13 by the switch driving signal Vqsw provided by the amplifier control unit 150, and as a result The low pulse width modulator 130 outputs a high frequency PWM switching signal Vpwm as shown in FIGS. 12 and 13.

First, [Fig. 12] shows a unipolar operation waveform of the pulse width modulator 130. When Vout> 0, switch Q1 is always on and switch Q3 is always off, and switches Q2 and Q4 are complementary to each other by modulating the pulse width according to the error between the feedback audio output signal (Vout) and audio input signal (Vs). complementary). When Vout <0, switch Q2 is always off and switch Q4 is always on, and switches Q1 and Q3 are complementary by modulating the pulse width according to the error between the audio input signal (Vs) and the audio output signal (Vout) fed back. It works as

Next, Fig. 13 shows a bipolar operation waveform of the pulse width modulator 130. When Vout> 0, the driving signals of switches Q1 and Q2 have a duty of 50% or more, and the pulse width is modulated according to the error between the audio input signal (Vs) and the feedback audio output signal (Vout), and switches Q3 and Q4 are switches Q1. And operates complementarily with Q2. When Vout <0, the driving signals of switches Q1 and Q2 have a duty of less than 50%, and the pulse width is modulated according to the error between the audio input signal (Vs) and the audio output signal (Vout) fed back, and switches Q3 and Q4 are switches Q1. And Q2 and operate complementarily.

Meanwhile, in Figs. 12 and 13, an operation clock for switching on and off the internal switch transistors Q1 to Q4 and a switching clock of the high frequency PWM switching signal Vpwm accordingly correspond to the carrier signal Vcar.

[Fig. 14] is a diagram showing a summary of the driving method of the pulse width modulator 130 in the present invention. In the present invention, the synchronous type and the asynchronous type are determined depending on whether the high frequency pulse signal of the pulse generator 110 and the carrier signal Vcar of the amplifier control unit 150 are synchronized with each other in terms of frequency and phase. The concepts of unipolar and bipolar have already been shown in [Fig. 12] and [Fig. 13].

[Fig. 15] is a diagram showing a main switching operation waveform according to the driving method of the pulse width modulator 130 in the present invention.

In FIG. 15, Vpg represents a waveform of a high-frequency pulse signal output from the high-frequency pulse generator 110, and is operated at a fixed frequency (fpg) and a fixed duty (50%) as described above. In addition, Vpwm in FIG. 15 indicates that the high-frequency pulse signal Vpg of the high-frequency pulse generator 110 is boosted or stepped down by the transformer turn ratio through the high-frequency transformer unit 120, and the carrier through the pulse width modulator 130 A signal waveform after pulse width modulation is performed at the operating frequency fpwm of the signal Vcar is shown. Here, in the signal waveforms of Vpwm and Vpg, the waveforms of the hatched portion and the dotted line correspond to each other. That is, the hatched and dotted waveforms of the Vpwm waveform mean waveforms generated by pulse width modulation of the hatched and dotted waveforms of Vpg.

Referring to FIG. 15, the high frequency pulse signal Vpg and the carrier signal Vcar have a switching frequency (fpg, fcar) and a phase respectively set. In particular, as shown in (a), (b), (e), and (f) of [Fig. 15], in the synchronous driving method, the high frequency pulse signal (Vpg) and the carrier signal (Vcar) are the switching frequencies (fpg, fcar). ) And phase must be fully synchronized. Referring to FIG. 15, it can be seen that Vpwm performs switching according to the operating frequency fcar of the carrier signal Vcar.

Since (a) and (b) of Fig. 15 are synchronous, the snubber capacitor (Csb) is not required or can be minimized in the pulse width modulator 130, and (c) of [Fig. 15] is asynchronous and switching transient Since the snubber capacitor Csb must intervene in the section, a small-capacity snubber capacitor needs to be disposed.

In the case of the bipolar driving method as shown in (d), (e), and (f) of Fig. 15, one switching period of the pulse width modulator 130 is an energy transfer section in which electric energy is transmitted to the audio output terminal Cf. It is divided into an energy regeneration section where electric energy is extracted from the audio output terminal (Cf). When the output load current is positive (+), the time period in which Vpwm is positive (+) can be referred to as the energy transfer period, and the time period in which Vpwm is negative (-) can be referred to as the energy regeneration period. When the output load current is negative (-), it is the opposite. The energy transfer section and the energy regeneration section are considerably short since the driving frequency fcar of the pulse width modulator 130 is a high speed of about several hundred kHz to several MHz. In order to temporarily store the electric energy extracted from the audio output terminal (Cf) during such a short energy regeneration period, the pulse width modulator 130 requires a snubber capacitor (Csb), such as PM-3 and PM-4. The regenerative energy stored in the null capacitor Csb is then transferred back to the audio output terminal Cf in the energy transfer section and used as load energy.

In particular, the present invention proposes a unipolar synchronous isolation type Class D audio amplifier of a center tap transformer.

In the present invention, the high-frequency pulse generator 110 includes any one of a full-bridge circuit, a half-bridge circuit, and a push-pull circuit, thereby generating a high-frequency pulse signal Vpg by switching the DC input power supply Vin at high speed.

In addition, the high frequency transformer 120 has a center tap insulation transformer element in which the secondary side is composed of a center tap winding, and electrically insulates the amplifier output terminal from the DC input power source Vin through this center tap insulation transformer element. , From the high-frequency pulse signal Vpg, a high-frequency transforming pulse signal Vpm obtained by step-up or step-down adjustment of the pulse amplitude of the DC input power supply Vin level by the turn-ratio of the center tap insulating transformer element is generated.

In addition, the pulse width modulator 130 is a switch driving signal Vqsw for switching and controlling a plurality of internal switching transistors Q1 to Q4 constituting a current flow path for the high frequency transforming pulse signal Vpm in a unipolar synchronous manner. According to the on-off switching, a high-frequency PWM switching signal Vpwm is generated in which the high-frequency transforming pulse signal Vpm is pulse-width modulated in a unipolar synchronous manner in response to the switch driving signal Vqsw.

The low-pass filter unit 140 generates an audio output signal Vout by low-pass filtering the high-frequency PWM switching signal Vpwm.

The amplifier control unit 150 receives feedback of the audio output signal Vout output from the low-pass filter unit 140, and corresponds to an error between the audio input signal Vs and the audio output signal Vout. ), a switch driving signal Vqsw for on-off switching of the internal switching transistors Q1 to Q4 of the pulse width modulator 130 according to a unipolar synchronization method is generated and provided to the pulse width modulator 130. Since the present invention is a unipolar synchronous method, the amplifier control unit 150 generates a high-frequency pulse signal Vpg and a high-frequency carrier signal Vcar whose frequency and phase are all synchronized and utilizes it to generate the switch driving signal Vqsw. .

[Fig. 16] is a diagram showing a first embodiment of a unipolar synchronous isolation type Class D audio amplifier of a center tap transformer according to the present invention.

Referring to FIG. 16, in the first embodiment of the unipolar synchronous insulation type Class D audio amplifier of the center tap transformer according to the present invention, a full bridge topology is adopted for the high frequency pulse generator 210 and the high frequency transformer 220 The secondary side of) is composed of a center tapped winding, and the pulse width modulator 230 is driven in a unipolar manner, and the frequency and phase of the high frequency pulse generator 210 and the pulse width modulator 230 are [Fig. 17] ] And [Fig. 19], they are synchronized with each other. In this case, the frequency of the high frequency pulse generator 210 may be set to be greater or less than the frequency of the pulse width modulator 230.

[Fig. 17] is a diagram showing a detailed operation waveform when the audio output signal Vout> 0 in the case of unipolar operation of the circuit according to the first embodiment of the present invention, and [Fig. 18] is a diagram showing a case where Vout> 0 A diagram showing a conduction path for each time section during one switching period of the high frequency pulse generator 210. Referring to FIG. 17, the switches Q2 and Q4 signals of the pulse width modulator 230 are pulse-width modulated according to an error between the audio output signal Vout and the audio input signal Vs, and thus operate complementarily.

In addition, [Fig. 19] is a diagram showing detailed operation waveforms when the audio output signal Vout <0 in the case of unipolar operation of the circuit according to the first embodiment of the present invention, [Fig. 20] In this case, a diagram showing a conduction path for each time section during one switching period of the high frequency pulse generator 210. Referring to FIG. 19, the switches Q1 and Q3 signals of the pulse width modulator 230 are pulse-width modulated according to an error between the audio output signal Vout and the audio input signal Vs, and operate complementarily.

17 and 19, in the first embodiment of the unipolar synchronous insulation type Class D audio amplifier of the center tap transformer according to the present invention, one switch M1, M2 in the high frequency pulse generator 210 A group of, and switches M3 and M4 form another group, and these two switch groups operate at a fixed frequency and 50% fixed duty, respectively.

21 is a view showing a second embodiment of a unipolar synchronous insulation type Class D audio amplifier of a center tap transformer according to the present invention.

Referring to Fig. 21, in the second embodiment of the unipolar synchronous insulation type Class D audio amplifier of the center tap transformer according to the present invention, the input voltage is applied by being divided into C1 and C2 by Vin/2, respectively, and the high frequency pulse The half-bridge topology is adopted for the generator 310 and the secondary side of the high frequency transformer 320 is composed of a center tapped winding, and the pulse width modulator 330 is driven in a unipolar manner and a high frequency pulse generator 310 ) And the frequency and phase of the pulse width modulator 330 are synchronized with each other as shown in [FIG. 22] and [FIG. 23]. In this case, the frequency of the high frequency pulse generator 310 may be set to be greater or less than the frequency of the pulse width modulator 330.

[Fig. 22] is a diagram showing detailed operation waveforms when the audio output signal Vout> 0 in the case of unipolar operation of the circuit according to the second embodiment of the present invention, and [Fig. 23] is a second embodiment of the present invention. A diagram showing detailed operation waveforms when the audio output signal Vout <0 when the circuit according to the example is unipolarly operated. 22 and 23, in the high frequency pulse generator 310, the switches M1 and M2 operate complementarily with a fixed frequency and a fixed duty of 50%, and the overall operation is the high frequency pulse generator 310 It is similar to the first embodiment except ).

[Fig. 24] is a diagram showing a third embodiment of a unipolar synchronous insulation type Class D audio amplifier of a center tap transformer according to the present invention.

Referring to FIG. 24, in the third embodiment of the unipolar synchronous insulation type Class D audio amplifier of the center tap transformer according to the present invention, a push-pull topology is adopted for the high frequency pulse generator 410, and the high frequency transformer 420 The secondary side of) is composed of a center tapped winding, the pulse width modulator 430 is driven in a unipolar manner, and the frequency and phase of the high frequency pulse generator 410 and the pulse width modulator 430 are [Fig. 25 ] And [Fig. 26], they are synchronized with each other. In this case, the frequency of the high frequency pulse generator 410 may be set to be greater or less than the frequency of the pulse width modulator 430.

[Fig. 25] is a diagram showing detailed operation waveforms when the audio output signal Vout> 0 when the circuit according to the third embodiment of the present invention is unipolarly operated, and [Fig. 26] is a third embodiment of the present invention. A diagram showing detailed operation waveforms when the audio output signal Vout <0 when the circuit according to the example is unipolarly operated. Referring to [FIG. 25] and [FIG. 26], switches M1 and M2 in the high frequency pulse generator 410 operate complementarily with a fixed frequency and a fixed duty of 50%, and the overall operation is the high frequency pulse generator 410 It is similar to the first embodiment except ).

100: Isolated Class D audio amplifier
110, 210, 310, 410: high frequency pulse generator
120, 220, 320, 420: high frequency transformer
130, 230, 330, 430: pulse width modulator
140, 240, 340, 440: low pass filter unit
150: amplifier control unit
151: Scaler
152: subtractor
153: switching control unit

Claims (5)

A high-frequency pulse generator 210, 310, and 410 having any one of a full-bridge circuit, a half-bridge circuit, and a push-pull circuit, thereby generating a high-frequency pulse signal Vpg by switching the DC input power supply Vin at high speed;
The secondary side electrically insulates the amplifier output terminal from the DC input power source Vin through an internal center tap insulation transformer element composed of a center tap winding, and the high frequency pulse signal Vpg by the turn ratio of the center tap insulation transformer element. A high-frequency transformer (220, 320, 420) for generating a high-frequency transforming pulse signal (Vpm) obtained by boosting or stepping down the pulse amplitude of the DC input power (Vin) level;
The high-frequency transformation by switching on and off a plurality of internal switching transistors (Q1 to Q4) constituting a current flow path for the high-frequency transformation pulse signal (Vpm) in accordance with a switch driving signal (Vqsw) for switching and controlling a unipolar synchronous method A pulse width modulator (230, 330, 430) generating a high frequency PWM switching signal (Vpwm) obtained by modulating a pulse width of a pulse signal (Vpm) in a unipolar synchronous manner in response to the switch driving signal (Vqsw);
A low-pass filter unit (240, 340, 440) configured to generate an audio output signal (Vout) by low-pass filtering the high-frequency PWM switching signal (Vpwm);
Receiving feedback from the audio output signal Vout output from the low-pass filter units 240, 340, 440, and generating a high frequency carrier signal Vcar whose frequency and phase are synchronized with the high frequency pulse signal Vpg, , In response to an error between the audio input signal Vs and the audio output signal Vout, the internal switching transistors Q1 to Q4 of the pulse width modulators 230, 330, and 430 are united from the carrier signal Vcar. An amplifier control unit 150 for generating the switch driving signal Vqsw for on-off switching according to a polar synchronization method and providing it to the pulse width modulators 230, 330, and 430;
It is composed including,
The center tap insulating transformer element of the high frequency transformers 220 and 320 is configured to form a path for subtracting electric energy from the parasitic capacitors of the internal switching transistors M1 to M4 of the high frequency pulse generators 210 and 310,
The high-frequency pulse generators 210 and 310 switch the DC input power Vin at a fixed frequency and a fixed duty of 50% at a high speed to the internal switching transistors M1 to M4 of the high-frequency pulse generators 210 and 310. Unipolar synchronous isolation type Class D audio amplifier of a center tap transformer, characterized in that configured to perform zero voltage switching for.
delete The method according to claim 1,
The high frequency pulse generator 410 is configured in a push-pull topology,
The pulse width modulator 430 is driven in a unipolar manner,
A unipolar synchronous isolation type Class D audio amplifier of a center tap transformer, characterized in that the frequency and phase of the high frequency pulse generator 410 and the pulse width modulator 430 are synchronized with each other.
The method according to claim 1,
The amplifier control unit 150,
A scaler (151) for matching a voltage scale between the audio input signal (Vs) and the audio output signal (Vout);
A subtractor (152) obtaining an error value between the audio input signal (Vs) and the audio output signal (Vout);
In order to generate a high frequency carrier signal Vcar in which the frequency and phase are synchronized with the high frequency pulse signal Vpg, the audio output signal Vout is increased or decreased so that the error value converges to zero. From the carrier signal Vcar, the switch driving signal Vqsw for switching on and off the internal switching transistors Q1 to Q4 of the pulse width modulators 230, 330, and 430 according to a unipolar method is generated to generate the pulse width. A switching control unit 153 provided to the modulators 230, 330, and 430;
Unipolar synchronous isolation type Class D audio amplifier of the center tap transformer, characterized in that configured to include.
The method according to claim 1,
The high-frequency transforming unit 220, 320, 420,
LC resonance tank configured to be connected in series with the primary side or secondary side of the center tap insulation transformer element by matching its own resonance frequency with the switching frequency (fpg) of the high frequency pulse generators 210, 310, 410 Lr, Cr);
Unipolar synchronous isolation type Class D audio amplifier of a center tap transformer, characterized in that it comprises a.

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