KR101594303B1 - Phase-Shifted Dual Full-Bridge Converter - Google Patents

Abstract

The present invention relates to a phase-shifted dual full-bridge converter which comprises: a pair of transformers; a first stage including first and second legs composed of a pair of switches connected in series, a lagging full-bridge inverter including a first transformer from the pair of transformers, third and fourth legs, and a leading full-bridge inverter including a second transformer from the pair of transformers; and a second stage including fifth, sixth, and seventh legs composed of a pair of diodes connected in series, a first full-bridge rectifying circuit including the first transformer, a second full-bridge rectifying unit including the sixth and seventh legs and the second transformer. Accordingly, the phase-shifted dual full-bridge converter has a small power loss and high conversion efficiency during the total charging process.

Description

Phase-Shifted Dual Full-Bridge Converter < RTI ID = 0.0 >

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a phase-shift dual full bridge converter, and more particularly, to a phase-change dual full bridge converter capable of reducing power loss and improving conversion efficiency by simplifying a structure of a rectification stage and using a diode of a low- .

Due to accelerated global warming, the reduction of natural resources, increased fuel prices, and economic problems, hybrid electric vehicles, plug-in hybrid electric vehicles (PHEVs), battery electric vehicles and fuel cell vehicles Electric propulsion vehicles are gradually increasing.

These vehicles generally require a rechargeable battery as an energy source. Among them, a plug-in hybrid electric vehicle or a battery electric vehicle requires a higher capacity and larger size battery pack than other vehicles because the battery is the main energy source of the vehicle.

These high energy density battery packs are charged from the AC power grid through an AC-to-DC converter, commonly referred to as a battery charger. In order to have low harmonics and high efficiency for the AC grid, most of the battery chargers have a basic form of an AC-DC converter that generally includes an isolated DC-DC converter and a power factor corrector (PFC).

There are several key requirements for the development of these battery chargers. First, its size and weight must be essentially reduced to facilitate packaging and increase energy utilization. That is, high power density and low weight design are required. In addition, to maximize fuel economy and emission reduction, conversion efficiency must be maximized throughout the battery recharging process or at all output conditions.

In order to achieve these requirements, it is necessary to adopt a high switching frequency and a soft switching technique. Here, the high switching frequency is key to reducing the size and weight of passive components such as transformers, inductors, capacitors, etc. used in high power applications. And soft switching technology is key to lower switching losses.

PFC, on the other hand, requires a bridgeless design to reduce excessive conduction losses due to forward voltage drop across the bridge diodes. Particularly, at low voltage, the current rises and the conduction loss becomes larger. This conduction loss deteriorates the overall efficiency and increases the size and weight of the heatsink.

Also, in order to obtain higher conversion efficiency, especially at high power levels, an interleaving or parallel scheme that reduces conduction losses needs to be considered.

Also, when the output voltage of the battery charger is high, the rectifier diode of the DC-DC converter can experience severe voltage spikes and oscillations. This requires lossy snubber circuits and higher voltage rated diodes, which cause power loss and increase in size. Therefore, at the rectification stage of the DC-DC converter, a design that can prevent this problem should also be considered.

The present invention proposes a phase-change dual full-bridge converter capable of reducing power loss and improving conversion efficiency by simplifying the structure of a rectification stage and using a diode of a low voltage rating.

This object is achieved by a transformer comprising: a pair of transformers; A legged full bridge inverter comprising a first leg, a second leg and a first transformer of a pair of transformers, each of the legs comprising a pair of series-connected switches; and a third leg, a fourth leg, and a pair of transformers A first stage including a leading full bridge inverter including a second transformer; A first full bridge rectifying circuit including a fifth leg, a sixth leg and a seventh leg composed of a pair of diodes connected in series, the fifth leg, the sixth leg, and the first transformer; And a second stage composed of a first full bridge rectifier circuit including first, sixth, seventh, and second transformers.

This object is achieved by a transformer comprising: a pair of transformers; A legged full bridge inverter comprising a first leg, a second leg and a pair of transformers, the leg comprising a pair of series-connected switches and a primary side of a first transformer; and a third leg, a fourth leg, A first stage including a leading full bridge inverter including a primary side of a second transformer of the transformers of the first stage; A first full bridge rectifying circuit including a fifth leg, a sixth leg and a seventh leg composed of a pair of diodes connected in series, the fifth leg, the sixth leg and the secondary side of the first transformer, , A second full bridge rectifier circuit including the sixth leg, the seventh leg, and the secondary side of the second transformer, and the sixth leg comprises a low-voltage rated diode To-peak converter.

The full bridge converter according to the present invention can reduce the primary and secondary side conduction losses of the transformer, has no circulating current, has less duty cycle loss, and can maximize the ZVS range. In addition, the size of the output inductor can be made smaller, and the power handling capacity can be easily increased. As a result, this converter has a small power loss and high conversion efficiency during the entire charging process, so it can be used as a DC-DC converter of a high power battery charger with a power capacity of 6.6 kW or more.

1 is a circuit diagram of a full bridge converter according to the present invention,
FIG. 2 is a main operation waveform diagram of the full bridge converter of FIG. 1,
3 (a) to 3 (h) are circuit diagrams showing modes 1 to 8 of the converter circuit of Fig. 1,
Figure 4 (a) is a simplified rectifier waveform in the converter according to the invention,
4 (b) is a waveform diagram of the rectifier output voltage in the conventional converter
5 is a graph showing the relationship between the normalized DC conversion ratio and the duty cycle of the converter according to the present invention,
6 is a graph showing the value of the output inductor required in the duty cycle function,
7 is a graph showing a charge analysis by a 6.6 kW battery charger,
8 (a) is an operational circuit diagram of the full-bridge converter in the pre-charging mode according to the present invention,
FIG. 8 (b) is a graph showing the output voltage and the battery voltage at point B of the full bridge converter according to the present invention,
8 (c) is a graph and an equivalent circuit diagram showing the output voltage and the battery voltage at point C of the full bridge converter according to the present invention,
9A is a circuit diagram of a conventional single PSFB converter,
9 (b) is a circuit diagram of a conventional parallel operation PSFB converter,
10 is an equivalent circuit diagram of a rectifying stage in the power ring mode,
11 (a) to 11 (d) are graphs showing the primary side conduction loss, the primary side switching loss, the secondary side conduction loss, and the total loss,
12 is a circuit diagram showing a snubber circuit applied to a full bridge converter according to the present invention,
13 is a graph showing the main operating waveforms generated when the converter of the present invention moves from point B to point C shown in Fig. 7,
14 is a graph showing the efficiency of the full bridge converter of the present invention measured under different operating conditions.

Hereinafter, some embodiments of the present invention will be described in detail with reference to exemplary drawings. It should be noted that, in adding reference numerals to the constituent elements of the drawings, the same constituent elements are denoted by the same reference symbols as possible even if they are shown in different drawings. In the following description of the present invention, a detailed description of known functions and configurations incorporated herein will be omitted when it may make the subject matter of the present invention rather unclear.

In describing the components of the present invention, terms such as first, second, A, B, (a), and (b) may be used. These terms are intended to distinguish the constituent elements from other constituent elements, and the terms do not limit the nature, order or order of the constituent elements. When a component is described as being "connected", "coupled", or "connected" to another component, the component may be directly connected to or connected to the other component, It should be understood that an element may be "connected," "coupled," or "connected."

FIG. 1 is a circuit diagram of a full bridge converter according to the present invention, and FIG. 2 is a main operation waveform diagram of the full bridge converter of FIG.

The full bridge converter according to the present invention includes a first stage including a primary side of a first transformer T ₁ and a second transformer T ₂ connected to a power source and a second stage connected to the load and including a secondary side of the transformer .

The first stage may include a leading full-bridge inverter and a full-bridge inverter.

The legged full bridge inverter may include a first leg, a second leg, and a first transformer T ₁ . Here, the first leg is constituted by the series connection of the first switch Q ₁ and the second switch Q ₂ , and the second leg is constituted by the series connection of the third switch Q ₃ and the fourth switch Q ₄ .

The contacts of the first switch Q ₁ and the second switch Q ₂ in the first leg may be connected directly or indirectly to one end of the primary side of the first transformer T ₁ and the third switch Q ₃ and the fourth switch Q ₄ May be directly or indirectly connected to the other end of the primary side of the first transformer T ₁ .

The leading full bridge inverter may include a third leg, a fourth leg, and a second transformer T ₂ . Here, the third leg is constituted by the series connection of the fifth switch Q ₅ and the sixth switch Q ₆ , and the fourth leg is constituted by the series connection of the seventh switch Q ₇ and the eighth switch Q ₈ .

The contacts of the fifth switch Q ₅ and the sixth switch Q ₆ in the third leg may be directly or indirectly connected to one end of the primary side of the first transformer T ₁ and the seventh switch Q ₇ and the eighth switch Q ₈ May be directly or indirectly connected to the other end of the primary side of the first transformer.

The first transformer T ₁ and the second transformer T ₂ may each include a leakage inductance L _m1 , L _m2 .

The second stage may include two full bridge rectifier circuits.

The first full bridge rectifier circuit may include a fifth leg, a sixth leg, and a first transformer T ₁ . Here, the fifth leg is constituted by the series connection of the first diode D ₁ and the second diode D ₂ , and the sixth leg may be constituted by the series connection of the third diode D _a1 and the fourth diode D _a2 .

In the fifth leg, the contacts of the first diode D ₁ and the second diode D ₂ may be directly or indirectly connected to one end of the first transformer T ₁ secondary side. In the sixth leg, the contact point of the third diode D _a1 and the fourth diode D _a2 may be directly or indirectly connected to another end of the first transformer T ₁ secondary side.

The second full bridge rectifier circuit may include a sixth leg, a seventh leg, and a second transformer T ₂ . Here, the seventh leg may be constituted by a series connection of a fifth diode D ₃ and a sixth diode D ₄ .

In the sixth leg, the contacts of the third diode D _a1 and the fourth diode D _a2 may be directly or indirectly connected to one end of the secondary side of the second transformer T ₁ . In the seventh leg, the contacts of the fifth diode D ₃ and the sixth diode D ₄ may be directly or indirectly connected to the other end of the second transformer T ₂ secondary side. An output inductor L ₀ is connected between the second bridge rectifier circuit and the battery.

All the switches in the converter circuit shown in Fig. 1 are driven with a constant ratio in a state in which the dead time is ignored. The driving signals of the switches of the leading full bridge inverter are preceded by the driving signals of the switches of the legged full bridge inverter.

In the following, the switches of the leading full bridge inverter are called the leading switches, and the switches of the legged full bridge inverter are called the legging switches. Two Full-Bridge Inverters (TFBIs) are driven by regulating the phase transition time to regulate the output voltage and current.

Each switching cycle is divided into two half cycles, t ₀ -t ₈ and t ₈ -t ₁₆ . Since the operating principles of the two half cycles are symmetric, only the first half cycle is described below. This half cycle can be divided into eight modes again. In Fig. 3, operation circuits for eight modes are disclosed.

First, the following five assumptions are set to simplify the analysis.

1) The output inductor L ₀ is large enough to be considered a constant current source during the switching period.

2) The first transformer T ₁ and the second transformer T ₂ have a leakage inductance of L _lk1 and L _lk2 , _{respectively,} and a turn _ratio of n = N _S1 / N _P1 , n = N _S2 / N _P2 .

3) The excitation inductor L _m2 of the second transformer T ₂ is large enough to ignore the effect of the excitation current occurring during the switching period.

4) The main active switches are all formed of MOSFETs with parasitic diodes D _b1 , D _b2 , D _b3 , D _b4 , D _b5 , D _b6 , D _b7 , D _b8 .

5) The output capacitances of all MOSFETs have the same capacitance C _oss .

3 (a) to 3 (h) are circuit diagrams showing modes 1 to 8 of the converter circuit of Fig.

[Mode ₁ (t ₀ -t 1)]

In mode 1 shown in FIG. 3A, the fifth and seventh switches Q ₅ and Q ₇ of the reading switches and the first and third switches Q ₁ and Q ₃ of the legging switch are on, 5 Starts when current flows to diodes D ₁ , D ₃ . During mode 1, the primary voltages V _p1 (t) and V _p2 (t) have (-) and (-) values of the input voltage, respectively. Since the excitation current i _Lm1 (t) linearly increases from the initial value, whereas the excitation inductor L _m2 of the second transformer T ₂ has a very large value, the excitation current i _Lm2 (t) has a value of almost zero . The secondary voltages V _s1 (t) and V _s2 (t) each have (+) and (-) values of the input voltage having a turn ratio of n. The output voltage V _rec (t) of the rectifier becomes 2nV _IN by the sum of the secondary voltages V _s1 (t) and -V _s2 (t). In mode 1, power is transferred from the input port to the output port via the first and second transformers T ₁ , T ₂ and the first and fifth diodes D ₁ , D ₃ , and the primary current of the inverter stage is expressed as .

[Mode 2 (t ₁ -t ₂ )]

The mode 2 shown in FIG. 3 (b) starts when the fifth and seventh switches Q ₅ and Q ₇ of the reading switches at t ₁ are turned off. Then, the energy stored in the output inductor L _{O charges the} capacitors C _OSS5 and C _OSS7 linearly, and the capacitances C _OSS6 and C _OSS8 are discharged. The primary voltage V _p2 (t) increases from -V _IN to zero, and the primary voltage V _p1 (t) continues to increase V _IN , thereby continuously increasing the excitation current i _Lm1 (t). As the secondary voltage V _s2 (t) increases from -nV _IN to zero and the secondary voltage V _s1 (t) maintains nV _IN , the rectifier output voltage V _rec (t) is increased from 2nV _IN to nV _IN . The primary current in mode 2 is the same as the primary current in mode 1.

[Mode 3 (t ₂ -t ₃ )]

The mode 3 shown in Fig. 3 (c) starts when the primary voltage V _p2 (t) reaches 0 in mode 2. At the same time, the secondary voltage V _s2 (t) becomes zero and a current flows through the fourth diode D _a2 . The fourth diode D, because the state in which a current flows to _a2, mode 3 while, the secondary-side voltage V _s2 (t) continues to maintain a zero, reading the full-bridge output capacitance of all the reading switches of the inverter and leakage inductance L _lk2 Resonance occurs between the two. With this resonance, the voltage _{across the} capacitances C _OSS5 and C _OSS7 is constantly charged, and the voltage across the capacitances C _OSS6 and C _OSS8 is continuously discharged. The primary voltage V _p2 (t) is increased from zero to V _IN in a sinusoidal fashion and the primary voltage V _p1 (t) continues to maintain V _IN .

[Mode 4 (t ₃ -t ₄ )]

The mode 4 shown in Fig. 3 (d) starts in mode 3 when the primary voltage V _p2 (t) reaches V _IN . Then, the reading switches of the sixth and the eighth switch Q _6, the parasitic diode D _b6, D _b8 of Q _8, a current starts flowing, and the reading switch sixth and eighth switches Q _6, Q ₈ is a zero voltage switching (ZVS : Zero Voltage Switching). During mode 4, the secondary voltage V _s2 (t) remains at zero, whereby the input voltage V _IN is provided to a leakage inductor having a leakage inductance L _lk2 . As the input voltage V _IN is provided to the leakage inductor, the commutation between the fifth and fourth diodes D ₃ and D _a2 is improved. The primary side voltage V _p1 (t) and the rectifier output voltage V _rec (t) maintain V _IN and nV _IN , respectively. Thus, in mode 4, power is transferred from the input port to the output port through the first and second transformers T ₁ and T ₂ and the first, fourth, and fifth diodes D ₁ , D _a2 , and D ₃ , The side current can be expressed as follows.

[Mode 5 (t ₄ -t ₅ )]

The mode 5 shown in FIG. 3 (e) is started when the commutation between the fifth diode D ₃ and the fourth diode D _a2 is completed at t ₄ and a current flows through the first diode D ₁ and the fourth diode D _a2 . For example, in mode 5, if the primary current is zero, the power only moves from the input port to the output port via the first transformer T ₁ and the first and fourth diodes D ₁ , D _a2 .

[Mode 6 (t ₅ -t ₆ )]

The mode 6 shown in FIG. 3 (f) starts when the first and third switches Q ₁ and Q ₃ , which are the legging switches, are turned off at t ₅ . At the same time, a current starts to flow in the sixth diode D ₄ , and another resonance occurs in the legged full bridge inverter consisting of two leakage inductors having the output capacitance of all the _legging switches and the leakage inductance L _lk2 . With this resonance, the voltage _{across the} capacitances C _OSS1 and C _OSS3 is continuously discharged, and the voltage across the capacitances C _OSS2 and C _OSS4 is constantly charged. The primary voltage V _p1 (t) decreases from V _IN to -V _IN and the rectifier output voltage V _rec (t) drops to zero. Also, the commutation between the first diode D ₁ and the sixth diode D ₄ is improved.

[Mode 7 (t ₆ -t ₇ )]

The mode 7 shown in Fig. 3 (g) starts in mode 6 when the primary voltage V _p1 (t) reaches -V _IN . Then, leggings switch the second and fourth switches Q _2, parasitic Q _4, a diode D _b2, D for a current starts flowing _b4, the second and fourth switches Q _2, and Q ₄ is a zero-voltage switching (ZVS: Zero Voltage Switching). In mode 7, the secondary voltages V _s1 (t) and V _s2 (t) are all zero, and accordingly, the output voltage V _rec (t) of the rectifier also becomes zero. Since the output voltage V _rec (t) is zero for the rectifier, the load power is supplied from t ₆ from the stored energy in the output inductor L _0. Since the primary side voltage V _p1 (t) is -V _IN and the primary side voltage V _p2 (t) is V _IN and the secondary side voltages V _s1 (t) and V _s2 (t) are all 0, the inductance L _lk1 And two leakage inductors with L _lk2 are _equal to -V _IN and V _IN , respectively. Due to the voltage of the two leakage inductors, the primary currents i _p1 (t) and i _D1 (t) decrease linearly and the primary currents i _p2 (t) and i _D4 (t) increase linearly . The primary currents i _p1 (t) and i _D1 (t) can be expressed as:

[Mode 8 (t ₇ -t ₈ )]

The mode 8 shown in FIG. 3 (h) starts when the current i _D4 (t) flowing through the sixth diode D ₄ reaches the output current I ₀ and the first diode D ₁ is naturally turned off. At the same time, the secondary voltage V _s1 (t) becomes zero and the secondary voltage V _s2 (t) becomes equal to nV _IN . The current in mode 8 is as follows.

At the end of mode 8, i _D2 (t) reaches the output current I _o and the fourth diode D _a2 is naturally turned off. The power then moves from the input port to the output port via the first transformer T ₁ , the second transformer T ₂ , the second and sixth diodes D ₂ , D ₄ .

[Mode 9-16 (t ₈ -t ₁₆ )]

The operations from mode 9 to 16 are the same as the previous modes except for the direction of the power supply path.

Figure 4 (a) shows a simplified rectifier waveform in a converter according to the invention.

As shown in FIG. 2, the simplified rectifier output voltage is given as shown in FIG. 4 (a) because the periods of modes 2, 6, and 8 are substantially narrow.

4 (b) shows the rectifier output voltage waveform in the conventional converter.

Conventional converters are configured with a single phase-shift full-bridge (PSFB) or a topology in which two PSFBs are configured in parallel for high power design. Compared with the conventional converter, in the converter of the present invention, there is almost no time interval in which the voltage is applied to the rectifier output voltage _Vrec (t). This means that the converter according to the present invention has no circulating current on the inverter side.

When the voltage waveform of the output voltage of Fig. 4 (a) is averaged, the DC conversion ratio of this converter is given as follows.

Where T _S is the switching period.

5 is a graph showing the relationship between the normalized DC conversion ratio and the duty cycle of the converter according to the present invention.

As shown in FIG. 5, the gain of the converter according to the present invention is much higher than that of the conventional converter. As a result, the turn ratio n of the converter according to the present invention is much smaller than that of the conventional converter, thereby significantly reducing the primary side conduction loss and the secondary side voltage stress.

On the other hand, the value of the output inductor L _O can be derived from the rectifier output voltage of FIG.

Where I _PP is the peak-to-peak value of the current flowing through the output inductor L _O and V is the voltage applied to the output inductor L _O during? T.

Equation (6) is transformed as shown in Equation (7).

For quantitative analysis, the following parameters are used. The parameters are the output voltage V _O = 250V, the peak current I _PP = tupikeu 1A, the switching frequency f _s = 100kHz.

6 is a graph showing the value of the output inductor required in the duty cycle function.

FIG. 6 shows the values of the output inductor L _O required in the converter according to the present invention and the values of the output inductors required in the conventional converter. As shown, the value of the output inductor L _O required in the converter of the present invention is much smaller than the value of the output inductor required in a conventional converter. In particular, in the case of a conventional converter in the form of a converter consisting of two parallel PSFBs to provide high power, two large output inductors with large output values are required. Thus, the converter of the present invention has a very large size gain in that it has an output inductor L _O with a very small value.

As described above in mode 2 and mode 3, the ZVS operation of the reading switches occurs in two resonant states. First, the voltage across the reading switches is linearly reduced from V _IN to 0.5V _IN by the energy stored in the output inductor L _O in mode 2. [ And the voltage remaining in the output capacitors of the reading switches is completely discharged by the energy stored in the leakage inductor L _lk2 in mode 3. The ZVS state of the reading switches is obtained by:

From the energy point of view, Equation (8) can be transformed into Equation (9).

As can be seen in equation (9), the energy remaining in the output capacitors of the reading switches must be discharged by the leakage inductor L _lk2 , but 25% of the original energy remains. Thus, the ZVS of the reading switches is easily achieved by a small leakage inductor L _lk2 to exceed a wide load range. If the ripple current of the inductor L _m2 is used to achieve ZVS in a load condition such as a load-free condition, then the ZVS condition is adjusted as shown in Equation 10 below and the full ZVS operation is achieved.

Here,? I ' _ripple is the ripple current of the inductor L _m2 . Since the energy discharged by the ripple current ΔI ' _ripple is 25% of the original energy, as shown in Equation 10, the ripple current ΔI' _ripple can be designed to a very small value.

The ZVS condition of the legging switch is obtained as follows.

In the energy dimension, Equation (11) is transformed into Equation (12).

By the ripple current ΔI _ripple item in the equation (11) or 12 L _lk1 + L _lk2 items and exciting current i _Lml (t), ZVS of leggings switch can know that it can be easily carried out over a wide load range. Typically, because a large leakage inductors increases the duty cycle loss, to design the converter at best, to improve the trade-off (Trade-off) between the leakage inductor values, ripple current magnitude of ΔI _ripple is in equation (12) Should be processed.

To extend the ZVS range to a no-load state with proper leakage inductor values, this converter uses the excitation inductor L _m1 (or ripple current ΔI _ripple ) of the first transformer T ₁ . Despite the use of the small excitation inductance of the first transformer T ₁ , the effective current stress of the legged full bridge inverter does not increase. This is because the average of the excitation current in the first transformer T ₁ in the half-switching period is zero and its contribution to the total effective current in the overload state is negligibly small.

Generally, to extend the ZVS range, adding a large resonant inductor in series with the transformer reduces the effective duty cycle or increases the duty cycle loss. To compensate for this, the turn ratio n of the transformer is designed to be large, thereby increasing the primary side conduction loss and the secondary side voltage stress. However, the ZVS range of this converter is extended using the excitation inductance of the first transformer T ₁ , in particular parallel to the path powered by the leggings full-bridge inductor. As a result, the extended ZVS range does not affect the duty cycle loss.

FIG. 7 is a graph showing an analysis of charging by a 6.6 kW battery charger.

Initially, the DC-DC converter gradually increases the charging current in a step-wise fashion to prevent damage to the battery, and the charging current is maintained until each battery cell voltage reaches a threshold voltage or the battery voltage reaches point A . This initial charging step is referred to as a pre-charging mode.

After the precharging mode, the DC-DC converter is switched to a constant current mode and the battery voltage is charged linearly. At this time, the maximum battery voltage is charged until it reaches 450V or point C. When the battery voltage reaches point C, the DC-DC converter is placed in a constant voltage mode, and the charging current gradually decreases. This mode is terminated when the charge current reaches a preset settling current.

Since the power is small in the precharging mode, the leading full bridge inverter is not operated and the phase transition time T _alpha between the two legs of the legged full bridge inverter is adjusted as shown in Fig. 8 (a) Only the inverter should charge the battery. In the precharging mode, the leggings full bridge inverter operates like a conventional single PSFB converter shown in Fig. 9 (a). As the output voltage or the battery voltage increases during the pre-charging mode, the phase transition time T _alpha decreases. When the battery voltage reaches point A, the phase transition time T _α becomes zero, the gate signal Q ₁ becomes equal to Q ₃ , or the gate signal Q ₂ becomes equal to Q ₄ .

At the same time, when the leading full bridge inverter is operated, the converter adjusts the phase transition time T _? Between the two inverters shown in FIG. 2 so that the present converter operates. Most of the power at point B moves to the battery through the legged full bridge inverter as shown in Fig. 8 (b) because the battery voltage is still low and the phase transition time T _? Is small. As the charging point moves from point A to point C, the battery voltage or charging power increases and the phase transition time T _? Also increases. Then, as shown in Fig. 8 (c), the power processed by the leading full bridge inverter gradually increases, and each inverter naturally processes half of the total power at the maximum power point C. [ These operations allow the converter to achieve high efficiency through the entire battery charging process.

The loss distribution is analyzed below to illustrate the fundamentals of high efficiency of this converter.

The performance of this converter is compared with the conventional converter shown in Fig.

10 is an equivalent circuit diagram of a rectifying stage in two powering modes. Referring to FIG. 10, the voltage stress of the rectifier diode is compared as follows.

The stress of the rectifier diode from the equivalent circuit can be obtained as shown in Table 1. The rectifier diode voltage stresses of the two conventional converters are also shown in Table 1.

Diode voltage stress The converter Single PSFB Converter Parallel Operation PSFB Converter Diodes D1 to D4 2nV _IN (488V) n _c1 V _IN (560V) n _c2 V _IN (488V) Diodes D5 to D8 - - n _c2 V _IN (488V) Diodes Da1 to Da2 nV _IN (244V) - -

Where n is the turn ratio of the present converter, and n _c1 and n _c2 are the turn ratio of two conventional converters, respectively. For quantitative analysis, it is assumed that the input voltage is 400V and the converters operate as the charge analysis shown in FIG. The turn ratio of the transformers can then be calculated using Equations 8 and 9. However, due to the dead time between the actual parameters, for example, the drive signals by the leakage inductors and the duty cycle loss, the turn ratio should actually be slightly larger than the value calculated in equation (5). Considering this, the turn ratios are determined as n = 0.61, n _c1 = 1.4, and n _c2 = 1.22, respectively. As a result, it can be seen that this converter requires four high voltage rated diodes and two low voltage rated diodes.

On the other hand, both conventional converters only require high-voltage rated diodes. The low-voltage rated diode has much lower on-state voltage and better reverse recovery characteristics than the high-voltage rated diode. The use of two low-voltage-rated diodes in the center leg of the rectifier end of the converter thereby reduces the power loss occurring in the rectification stages of the pre-charging mode and the constant current mode and improves the converter efficiency. Since the powering in the constant voltage mode is also performed using two low voltage rated diodes, the loss can also be reduced under the constant voltage mode.

Comparisons of losses between this converter and a conventional converter were performed at points B, C, and D, respectively, in Fig. First, as shown in Fig. 11, the primary side conduction loss of the present converter is much lower than that of the conventional converter. This is because the legged full bridge inverter handles most of the power at point B in this converter, and the power of maximum power point C is processed in parallel by both inverters. A lower turn ratio contributes to a reduction in primary side conduction losses. The primary side switching loss of this converter has the advantage of reducing the output power. It is due to the wide ZVS range. Compared to conventional converters, the secondary side conduction losses of this converter have been improved by the placement of two low-voltage rated diodes.

11 (d) is a graph showing the sum of the primary side conduction loss, the primary side switching loss, and the secondary side conduction loss. Because of the improvement in power loss as shown in Figure 11 (d), this converter is capable of achieving a higher conversion efficiency over conventional converters during the entire battery charging process.

Table 2 compares the number of components between this converter and conventional converters.

The converter Single PSFB Converter Parallel Operation PSFB Converter switch 8 4 8 diode 6 4 8 Tracer 2 One 2 Resonance inductor 0 One 2 Output inductor One One 2 sum 17 11 22

As shown in Table 2, although the present converter processes a high power of 6.6 kW, the number of components is smaller than a conventional parallel-operated PSFB converter.

In summary, depending on the power loss and the number of components in FIG. 11, this converter is said to be suitable for use with battery chargers having a high power capacity greater than 6.6 kW over conventional converters.

To demonstrate the efficiency of this converter, a specimen converter of 3.3kW power capacity was implemented in a battery charger with the following conditions.

Here, the battery charger has an input voltage V _IN = 400 V, an output voltage V _O = 250-450 V, a maximum output current I _{O (MAX)} = 7.33 A, and a switching frequency f _S = 100 kHz.

The sample converter was fabricated using the components listed in Table 3.

Switch (Q ₁ ~ Q ₈₎ IPW65R080CFDA (650V) The rectifying diodes (D ₁ to D ₄ ) IDH15S120 (1200 V, 15 A, V _f = 2.5 V) The auxiliary diodes ( _Da1 , _Da2 ) FEP16GTA (400 V, 16 A, V _f = 1.3 V) Transformers (T ₁ , T ₂ ) Core: EE7072, turn ratio: 0.61
The first transformer second transformer
L _m : 855.8 μH L _m : 845.2 μH
L _lk: 8.46μH L _lk: 8.32μH Output inductor (L _O ) 200 μH, MPP core Output capacitor (C _o ) 47μF / 650V controller TMS320F28027

In order to reduce the voltage overshoot and the oscillation problem, two snubber circuits are employed in the rectification stage, as shown in Fig.

13 is a graph showing the main operation waveforms generated when the converter of the present invention moves from point B to point C shown in Fig.

As shown, all of the measured waveforms follow the theoretical waveforms described in FIGS. 2 and 8 well. In addition, the converter is confirmed to have no circulating current on the primary side during battery charging. Also, looking at the rectifier output voltage waveform in Figure 13, when power is shifted from point B to point C in the present converter, two low-voltage rated diodes are delivered using a smaller on-state voltage. This operation contributes to reduction of the conduction loss of the rectification stage.

14 is a graph showing the efficiency of the full bridge converter of the present invention measured under different operating conditions.

As shown in FIG. 14, the converter has a maximum load or efficiency of 96.65% at a maximum at point C in FIG. 7, and maintains high efficiency in the entire battery charging process. This high efficiency can be achieved because the ZVS operates under all operating conditions, has no duty cycle loss effect, no circulating current, and uses two low-voltage rated diodes.

Thus, the soft switching DC-DC converter according to the present invention is applied to an onboard high power battery charger having a power capacity exceeding 6.6 kW in an electric vehicle. The converter according to the present invention has the following advantages.

1) Reduce primary and secondary conduction losses.

2) No circulating current, less loss of duty cycle, and maximized ZVS range.

3) The size of the output inductor can be made smaller.

4) Power handling capacity can be easily increased.

With these advantages, the converter has a small power loss and high conversion efficiency during the entire charging process. Experimental results of this converter implemented in a 3kW battery charger demonstrate all the advantages mentioned above and confirmed the efficiency and practicality of the converter. As a result, this converter is suitable for use as a DC-DC converter of a high-power battery charger with a power capacity of 6.6 kW or more, which is used in electric vehicles and plug-in hybrid electric automobiles.

The standard content or standard documents referred to in the above-mentioned embodiments constitute a part of this specification, for the sake of simplicity of description of the specification. Therefore, it is to be understood that the content of the above standard content and portions of the standard documents are added to or contained in the scope of the present invention.

The foregoing description is merely illustrative of the technical idea of the present invention and various changes and modifications may be made by those skilled in the art without departing from the essential characteristics of the present invention. Therefore, the embodiments disclosed in the present invention are intended to illustrate rather than limit the scope of the present invention, and the scope of the technical idea of the present invention is not limited by these embodiments. The scope of protection of the present invention should be construed according to the following claims, and all technical ideas within the scope of equivalents should be construed as falling within the scope of the present invention.

Claims (9)

A pair of transformers;
A legged full bridge inverter comprising a first leg, a second leg and a first transformer of a pair of transformers, each of the legs comprising a pair of series-connected switches; and a third leg, a fourth leg, and a pair of transformers A first stage including a leading full bridge inverter including a second transformer; And
A first full bridge rectifying circuit including a fifth leg, a sixth leg and a seventh leg composed of a pair of diodes connected in series, the fifth leg, the sixth leg, and the first transformer; And a second stage comprising a second full bridge rectifier circuit including a sixth leg, a seventh leg, and the second transformer,
The legging full bridge inverter only charges the battery by adjusting the phase transition time between the first leg and the second leg of the legging full bridge inverter until the battery voltage reaches a preset threshold voltage,
Adjusting the phase transition time between the third leg and the fourth leg of the leading full bridge inverter to increase power to be processed in the leading full bridge inverter when the battery voltage reaches a preset threshold voltage,
A snubber circuit is mounted in parallel to each of the pair of low-voltage rated diodes constituting the sixth leg,
And the inductance (Lo) of the output inductor connected in series to the output stage of the second stage is determined according to the following equation.

Where Ipp is the peak-to-peak value of the current flowing into the output inductor, Vo is the output voltage, Ts is the switching period, and D is the duty of the switch located in the first stage. The method according to claim 1,
And the pair of diodes constituting the sixth leg are low-voltage rated diodes. The method according to claim 1,
A precharge mode in which the charge current is increased until the battery voltage reaches a predetermined threshold voltage, a constant current mode in which the battery voltage is linearly charged, a constant voltage mode in which the charge current is reduced when the battery voltage reaches a predetermined charge voltage And the battery is charged through the battery. delete delete delete The method according to claim 1,
Wherein the excitation inductor of the second transformer is sufficiently large to neglect the effect of the excitation current occurring during the switching period. The method according to claim 1,
Wherein the ZVS range is extended by the excitation inductance of the first transformer of the legged full bridge inductor parallel to the power path. delete

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