KR101204470B1 - Signal converting apparatus and receiving apparatus for supporting dual bandwidth in wireless communication system - Google Patents

Abstract

A receiving apparatus of a wireless communication system is provided. The receiving device amplifies an antenna for receiving a radio frequency signal comprising a signal of a first frequency band and a signal of a second frequency band, amplifies the radio frequency signal, and outputs the signal of the first frequency band as a differential phase signal, A low noise amplifier for outputting the signal of the second frequency band as a common phase signal, a differential amplifier for passing only the differential phase signal among the signals output from the low noise amplifier, and only the common phase signal among the signals output from the low noise amplifier To a mixer.

Description

SIGNAL CONVERTING APPARATUS AND RECEIVING APPARATUS FOR SUPPORTING DUAL BANDWIDTH IN WIRELESS COMMUNICATION SYSTEM}

The present invention relates to a signal conversion device and a reception device supporting dual band in a wireless communication system.

There is an active research on intelligent RF (Radio Frequency) receivers that can handle various wireless communication applications in various frequency bands. The intelligent RF receiver has the advantage of lowering the unit cost by increasing the flexibility of the chip, but has the disadvantage of lowering the quality of wireless communication due to interference between frequency bands corresponding to each wireless communication application field.

Currently, intelligent RF receivers can be implemented in wideband or multiple bands. Broadband intelligent RF receivers, which can handle a wide frequency band with one receiver, can achieve the effect of implementing multiple systems at low cost, but they consume high power and have high interference problems between multiple frequency bands. On the other hand, a multi-band intelligent RF receiver that realizes narrowband characteristics in two or more frequency bands handles only the required frequency bands so that power consumption is low, but interference problems still exist.

The technical problem to be solved by the present invention is to provide an intelligent RF receiving apparatus that can minimize the interference between frequency bands.

An apparatus for receiving a wireless communication system according to an aspect of the present invention includes an antenna for receiving a radio frequency signal comprising a signal of a first frequency band and a signal of a second frequency band, amplifying the radio frequency signal, and amplifying the first frequency. A low noise amplifier which outputs a signal of a band as a differential phase signal and outputs a signal of the second frequency band as a common phase signal, a differential amplifier which passes only the differential phase signal among the signals output from the low noise amplifier, and the low noise amplifier The mixer outputs a mixer that passes only the common phase signal.

A load circuit of a low noise amplifier according to an aspect of the present invention includes a first capacitor, a second capacitor, a third capacitor, a first inductor, and a second inductor, wherein the third capacitor is between the first node and the second node. One end of the first inductor and the first capacitor is connected between the first node and the third capacitor, and the second inductor and the second capacitor are connected between the second node and the third capacitor. The other end of the first inductor, the first capacitor, the second inductor, and the second capacitor is connected to ground.

A low noise amplifier according to an aspect of the present invention is connected to an input terminal, an impedance matching unit for matching a single phase signal input through the input terminal, and the impedance matching unit, and one end is symmetrically connected to each other through a first tap. And a first parasitic capacitor and a second capacitor symmetrical with respect to the first inductor, the second inductor, and the first tap, and converting the single phase signal output from the impedance matcher into a differential phase signal and a common phase signal. And a first intermediate tap inductor.

According to an embodiment of the present invention, in a wireless communication system for receiving a signal of a single phase type, two different wideband signals having a bandwidth of 1 GHz or more are received and isolation between input and output by minimizing reflection and noise figure of the received signal After securing the signal of each band can be amplified.

In addition, in processing two different single-phase signals having wideband characteristics, the signals are output in a precise differential phase form for low frequency band signals and in a common common phase form for high frequency signals. Outputs The signals of each frequency band output in this way are transferred to the next stage without interference to the relative signals.

In addition, since the receiving device according to the embodiment of the present invention uses a circuit used as a load of the low noise amplifier stage, it does not need an additional amplifier stage to obtain a dual band, it can be implemented in a simple and small area.

The receiving device according to the embodiment of the present invention can be applied to both the direct conversion (conversion) and the heterogeneous receiving device.

1 is a view showing a receiving device of a wireless communication system according to an embodiment of the present invention.
2 is a diagram illustrating a load unit 114 of the signal conversion device according to the first embodiment of the present invention.
FIG. 3 is a graph showing simulation results of transfer function values at nodes P1 and P2 in the circuit diagram of FIG. 2.
FIG. 4 is a graph showing phases at the nodes P1 and P2 in the circuit diagram of FIG. 2.
FIG. 5 is a graph illustrating a phase difference between a node P1 and a node P2 in the circuit diagram of FIG. 2.
6 is a diagram illustrating a load unit 114 of the signal conversion device according to the second embodiment of the present invention.
FIG. 7 is a graph illustrating simulation results of transfer function values at nodes P1 and P2 in the circuit diagram of FIG. 6.
8 is a diagram showing a load unit 114 of the signal conversion apparatus according to the third embodiment of the present invention.
FIG. 9 is a graph illustrating simulation results of transfer function values at nodes P1 and P2 in the circuit diagram of FIG. 8.
FIG. 10 shows an example of applying the circuit diagram of FIG. 2 to a low noise amplifier design having two stages.
FIG. 11 is a graph illustrating gain and phase difference characteristics of the circuit diagram of FIG. 10.
FIG. 12 is a graph illustrating a transient simulation result of the dual band output signal of the circuit diagram of FIG. 10.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings so that those skilled in the art can easily carry out the present invention. As those skilled in the art would realize, the described embodiments may be modified in various different ways, all without departing from the spirit or scope of the present invention. In the drawings, parts irrelevant to the description are omitted in order to clearly describe the present invention, and like reference numerals designate like parts throughout the specification.

Throughout the specification and claims, when a part includes a certain component, it means that it can further include other components, except to exclude other components unless specifically stated otherwise.

Now, a signal conversion device and a reception device of a wireless communication system according to an embodiment of the present invention will be described in detail with reference to the accompanying drawings.

1 is a view showing a receiving device of a wireless communication system according to an embodiment of the present invention.

Referring to FIG. 1, a receiver of a wireless communication system according to an exemplary embodiment of the present invention includes an antenna 100, a low noise amplifier (LNA) 110, a differential amplifier 120, a mixer 130, and a frequency. Down converters 140-1 and 140-2.

The antenna 100 directly receives a radio frequency (RF) signal from a transmitting device of a wireless communication system.

The low noise amplifier 110 includes a transconductance (Gm) part 112 for converting an input voltage into a current and a load part 114 for converting the current back to a voltage. The low noise amplifier 110 amplifies the selected signal while minimizing noise. The low noise amplifier 110 converts the amplified signal into a differential phase signal and a common phase signal for each frequency band. Thus, in this specification, the low noise amplifier 110 may be mixed with a signal conversion device.

The differential amplifier 120 passes only the differential phase signal among the signals output from the low noise amplifier 110 and removes the common phase signal.

The mixer 130 passes only in-phase phase signals among the signals output from the low noise amplifier 110 and removes the differential phase signals.

The frequency down converter 140-1 converts the differential phase signal output from the differential amplifier 120 to a predetermined frequency using a local oscillator.

The frequency down converter 140-2 converts the common phase signal output from the mixer 130 to a predetermined frequency using a local oscillator.

Here, the differential amplifier 120 and the mixer 130 are shown as separate components from the frequency down converters 140-1 and 140-2, but the differential amplifier 120 and the mixer 130 have a frequency down converter ( 140-1 and 140-2 may be implemented in the form of a transconductance.

Next, a signal conversion apparatus of a wireless communication system according to an embodiment of the present invention will be described in detail with reference to FIGS. 2 to 12.

2 is a diagram illustrating a load unit 114 of the signal conversion device according to the first embodiment of the present invention.

Referring to FIG. 2, the load unit 114 of the signal converter includes capacitors C1, C2, and C3 and inductors L1 and L2. For convenience of description, it is assumed that capacitors C1, C2, C3 and inductors L1, L2 are ideal passive elements.

Capacitor C3 is connected between node P1 and node P2, one end of capacitor C1 and inductor L1 is connected between node P1 and capacitor C3, and capacitor C2 and One end of the inductor L2 is connected between the node P2 and the capacitor C3. The other ends of the capacitors C1 and C2 and the inductors L1 and L2 are connected to ground.

A large impedance value is applied to the nodes P1 and P2 similar to the open state. In addition, C = C1 = C2 and L = L1 = L2. Therefore, the impedance Za at the node P1 may be represented by Equation 1 below.

Where C is the capacitance and L is the inductance. As in Equations 2 and 4, parallel resonance occurs in the low frequency band ω _low and the high frequency band ω _high , and the impedance Za becomes infinite. On the other hand, as shown in Equation 3, series resonance occurs in the intermediate frequency band ω _mid , and the impedance Za becomes zero.

Meanwhile, when an arbitrary input signal Vs is applied to the node P1, a transfer function for confirming the signal V _P1 at the node P1 may be expressed by Equation 5.

Where ω is the angular frequency (2 [pi] f), C is the capacitance, L is the inductance, and Rs is the resistance of the arbitrary input signal Vs. Since it is a circuit when acting as the load 114 of the low noise amplifier 110, Rs can be set to a very large value. The transfer function value of Equation 5 is 1 at the low frequency of Equation 2 and the high frequency of Equation 4, and the transfer function value of Equation 5 is 1 at the intermediate frequency of Equation 3.

Similarly, when the arbitrary input signal Vs is applied to the node P2, the transfer function for confirming the signal V _P2 at the node P2 may be expressed as in Equation (6).

Where ω is the angular frequency (2 [pi] f), C is the capacitance, L is the inductance, and Rs is the resistance of the arbitrary input signal Vs. The denominators of Equations 6 and 5 are the same. This means that in the case of parallel resonance, the transfer functions of the nodes P1 and P2 are the same, but in the case of series resonance, the transfer functions of the nodes P1 and P2 are different.

FIG. 3 is a graph showing simulation results of transfer function values at nodes P1 and P2 in the circuit diagram of FIG. 2.

Referring to FIG. 3, the graph drawn by the solid line represents the transfer function of Equation 5, that is, the transfer function at node P1 in dB units, and the graph drawn by the dotted line is the transfer function of Equation 6, that is, node P2. The transfer function in) is expressed in dB. In the case of parallel resonance, which is a low frequency and a high frequency, the transfer function of Equation 5 and the transfer function of Equation 6 have the same value, but in the case of an intermediate frequency series resonance, it has a different value.

As a result of the simulation, the reason why the low frequency and the high frequency do not become 1 (0 in dB) is because the Rs value is set very large. If you set the resolution of the simulation to a very large value, the transfer function value can be 1. The same reason is that the transfer function does not become 0 (negative infinity in dB) at intermediate frequencies.

FIG. 4 is a graph showing phases at the nodes P1 and P2 in the circuit diagram of FIG. 2.

Referring to FIG. 4, the graph drawn by the solid line represents the phase characteristic at the node P1, and the graph drawn by the dotted line represents the phase characteristic at the node P2. According to the inherent characteristics of the LC resonant circuit, the phase changes by -180 degrees in the case of parallel resonance, and the +180 degrees in the case of series resonance. The phase characteristic of node P1 is phase shifted from +90 degrees to -90 degrees due to parallel resonance at the low frequency of Equation 2, and phase shifted from -90 degrees to +90 degrees due to series resonance at Equation 3 intermediate frequency. Change. Similarly, at high frequencies in Equation 4, due to parallel resonance, the phase changes again from +90 degrees to -90 degrees. On the other hand, since the phase characteristic of the node P2 does not include the series resonance, the phase change of -180 degrees occurs at the low frequency of Equation 2 and the high frequency of Equation 4, respectively, and finally has a phase of -450 degrees. .

FIG. 5 is a graph illustrating a phase difference between a node P1 and a node P2 in the circuit diagram of FIG. 2.

Referring to FIG. 5, since there is a series resonance only at the node P1 and there is no series resonance at the node P2, the phase changes only +180 degrees at the node P1. Therefore, the phase difference between node P1 and node P2 in the band before the intermediate frequency of Equation 3 is 180 degrees, and the phase difference between node P1 and node P2 in the band after the intermediate frequency of Equation 4 is 360 degrees. That is, a low frequency of Equation 2 generates a differential phase characteristic of 180 degrees between nodes, and a high frequency of Equation 4 generates a signal having in-phase phase characteristics of 360 degrees between nodes.

As such, when the circuit of FIG. 2 is used as the load 114 of the low noise amplifier 110, the dual band wireless communication system can provide high gain in each frequency band while completely eliminating interference between the frequency bands. It can be seen.

6 is a diagram illustrating a load unit 114 of the signal conversion device according to the second embodiment of the present invention.

Referring to FIG. 6, the varactor Va, which is a variable capacitor, is connected in parallel to the capacitor C3 of the circuit diagram of FIG. 2. The addition of the varactor Va affects the low frequency of Equation 2 and the intermediate frequency of Equation 3, but does not affect the high frequency of Equation 4. That is, the varactor Va can adjust the gain characteristics of the low frequency band among the dual bands to a desired frequency by changing the low parallel resonance of Equation 2, and change the series resonance of Equation 3 together to share the differential phase difference. The frequency used as the boundary of the frequency band showing the phase difference is changed. Therefore, the phase difference at the low frequency of Equation 2 and the high frequency of Equation 4 is always kept in common with the differential phase.

FIG. 7 is a graph illustrating simulation results of transfer function values at nodes P1 and P2 in the circuit diagram of FIG. 6.

Referring to FIG. 7, the simulation was performed for each of the nodes P1 and P2 when the varactor Va is 10fF and 500fF. As shown in the graph, when the varactor Va is 10fF and 500fF, the transfer function at the node P1 and the transfer function at the node P2 at the high frequency appear to have the same value. On the other hand, the transfer function at node P1 and the transfer function at node P2 appear to have different values at intermediate frequencies, which are in series resonance with low frequencies.

8 is a diagram showing a load unit 114 of the signal conversion apparatus according to the third embodiment of the present invention.

Referring to FIG. 8, in the circuit diagram of FIG. 2, the varactor Va1 is connected in parallel to the capacitor C1, and the varactor Va2 is connected in parallel to the capacitor C2. The addition of varactors Va1 and Va2 affects both the low frequency of equation (2), the intermediate frequency of equation (3) and the high frequency of equation (4). That is, the circuit diagram of FIG. 8 can control both the gain of the low frequency band and the high frequency band of the dual band, and the frequency that is the boundary of the phase difference also changes so that the condition for removing interference is always maintained.

FIG. 9 is a graph illustrating simulation results of transfer function values at nodes P1 and P2 in the circuit diagram of FIG. 8.

Referring to FIG. 9, the simulation was performed for each of the nodes P1 and P2 when the varactor Va is 10fF and 100fF. As shown in the graph, when the varactor Va is 10fF and 100fF, the transfer function at node P1 and the transfer function at node P2 are the same value at low, intermediate and high frequencies. Appears to have

FIG. 10 shows an example of applying the circuit diagram of FIG. 2 to a low noise amplifier design having two stages.

Referring to FIG. 10, the low noise amplifier 1000 includes middle tap inductors 1020 and 1050, cascode amplifiers 1010 and 1040, and an impedance matching means 1030.

The impedance matching means 1030 includes a capacitor C3, an inductor L2 and a transistor M5.

Capacitor C3 is connected between input terminal IN and the source of transistor M5, and inductor L2 is connected between the source of transistor M5 and ground. The drain of the transistor M5 is connected to the gate of the transistor M1 of the cascode amplifier 1010, and the gate of the transistor M5 is connected to a power supply for supplying a bias voltage Vb2.

The transistor M5 is a transistor having a common gate structure, and the transistor M5 is an amplifier having a control terminal, an input terminal, and an output terminal, respectively. In FIG. 10, the transistor M5 is illustrated as an n-channel field effect transistor (FET).

The capacitor C3 blocks the DC voltage at the single phase signal VRF input to the input terminal IN, and the inductor L2 blocks the AC voltage at the single phase signal VRF.

The single phase signal VRF is input to the source of the transistor M5 and is output to the gate of the transistor M1 through the drain of the transistor M5. That is, the single phase signal VRF may be input to the gate of the transistor M1 without reflection by the transistor M5. In this case, the input impedance of the transistor M1 may be always constant regardless of the frequency band by adjusting the magnitude of the direct current flowing through the source of the transistor M5 and the transistor M5.

Meanwhile, the middle tap inductors 1020 and 1050 may be modeled similarly to the circuit diagram of FIG. 2. The middle tap inductors 1020 and 1050 include inductors Lc1 and Lc2, Lc3 and Lc4 which are symmetrically connected to each other, and parasitic capacitors existing between the middle tap and the substrate also exist symmetrically. The intermediate tap inductor and the parasitic capacitor may be modeled as an inductor L1, an inductor L2, a capacitor C1, and a capacitor C2 in the circuit diagram of FIG. 2.

In addition, capacitor C3 in the circuit diagram of FIG. 2 is basically present due to the characteristic that the intermediate tap inductor lines are physically located close to each other. Therefore, the two-stage low noise amplifier according to an embodiment of the present invention serves to lower the noise of the signal of the dual band wireless communication system simultaneously, provide gain, and completely remove interference between the bands.

The wideband single phase signal VRF output to the drain of the transistor M5 is converted into an input differential phase signal and an in-phase phase signal by the middle tap inductor 1050.

One end of each of the inductors Lc3 and Lc4 of the middle tap inductor 1050 is shared with each other through the tap and is AC grounded. That is, one end of each inductor Lc3 and Lc4 may be connected to the power supply Vdd or the ground and may be connected to a power supply for supplying a specific bias voltage.

The other end of the inductor Lc3 is connected to the input terminal N1 of the cascode amplifier 1010 and the drain of the transistor M5, and the other end of the inductor Lc4 is input terminal N1 ′ of the cascode amplifier 1040. Is connected to. Inductor Lc3 acts as a load on transistor M5 and generates resonance at two different frequencies.

In this case, the first resonance is between the inductor Lc3 of the middle tap inductor 1050 and the parasitic capacitor connected between the drain and the source of the transistor M5 and the inductors Lc3 and Lc4 of the middle tap inductor 1050. Resonance is caused by the parasitic capacitor present. This corresponds to the low frequency band of a dual band system. Low frequency output signals appear in differential phases.

The second resonance is caused by a parasitic capacitor connected between the inductor Lc3 of the middle tap inductor 1050 and the drain and the source of the transistor M5. This corresponds to the high frequency band of the dual band system, and the output signal of the high frequency band is output in a common phase.

Since the resonant frequency is determined by the various parasitic capacitors of the intermediate tap inductor 1050, both dual bands exhibit narrowband gain characteristics.

In an embodiment of the present invention, the differential phase signal and the common phase signal converted by the middle tap inductor 1050 are applied to the cascode amplifiers 1010 and 1040 of the two stages to improve the narrow band gain characteristics.

In the present invention, the second amplifier stage has been proposed to output a differential phase signal as a differential phase signal and a common phase signal as a common phase signal.

Thereafter, the low frequency band signal is still a differential phase signal through the middle tap inductor 1020, and the high frequency signal signal is still finally output as a common phase signal. However, the value of the middle tap inductor 1020 is different from the value of the middle tap inductor 1050 of the first stage, thereby implementing a slightly different resonance frequency. This allows each band of the dual band to be implemented in a wide band rather than a narrow band.

FIG. 11 is a graph illustrating gain and phase difference characteristics of the circuit diagram of FIG. 10.

Referring to FIG. 11, unlike the gain characteristics of the low frequency band, the gain characteristics of the high frequency band among the dual bands are not sensitive to the resonance point depending on the value of the inductor. Since the capacitor value multiplied by the inductor value of Equation 2, which is a low frequency, is significantly larger than the value of the capacitor multiplied by the same inductor value of Equation 4, which is a high frequency, the change of the low frequency of Equation 2 according to the change of the inductor value This is because is more sensitive than the change in the high frequency of Equation 4.

On the other hand, when looking at the phase difference between the output of the low noise amplifier, it can be seen that the phase difference of 180 degrees for a low frequency, 0 (or 360 degrees) for a high frequency.

FIG. 12 is a graph illustrating a transient simulation result of the dual band output signal of the circuit diagram of FIG. 10.

Referring to FIG. 12, it can be seen that a differential phase signal is output at a frequency of 4.5 GHz corresponding to a low frequency band, but a common phase signal is output at a frequency of 10.5 GHz corresponding to a high frequency band.

The embodiments of the present invention described above are not implemented only by the apparatus and method, but may be implemented through a program for realizing the function corresponding to the configuration of the embodiment of the present invention or a recording medium on which the program is recorded.

While the present invention has been particularly shown and described with reference to exemplary embodiments thereof, it is to be understood that the invention is not limited to the disclosed exemplary embodiments, It belongs to the scope of right.

Claims (8)

In the load circuit of a low noise amplifier,
A first capacitor, a second capacitor, a third capacitor, a first inductor, and a second inductor,
The third capacitor is connected between the first node and the second node,
One end of the first inductor and the first capacitor is connected between the first node and the third capacitor,
The second inductor and the second capacitor are connected between the second node and the third capacitor,
The other ends of the first inductor, the first capacitor, the second inductor, and the second capacitor are connected to ground,
The low noise amplifier amplifies a radio frequency signal including a signal of a first frequency band and a signal of a second frequency band,
And a load circuit for outputting the signal of the first frequency band as a differential phase signal and the signal of the second frequency band as a common phase signal. The method of claim 1,
Further includes varactors,
The varactor is connected in parallel with the third capacitor. The method of claim 1,
Further comprising a first varactor and a second varactor,
And the first varactor is connected in parallel with the first capacitor and the second varactor is connected in parallel with the second capacitor. delete delete delete delete delete

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