KR100871209B1 - Frequency converter - Google Patents

Abstract

Provided is a frequency converter capable of achieving compatibility with the balance characteristics between broadband and good IQ. In the frequency converter 100 for signal reception, a polyphase filter 130 for converting a real RF signal into a complex RF signal, a divider 140 and 150 for converting a local signal into a complex local signal, The input / output signal is a complex signal, and includes an orthogonal mixer 160 to which a complex RF signal and a complex local signal are input. It features. The frequency characteristics can be received in a wide frequency band such as 400 MHz to 6 GHz by setting the frequency characteristics in advance so as to complement each other in the RF signal and the LO signal.

Receiving Frequency Converters, Quadrature Down Converters, Complex Coefficient Transversal Filters

Description

Frequency converter {FREQUENCY CONVERTER}

1 is a block diagram of a receiving frequency converter according to a first embodiment of the present invention;

2 is a block diagram illustrating a polyphase filter in the embodiment of FIG. 1;

3 is an exemplary diagram for explaining correspondence between switching of a switch and a division ratio of a local signal according to a first embodiment of the present invention;

4 is a configuration diagram of a frequency converter for transmission according to a second embodiment of the present invention;

5 is a configuration diagram of a receiving frequency converter according to a third embodiment of the present invention;

6 is a block diagram of a complex coefficient transversal filter according to a third embodiment of the present invention;

7 is an exemplary diagram for explaining correspondence between switching of a switch, a complex coefficient filter of RF, and a division ratio of a local signal according to a third embodiment of the present invention;

8 is a configuration diagram of a frequency converter for transmission according to a fourth embodiment of the present invention;

9 is a block diagram showing the configuration of a 90 ° distributor according to the prior art,

10 is a block diagram showing the configuration of a quadrature down converter according to the prior art;

[Description of the code]

100 frequency converter

110 Bandpass Filter (BPF)

120 Low Noise Amplifier (LNA)

130 Polyphase Filters (PPF)

140 1/2 dispenser

150 1/4 divider

160 double quadrature mixer

161-164 multiplier

165 subtractor

166 adder

170, 180 Low Pass Filter (LPF)

190 Complex Factor Transversal Filter

198 Low Noise Amplifier

200 frequency converter

210 Bandpass Filter (BPF)

220 amplifier circuit

230 Polyphase Filters (PPF)

240 1/2 dispensers

250 1/4 divider

260 double quadrature mixer

261-264 multiplier

265 subtractor

266 adder

270, 280 Lowpass Filter (LPF)

290 Complex Coefficient Transversal Filter

298 amplifier circuit

300 frequency converter

400 frequency converter

S1 to S5 switch

RF RF Signal

LO Local Signal (Local Signal Oscillator)

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a frequency converter, and in particular, a wideband frequency converter that is compatible with the broadband characteristics of the downconverter / upconverter wideband technology and good imbalance between in-phase and quadrature (I / Q). It is about.

As mobile communication develops, terminals capable of communicating in a plurality of bands are required, and in particular, portable terminals are required to correspond to 400 MHz to 6 GHz. Here, an orthogonal detector (orthogonal demodulator) for inputting a real signal to obtain a complex signal, or an orthogonal modulator for inputting a complex signal to obtain a real signal is a general configuration, and is realized by injecting a complex signal into a local signal with two multipliers.

Such an orthogonal modulator and an orthogonal demodulator are used to determine the difference between the IQs by the difference between the phase and amplitude of the multiplication period, or the phase and amplitude difference between the real part I and the real Q (even symmetric signal: cos and odd symmetric signal: sin) of the complex local signal. There is an imbalance. This results in signal distortion in the homo IF and image signals in the heterodyne. Various attempts have been made to improve such a problem of signal distortion and image signal generation. In order to realize this in a wide band, for example, 400 MHz to 6 GHz as a single circuit, not only the wider circuit operation but also I / Q The difficulty for suppressing liver imbalance within a predetermined range is further increased.

As the above improvement method, for example, "SiGe-MMIC quadrature modulator for 0.8-5.2 GHz multiband / multimode direct conversion transmitter"村上 圭 司, Technical Report of the Institute of Electronics and Information Sciences SR 2005-36, pp.105-108, 2005). A description with reference to FIG. 9 is as follows. The LO (Local Ocillator) signal of the quadrature modulator is divided into a 1/2 divider (90 ° divider) (10), an emitter follower circuit (20), and a poly phase filter (PPF) ( When input to the circuit of the serial connection of 30, the phase error occurs because the balance of the output signal of the 1/2 divider 10 is broken in the polyphase filter 30, so that the phase error can be reduced by increasing the output frequency. As such, when the polyphase filter 30 is present, orthogonal precision, which is an imbalance characteristic between I / Q of the LO output, may be improved in a wide frequency band compared to the case where the polyphase filter 30 is not present.

As another improvement, "A single-Chip 900MHz CMOS Receiver Front-End with a High Performance Low-IF Topology" (Jan Crols, Michel sJ Steyaert, IEEE J. of solid-State Circuits, vo1.30, no.12, pp. 1483-1492, Dec. 1995). A description with reference to FIG. 10 is as follows. The RF signal and the LO signal are input to a double quadrature mixer (DQM) 60 through quadrature converters 40 and 50 each consisting of a complex coefficient filter. The double quadrature mixer 60 is composed of four multipliers 61 to 64, a subtractor 65, and an adder 66 to which the combination of the real part I and the virtual part Q from the quadrature converters 40 and 50 is input, respectively.

When the balance between I / Q of the mixer output of the double quadrature mixer 60 is expressed by the image reject rate (IMRR), it is assumed that there is no error in the mixer output. Can be calculated as

IMRR = IMRR_RF + IMRR_local

Here, 'IMRR_RF' is an IMRR of an RF (Radio Frequncy) input, and 'IMRR_local' is an IMRR of a local input. From this, if the IMRR of either RF or local is good, the IMMR of the I / Q balance can be good. Hereinafter, the effect by such a double orthogonal mixer 60 is called a complementary effect.

However, in the configuration as shown in FIG. 9, since the 1/2 divider 10, which is a frequency divider 10, and the polyphase filter 30 are connected in series, the LO output is phase stabilized in the frequency divider and the polyphase. The good state of IMRR in the filter 30 is applied. Since this LO output signal passes through the polyphase filter 30, it can only correspond to a 90 degree phase difference. For this reason, an even harmonic mixer that outputs a mixed wave with twice the LO output signal cannot be used. Also, IMRR may not be good in relation to the RF signal.

In the configuration as shown in Fig. 10, since the frequency characteristics of the RF signal and the LO signal coincide, the balance characteristics between I / Q are good, but the broadband characteristics are not disclosed at all. Cannot have good characteristics for broadband. For example, it is difficult to improve the balance characteristics between I / Q in a wide band such as 400 MHz to 6 GHz.

It is therefore an object of the present invention to provide a frequency converter capable of achieving both compatibility with broadband and good imbalance between I / Q.

In order to solve the above problems, the reception frequency converter 100 according to the first embodiment of the present invention includes a polyphase filter 130 which is an RF signal converter for converting a real RF signal into a complex RF signal, and a local signal. The 1/2 divider 140, the 1/4 divider 150, and the I / O signal, which are local signal converters for converting the signal into a complex local signal, are complex signals, and the complex RF signal and the complex local signal are inputted. It has a double quadrature mixer 160. The frequency band showing the best characteristic of the RF signal converter is different from the frequency band showing the best characteristic of the RF signal converter. In addition, in the above, the code | symbol shown in parentheses in the component is shown for the convenience of description only the corresponding component in the below-mentioned Example and drawing as an example, and this invention is not limited to this. . The same applies to the following.

According to such a configuration, the frequency characteristics of the RF signal conversion section including the complex coefficient filter and the like are set in advance so as to complement frequency bands having poor characteristics in the RF signal and the LO signal. For example, even in the case of designing a polyphase filter, if the polyphase filter 130 of a plurality of stages is inserted in the RF side, the loss due to the increase of insertion loss increases, so that the cover The frequency band which cannot be corrected is compensated on the local side. As a result, when the RF signal and the LO signal are respectively input to the double quadrature mixer 160 through a complex coefficient filter, the RF signal and the LO signal can be received in a wide frequency band such as 400 MHz to 6 GHz.

In the frequency converter of the present invention, the RF signal converter may be configured to include a polyphase filter (130). Alternatively, the RF signal converter includes a polyphase filter 130, a complex coefficient transversal filter 190, and a switch for switching the polyphase filter 130 and the complex coefficient transverse filter 190. It is also possible to. The polyphase filter 130 is excellent as a wideband phase divider, but since it is a filter composed of R and C, the loss increases at a high frequency and the precision also decreases. In this regard, according to the complex coefficient transversal filter 190, the present invention can be applied even at frequencies requiring high sensitivity and high IMRR.

The local signal converter may be configured to include a first divider for high frequency and a second divider for low frequency. In the first embodiment of the present invention, the first divider is described as a half divider 240, and the second divider is described as a quarter divider 250. By switching the division ratio for the same oscillator, it is possible to increase the frequency range input to the mixer while suppressing the frequency variable range of the oscillator. For example, if the divider is composed of dividers such as 1/2, 1/4, 1/8, the division of each divider is in octave units. When the first divider is 1/2 and the second divider is 1/4, when the low frequency, which is the output frequency of the first divider, is 400 MHz to 800 MHz, the output frequency of the second divider is 800 MHz to 1.6 GHz, Each input frequency and local oscillator frequency ranges from 1.6 GHz to 3.2 GHz.

It is also possible to use an even harmonic mixer for outputting a mixed wave with a double wave of the local signal in the mixer constituting the double quadrature mixer. By setting the phase difference of the local signal to 45 ° and multiplying it inside the mixer, it becomes a local signal of 90 ° phase difference.

In addition, in order to solve the above problems, the transmission frequency converter 200 according to the second embodiment of the present invention is a 1/2 divider, 1/4 which is a local signal conversion unit for converting a local signal into a complex local signal. The divider 240, 250, the input / output signal is a complex signal, the complex local signal is input, and a double quadrature mixer 260 which outputs a complex RF signal, and converts the complex RF signal into a real RF signal. A polyphase filter 230 that is an RF signal converter is provided. The frequency band showing the best characteristic of the RF signal converter is different from the frequency band showing the best characteristic of the local signal converter.

According to such a configuration, the frequency characteristics of the RF signal conversion section including the complex coefficient filter and the like are set in advance so as to complement frequency bands having poor characteristics in the RF signal and the LO signal. For example, even in the case of designing a polyphase filter, if the polyphase filter 230 of a plurality of stages is inserted in the RF side, the loss due to the increase of insertion loss increases, so that the frequency band which cannot be covered cannot be localized. Complement in As a result, when the RF signal and the LO signal are respectively input to the double quadrature mixer 260 through the complex coefficient filter, the RF signal and the LO signal can be received in a wide frequency band such as 400 MHz to 6 GHz.

In the transmission frequency converter of the present invention, the RF signal converter may adopt a configuration including a polyphase filter. Alternatively, the RF signal converter includes a polyphase filter 230, a complex coefficient transversal filter 290, and a switch for switching the polyphase filter 230 and the complex coefficient transverse filter 290. It is also possible to make a configuration. The polyphase filter 230 is excellent as a wideband phase divider. However, since the polyphase filter 230 is a filter composed of R and C, the loss is increased at high frequencies and the precision is also reduced. In this respect, according to the complex coefficient transversal filter, the present invention can be applied to frequencies requiring high sensitivity and high IMRR.

The local signal converter may be configured to include a first divider for high frequency and a second divider for low frequency. In the second embodiment of the present invention, the first divider is described as a half divider 240, and the second divider is described as a quarter divider 250. By switching the division ratio for the same oscillator, the frequency range input to the mixer can be expanded while suppressing the frequency variable range of the oscillator. For example, if the divider is composed of dividers such as 1/2, 1/4, 1/8, the division of each divider is in octave units. When the first divider is 1/2 and the second divider is 1/4, when the low frequency, which is the output frequency of the first divider, is 400 MHz to 800 MHz, the output frequency of the second divider is 800 MHz to 1.6 GHz, Each input frequency has a local oscillator frequency of 1.6GHz to 3.2GHz.

It is also possible to use an even harmonic mixer that outputs a mixed wave with the double wave of the local signal in the mixer constituting the double quadrature mixer 260. The phase difference of the local signal is set to 45 ° and doubled inside the mixer, resulting in a local signal having a 90 ° phase difference.

Hereinafter, with reference to the accompanying drawings will be described in detail a preferred embodiment of the frequency converter according to the present invention. In addition, in this specification and drawing, duplication description is abbreviate | omitted by using the same code | symbol about the component which has a substantially same functional structure.

First, the basic concept of each embodiment described below will be described.

In the following embodiment, it is possible to provide a frequency converter having wideband and high performance by using a double quadrature mixer having good I / Q balance. As described later with reference to Fig. 1, the double quadrature mixer uses a complex signal as all ports of the mixer. If there is no error in the mixer output at this time, the imbalance between the IQs of the mixer outputs may be expressed as shown in Equation 2 below.

IMRR = IMRR_RF + IMRR_local

Where IMRR_RF is the IMRR of the RF input and IMRR_local is the IMRR of the local input.

From such an IMRR, it can be seen that either IMRF of RF or local should be good in order to improve IMRR of IQ balance. Thus, a signal input to a double quadrature mixer is generated as follows.

First, a broadband complex coefficient filter such as a polyphase filter is used as a means for obtaining a complex RF signal from a real RF signal or a means for obtaining a real RF signal from a complex RF signal. Here, the frequency at which a good 90 ° phase difference can be obtained by the polyphase filter is set to a high frequency.

On the other hand, as a means for obtaining a complex local signal from a local signal or a means for obtaining a local signal from a complex local signal, when a phase difference of 90 degrees or 45 degrees is obtained by dividing the output of the oscillator, 1 / Use a two-division output, and use a quarter-division output at low frequencies. This improves the phase error at low frequencies even if the phase error increases at high frequencies.

Then, the frequency characteristics of the complex coefficient filter are set in advance so as to deviate from each other in the RF signal and the LO signal. In other words, it is assumed that the means for converting the RF signal exhibits the best characteristic and the frequency band for converting the local signal has the best characteristic. As a result, when the RF signal and the LO signal are respectively input to the double quadrature mixer through a complex coefficient filter, the RF signal and the LO signal can be received in a wide frequency band such as 400 MHz to 6 GHz.

In the above, the basic concept of the Example of this invention was demonstrated.

Hereinafter, the specific structure for realizing this basic concept is demonstrated as 1st-4th embodiment.

(First embodiment: reception frequency converter)

1 shows a signal obtained by converting a real RF signal of 400 GHz to 2.1 GHz and 2.5 GHz into a complex RF signal by a complex coefficient filter, and a local signal of the same frequency as a desired RF signal as a complex local signal. The converted signal is input to a double quadrature mixer (full complex mixer) composed of four multipliers, adders and subtractors, and is an embodiment of a frequency converter for obtaining a complex baseband signal (complex zero-IF signal).

As shown in Fig. 1, the receiving frequency converter 100 according to the present embodiment includes a band pass filter (BPF) 110 and a low noise amplifier circuit (LNA) for the purpose of band limitation on the RF signal side. 120 and a polyphase filter (PPF) 130. In addition, the local signal oscillator side LO is provided with a 1/2 divider 140, a 1/4 divider 150, and switches S1 and S2 for switching these two dividers. The complex RF signal from the RF signal side and the complex local signal from the local signal oscillator side are input to the subsequent double quadrature mixer 160. The complex baseband signal (complex zero-IF signal) from the double quadrature mixer 160 is output as a real signal I through a low pass filter (LPF) 170, and the low pass filter (LPF) 180 is output. Is output as a complex signal Q.

Below, each component of FIG. 1 is demonstrated in detail.

(RF signal side)

On the RF signal side, a band pass filter (BPF) 110, a low noise amplifier circuit (LNA) 120, and a polyphase filter (PPF) 130 are provided for band limitation. The bandpass filter 110 is a filter bank for switching the bandpass filter in accordance with the frequency. The low noise amplifier (LNA) 120 is an amplifier having a small noise figure applied to the initial stage of the receiver.

The polyphase filter 130 is a kind of wideband complex coefficient filter, and realizes a phase difference of 90 ° over a wideband. 2 is an explanatory diagram showing an example of the polyphase filter 130. As shown in Fig. 2, the polyphase filter 130 has a bridge structure having an input of four terminals and an output of four terminals, and the reception has four terminals of I +, I-, Q +, and Q-. At that time, input to I + and I- (Q +, Q- is ground), and get Q from I, Q + and Q- from I + and I-. The transmission inputs I to I + and I-, inputs Q to Q + and Q-, and obtains a real signal by obtaining I-jQ from I of outputs I + and I- and of -Q of Q- and Q +. Get

Here, it is important that there is no large variation in the amplitude characteristic also outside the frequency range where a phase difference of 90 degrees can be obtained. For example, even if the phase difference is 0 °, if the amplitude level does not change, only the same operation as the conventional frequency converter is performed, but when the level is lowered outside the predetermined frequency range such as a bandpass filter, the desired sensitivity cannot be obtained. Because. In addition, in the broadband complex coefficient filter other than the polyphase filter 130, a phase difference of 90 degrees is realized in a wide band by the phase difference of two phase periods using an all-pass filter by RC or LC. There is a means.

In this embodiment, the frequency characteristics of the polyphase filter 130, which is a complex coefficient filter, are set in advance so as to deviate from the RF signal and the LO signal. As a result, when the RF signal and the LO signal are respectively input to the double quadrature mixer 160 through the polyphase filter 130, reception is performed in a wide frequency band such as 400 MHz to 6 GHz due to the complementary effect of the double quadrature mixer 160. It is possible.

(Local signal side)

On the LO side of the local signal oscillator, a half divider 140, a quarter divider 150, and switches S1 and S2 for switching these two dividers 140 and 150 are provided. The switches S1 and S2 switch the frequency division ratios of the frequency dividers 140 and 150 according to the RF frequency. At high frequencies, divide by 1/2; at low frequencies, divide by 1/4.

The 1/2 divider 140 obtains an output having a phase difference of 90 ° by using a positive edge and a negative edge of the input signal, but the duty of the input signal is completely 50%. Otherwise, output with full 90 ° phase difference cannot be obtained. Since it is difficult to realize a full duty of 50%, the presence of phase error is inevitable at the output of the 1/2 divider 140.

On the other hand, in the quarter frequency divider 150, by using only one of the positive edge and the negative edge, an output having a phase difference of 90 ° can be obtained, so that a good complex signal output can be obtained. In general, by changing the frequency division ratio for the same oscillator, it is possible to expand the frequency range input to the mixer while suppressing the frequency variable range of the oscillator.

The selection of the dividers 140 and 150 by switching the switches S1 and S2 is summarized as shown in FIG. As shown in Fig. 3, when the RF frequency is 400 MHz to 1 GHz (low frequency), the quarter frequency divider 150 is selected. When the RF frequency is 1 GHz to 2.1 GHz and 2.5 GHz (high frequency), 1/2 is selected. Select the divider 140. By designing the pole of the polyphase filter 130 to 1 GHz or more, the IMRR of the polyphase filter 130 and the IMRR of the dividers 140 and 150 are set to complement each other, so that a good IMRR can be obtained in a wide band.

(Double quadrature mixer)

The double quadrature mixer 160 is comprised by four multipliers 161-164, the subtractor 165, and the adder 166, as shown in FIG.

The multiplier 161 receives the output of the polyphase filter 130 and the output of the switch S1 as inputs, and outputs a multiplication result to the subtractor 165. The multiplier 162 receives the output of the polyphase filter 130 and the output of the switch S2 as inputs, and outputs a multiplication result to the subtractor 165. The multiplier 163 takes the output of the polyphase filter 130 and the output of the switch S1 as inputs, and outputs the multiplication result to the adder 166. The multiplier 164 receives the output of the polyphase filter 130 and the output of the switch S2 as inputs, and outputs a multiplication result to the adder 166.

In order to cope with an input signal having a 45 ° phase difference, the multipliers 161 to 164 may use an even harmonic mixer for outputting a mixed wave with a double wave of the input signal. The multiplier is multiplied inside the multiplier with respect to the input signal having a 45 ° phase difference, resulting in a 90 ° phase difference signal.

The subtractor 165 inputs the output of the multiplier 161 and the output of the multiplier 162, and outputs the subtraction result to the low pass filter (LPF) 170 at the rear stage. The output of the low pass filter (LPF) 170 is the real I. The adder 166 takes the output of the multiplier 163 and the output of the multiplier 164 as inputs, and outputs the addition result to the low pass filter (LPF) 180 at the next stage. The output of the low pass filter (LPF) 180 becomes the hub Q.

(Effect of the first embodiment)

As described above, according to the present embodiment, the frequency characteristics of the polyphase filter 130, which is a complex coefficient filter, are set in advance so as to deviate from the RF signal and the LO signal. As a result, when the RF signal and the LO signal are respectively input to the double quadrature mixer 160 through a complex coefficient filter, the RF signal and the LO signal can be received in a wide frequency band such as 400 MHz to 6 GHz due to the complementary effect of the double quadrature mixer 160. have.

(Second embodiment: transmission frequency converter)

Fig. 4 shows a double orthogonal mixer composed of four multipliers, an adder and a subtractor for converting a complex baseband signal and a signal obtained by converting a local signal of the same frequency as a desired RF signal into a complex local signal by a divider. After inputting a complex RF signal to a complex RF signal and suppressing a negative or positive frequency with a complex coefficient filter, the real part of the complex RF signal is output and a real RF signal of 2.1 GHz and 2.5 kHz is applied from 400 MHz. An example of a frequency converter obtained.

As shown in Fig. 4, the transmission frequency converter 200 according to the present embodiment includes a band pass filter (BPF) 210, an amplifier circuit 220, for the purpose of band limitation on the RF signal side, A polyphase filter (PPF) 230. Further, on the LO side of the local signal oscillator, a half divider 240, a quarter divider 250, and switches S1 and S2 for switching these two dividers are provided. The real signal I is input to the double quadrature mixer 260 through the low pass filter (LPF) 270 and the complex signal Q through the low pass filter (LPF) 280, respectively. Further, the complex local signal at the local signal oscillator LO side is input to the double quadrature mixer 260 at the rear stage. The output signal of the double quadrature mixer 260 is an RF signal through the polyphase filter 230, the amplifying circuit 220, and the band pass filter (BPF) 210, and becomes a transmission signal.

Below, each component is demonstrated in detail.

(RF signal side)

On the RF signal side, a band pass filter (BPF) 210, an amplifier circuit 220, and a polyphase filter (PPF) 230 are provided for band limitation. The bandpass filter 210 is a filter bank that switches the bandpass filter in accordance with the frequency.

The polyphase filter 230 is a kind of wideband complex coefficient filter, and realizes a 90 ° phase difference in wideband. Here, it is important that there is no large variation in the amplitude characteristic even outside the frequency range where a phase difference of 90 degrees can be obtained. For example, even if the phase difference is 0 °, if the amplitude level does not change, only the same operation as that of the conventional frequency converter is performed. However, if the level decreases outside a predetermined frequency range such as a bandpass filter, the desired sensitivity can be obtained. Because there is no. In addition, in the broadband complex coefficient filter other than the polyphase filter 230, there is a means for realizing a phase difference of 90 degrees over a wide band by the phase difference between two phase periods using the all-pass filter by RC or LC.

(Local signal side)

On the LO side of the local signal oscillator, a half divider 240, a quarter divider 250, and switches S1 and S2 for switching these two dividers 240 and 250 are provided. The switches S1 and S2 switch the division ratios of the dividers 240 and 250 according to the RF frequency. At high frequencies, divide by 1/2; at low frequencies, divide by 1/4.

The 1/2 divider 240 obtains an output having a phase difference of 90 ° by using the positive and negative edges of the input signal, but obtains an output having a full 90 ° phase difference if the duty of the input signal is not 50% completely. Can not. Since it is difficult to realize 50% of full duty, the presence of phase error is inevitable at the output of the 1/2 divider 240.

On the other hand, in the quarter frequency divider 250, an output having a 90 ° phase difference can be obtained by using either the positive edge or the negative edge, so that a good complex signal output can be obtained. In general, by switching the frequency division ratio for the same oscillator, it is possible to expand the frequency range input to the mixer while suppressing the frequency variable range of the oscillator.

The selection of the dividers 240 and 250 by switching the switches S1 and S2 is summarized in the table shown in FIG. As shown in Fig. 3, the quarter frequency divider 250 is selected when the RF frequency is 400 MHz to 1 GHz (low frequency), and 1/2 when the RF frequency is 1 GHz to 2.1 GHz and 2.5 GHz (high frequency). Select divider 240. By designing the pole of the polyphase filter 130 to 1 GHz or more, the IMRR of the polyphase filter 230 and the IMRR of the dividers 240 and 250 are set to complement each other, and a good IMRR can be obtained in a wide band.

(Double quadrature mixer)

As shown in FIG. 4, the double quadrature mixer 260 is constituted by four multipliers 261 to 264, a subtractor 265, and an adder 266.

The multiplier 261 receives the output of the low pass filter 270 and the output of the switch S1 as inputs, and outputs a multiplication result to the subtractor 265. The multiplier 262 receives the output of the low pass filter 280 and the output of the switch S2 as inputs, and outputs the multiplication result to the subtractor 265. The multiplier 263 receives the output of the low pass filter 280 and the output of the switch S1 as inputs, and outputs the multiplication result to the adder 266. The multiplier 264 receives the output of the low pass filter 270 and the output of the switch S2 as inputs, and outputs a multiplication result to the adder 266.

The subtractor 265 takes the output of the multiplier 261 and the output of the multiplier 262 as inputs, and outputs the subtraction result to the polyphase filter 230 at the rear stage. The adder 266 takes the output of the multiplier 263 and the output of the multiplier 264 as inputs, and outputs the addition result to the polyphase filter 230 at the rear stage. The output of the polyphase filter 230 becomes a transmission signal as an RF signal through the amplifier circuit 220 and the band pass filter (BPF) 210.

(Effect of 2nd Example)

As described above, according to the present embodiment, the frequency characteristics of the polyphase filter 130, which is a complex coefficient filter, are set in advance so as to deviate from the RF signal and the LO signal. As a result, when the RF signal and the LO signal are respectively input to the double quadrature mixer 160 through the complex coefficient filter, the RF signal and the LO signal can be received in a wide frequency band such as 400 MHz to 6 GHz due to the complementary effect of the double quadrature mixer 160. have.

(Third Embodiment)

As a third embodiment of the present invention, an application example of the first embodiment (receiving frequency converter) will be described.

In the first embodiment, a configuration in which the polyphase filter 130 is provided on the RF signal side as shown in FIG. Since a wideband complex coefficient filter such as a polyphase filter does not have a band limit function, a band pass filter 110 for the purpose of a separate band limit is required, and a total including the loss of the polyphase filter 130 is required. The loss can be large. In addition, since the polyphase filter can be configured on an IC, it is easy to make one-chip, but a high IMRR may not be obtained due to IC manufacturing error and parasitic resistance, capacitor, and inductor.

Thus, in the receiving frequency converter 300 according to another embodiment of the present invention, as shown in FIG. 5, the complex coefficient transversal filter 190 and the low noise amplification, which are complex coefficient filters in parallel with the polyphase filter 130, are shown. The structure provided with the circuit 198 is employ | adopted.

6 is an explanatory diagram illustrating an example of the complex coefficient transverse filter 190. As shown in FIG. 6, the complex-coefficient transverse filter 190 has an inter-digital transducer having different cross-widths installed on the piezoelectric plate 191 and the piezoelectric plate 191 for each location. IDT ”) 192 to 195. Each comb-shaped part is also called electrode finger. IDTs 192 and 194 are connected to input terminals, and when an impulse electric signal is applied, mechanical distortion occurs due to piezoelectricity, surface acoustic wave (SAW) is excited, and piezoelectric plate 191 Will propagate in the left and right directions. The IDT 193 is connected to an output terminal I that outputs the component of the real axis, and is provided at a position capable of receiving surface acoustic waves from the IDT 192. In addition, the IDT 195 is connected to an output terminal Q that outputs the component of the imaginary axis, and is provided at a position capable of receiving surface acoustic waves from the IDT 194.

Then, one or both frequency characteristics of the complex coefficient filters 130 and 190 are set in advance so as to deviate from each other in the RF signal and the LO signal. In addition, while the switch (S5) is provided between the RF signal and the complex coefficient filter (polyphase filter 130 or complex coefficient transverse filter 190), and between the complex coefficient filter and the double quadrature mixer 160 Install switches S3 and S4 on the In addition, about the other component, since it is substantially the same as the component shown in FIG. 1, redundant description is abbreviate | omitted by using the same code | symbol.

In the receiving frequency converter 300, at a frequency requiring high sensitivity and high IMRR, the switching coefficients S3, S4, and S5 are switched to replace the complex coefficient transversal filter 190 instead of the polyphase filter 130. It is possible to select).

The selection of the complex coefficient filters 130 and 190 and the divider 140 and 150 by switching the switches S1 to S5 is summarized as shown in the table shown in FIG. As shown in Fig. 7, when the RF frequency is 400 MHz to 1 GHz, the polyphase filter 130 is selected on the RF signal side, and the quarter divider 150 is selected on the local signal side. When the RF frequency is 1 GHz to 2.1 GHz, the polyphase filter 130 is selected on the RF signal side, and the 1/2 divider 140 is selected on the local signal side. When the RF frequency is 2.5 GHz, the complex coefficient transversal filter 190 is selected on the RF signal side, and the 1/2 divider 140 is selected on the local signal side. By designing the pole of the polyphase filter to 1 GHz or more, the IMRR of the polyphase filter and the IMRR of the frequency divider are set to complement each other, and a good IMRR can be obtained in a wide band.

(Effect of 3rd Example)

As described above, according to the third embodiment of the present invention, in addition to the effects of the first embodiment described above, the filter passes through the filter by switching to a complex coefficient transversal 190 that does not require the bandpass filter. Resulting losses can be reduced. In addition, by switching to the complex coefficient transversal 190, the present invention can be applied to a frequency requiring high sensitivity and high IMRR.

(Example 4)

As a fourth embodiment of the present invention, an application example of the second embodiment (receiving frequency converter) will be described.

Similar to the third embodiment, in the transmission frequency converter 400 according to the fourth embodiment of the present invention, as shown in FIG. 8, a complex coefficient that is another complex coefficient filter in parallel with the polyphase filter 230. The structure provided with the transversal filter 290 (and the amplifying circuit 298) is employ | adopted. The frequency characteristics of either or both of the complex coefficient filters 230, 290 are set in advance so as to deviate from each other in the RF signal and the LO signal. Furthermore, the switch S5 is provided between the RF signal and the complex coefficient filter (the polyphase filter 230 or the complex coefficient transverse filter 290), and the complex coefficient filter and the double quadrature mixer 260 Install switches S3 and S4 between them. In addition, about the other component, since it is substantially the same as the component shown in FIG. 4, the duplicate description is abbreviate | omitted by using the same code | symbol.

In the frequency converter 400 for transmission, at a frequency requiring high sensitivity and high IMRR, by switching the switches S3, S4, and S5, the complex coefficient transversal filter 290 instead of the polyphase filter 130 is performed. It is possible to choose.

The selection of the complex coefficient filters 230 and 290 and the dividers 240 and 250 by switching the switches S1 to S5 is summarized in the table shown in FIG. As shown in Fig. 7, when the RF frequency is 400 MHz to 1 GHz, the polyphase filter 230 is selected on the RF signal side, and the quarter divider 250 is selected on the local signal side. When the RF frequency is 1 GHz to 2.1 GHz, the polyphase filter 230 is selected on the RF signal side, and the 1/2 divider 240 is selected on the local signal side. When the RF frequency is 2.5 GHz, the complex coefficient transversal filter 290 is selected on the RF signal side, and the 1/2 divider 240 is selected on the local signal side. By designing the pole of the polyphase filter to 1 GHz or more, the IMRR of the polyphase filter and the IMRR of the frequency divider are set to complement each other, and a good IMRR can be obtained in a wide band.

(Effect of 4th Example)

As described above, according to the fourth embodiment of the present invention, in addition to the effects of the second embodiment, the filter is interposed by switching to a complex coefficient transversal 290 that does not require a bandpass filter. The loss by doing can be reduced. In addition, by switching to the complex coefficient transversal 290, the present invention can be applied to a frequency requiring high sensitivity and high IMRR.

As mentioned above, although preferred embodiment of the frequency converter which concerns on this invention was described referring an accompanying drawing, this invention is not limited to this example. It will be apparent to those skilled in the art that various modifications or modifications can be predicted within the scope of the technical idea described in the claims, and those of course belong to the technical scope of the present invention.

For example, according to the present invention, since the imbalance characteristics between IQ better than those of the wideband frequency converter can be realized in the configuration as shown in FIG. 9, not only homodynes but also low IF frequency converters requiring high IMRR have suitable performance. The analog unit can also be applied to a radio such as a digital radio or a software radio in which frequency conversion is performed in sparse frequency steps, and digital conversion is performed in a compact frequency step.

In the above embodiment, the configuration in which the polyphase filter is used on the RF signal side and the divider is provided on the local signal side has been described. However, the present invention is not limited to this, but the RF signal side and the local signal side are described. The polyphase filter may be used in both directions.

Specifically, for example, assuming that there are no deviations between conventional inductors, capacitances, resistors, and capacitors, 3 poles having a pole at 500 MHz, 1.2 GHz, and 2.8 GHz can obtain an IMRR of 30 dB from 400 MHz to 3.5 GHz. By using a polyphase filter at the stage of the local signal side and a polyphase filter of one stage having a pole at 5.3 GHz for the phase of the RF signal side, an IMRR of 35 dB or more can be obtained from 2.1 MHz at 400 MHz. . In addition, an IMRR of 40 dB or more can be obtained at 6 GHz from 2.4 which can use OFDM or QAM. The element values of the local polyphase filter at this time are R: 200Ω, C1: 1.591pF, C2: 0.663pF, C3: 0.274pF, and the element values of the RF polyphase filter are R: 200Ω, C: 0.15pF. to be.

The present invention can be used for a frequency converter, and in particular, it can be used for a wideband frequency converter that is compatible with the balance characteristics between the wideband and good IQ in the widening technology of the down converter / up converter.

As mentioned above, according to this invention, it is possible to provide the frequency converter (broadband frequency converter) which is compatible with the balance characteristic between broadband and favorable IQ.

In other words, the local error is increased in phase error at high frequencies and the RF signal is increased in phase error at low frequencies, but the increase in phase error is suppressed because the IMRR at the output of the double quadrature mixer is the sum of both. Either high or low frequencies can achieve good IMRR.

In addition, in the configuration as shown in Fig. 9 of the prior art, the polyphase filter can only correspond to the 90 ° phase difference in order to output a signal having a 90 ° phase difference, but in the present invention, it is possible to cope with the phase difference of 45 ° by using an even harmonic mixer. Therefore, there is an effect that even harmonics excellent in distortion and DC offset in the down converter can be used.

In addition, in the double quadrature mixer, the critical demand for the cause of imbalance between the mixer and the local signal is alleviated. Therefore, by increasing the size of the transistor used in the mixer, it is necessary to improve the balance between the IQs. Similar countermeasures such as obstruction of widening of circuit operation are also unnecessary.

In addition, since the imbalance characteristics between the IQs that are better than those of the conventional broadband frequency converters as shown in FIG. 9 can be realized, not only homodynes but also low-frequency frequency converters requiring high IMRR have appropriate performance. It is also possible to apply to a radio such as a digital radio or a software radio in which the frequency conversion in the step is performed and the frequency conversion in the compact step is performed in the digital unit.

Claims (10)

In the frequency converter, An RF signal converter for converting a real RF signal into a complex RF signal; A local signal converter for converting the local signal into a complex local signal; The input / output signal is a complex signal, and has a double quadrature mixer to which the complex RF signal and the complex local signal are input. And a frequency characteristic of the RF signal converter and the local signal converter set to different frequencies. The method of claim 1, The RF signal converter is a frequency converter characterized in that it comprises a polyphase filter. The method of claim 1, The RF signal converter includes a polyphase filter, a complex coefficient transverse filter, and a switch for switching the polyphase filter and the complex coefficient transverse filter. The method according to any one of claims 1 to 3, The local signal converter includes a first divider for high frequency and a second divider for low frequency. The method according to any one of claims 1 to 3, And the double quadrature mixer comprises a plurality of mixers, and uses an even harmonic mixer for outputting a mixed wave of two times the local signal to the mixer. In the frequency converter, A local signal converter for converting the local signal into a complex local signal; A double quadrature mixer, in which an input / output signal is a complex signal and the complex local signal is input to output a complex RF signal; An RF signal conversion unit converting the complex RF signal into a real RF signal; And a frequency characteristic of the RF signal converter and the local signal converter set to different frequencies. The method of claim 6, The RF signal converter is a frequency converter, characterized in that it comprises a polyphase filter. The method of claim 6, The RF signal converter includes a polyphase filter, a complex coefficient transverse filter, and a switch for switching the polyphase filter and the complex coefficient transverse filter. The method of claim 6, wherein The local signal converter includes a first divider for high frequency and a second divider for low frequency. The method of claim 6, wherein And an even harmonic mixer for outputting a mixed wave of twice the wavelength of the local signal as a mixer constituting the double quadrature mixer.

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