KR100513255B1 - Control device for an induction motor - Google Patents

Abstract

The present invention relates to a control device of an induction motor capable of controlling the generated torque of the induction motor with high precision. The present invention relates to a rotational speed detector (2) for detecting the rotational speed of the induction motor (1) and one of the induction motor (1). From the current detector 3 for detecting the difference current, the amplifying means 6a for amplifying the deviation between the estimated primary current obtained from the magnetic flux observer and the primary current obtained from the current detector 3, and the rotational speed detector 2 A magnetic flux observer 4a for estimating the estimated secondary magnetic flux and the estimated primary current of the induction motor based on the obtained rotational speed, the primary voltage of the induction motor 1 and the deviation signal obtained from the amplifying means 6a, and the magnetic flux observer Control means (5) for controlling the voltage applied to the induction motor based on the estimated secondary magnetic flux obtained from the amplification means, and the amplifying means (6a) is based on a feedback gain composed of eight independent elements, respectively. Amplitude deviation is amplified, and high torque control is possible, especially at low speeds.

Description

CONTROL DEVICE FOR AN INDUCTION MOTOR}

The present invention relates to a control device of an induction motor capable of controlling the generated torque of the induction motor with high accuracy.

Background Art Conventionally, as a method of driving an induction motor with high performance, a vector control method is known in which the d-axis current and the q-axis current of the rotational coordinate axis (d-q axis) rotating in synchronization with the secondary magnetic flux are respectively controlled to desired values. In addition, it is often impossible to directly observe the secondary magnetic flux due to hardware limitations. Therefore, a sliding frequency type vector control method for calculating the estimated secondary magnetic flux based on the primary current of the induction motor has been proposed.

However, this sliding frequency type vector control method requires a secondary resistor for the calculation of the estimated secondary magnetic flux, so that there is a problem that the control performance deteriorates when the secondary resistance changes due to heat generation or the like.

12 is a graph showing a relationship between a torque command and a torque error in a control apparatus of a conventional induction motor to which a sliding frequency type vector control method is applied. In Fig. 12, the horizontal axis is the torque command, and the vertical axis is the torque error (= generated torque-torque command). 12 shows the relationship between the torque command and torque error when the rotational speed is 3 [rad / s], and the lower figure shows the torque command and torque error when the rotational speed is 188 [rad / s]. The relationship is shown. Moreover, the solid line shows the characteristic when the secondary resistance of an induction motor changes 1.3 times, and the broken line shows the characteristic when the secondary resistance of an induction motor changes 1/1/3 times.

As shown in Fig. 12, in the conventional induction motor control apparatus to which the sliding frequency type vector control method is applied, there is a problem that a torque error occurs when the secondary resistance value changes regardless of the rotational speed.

Moreover, although the method of identifying the secondary resistance value during the operation of the induction motor has been proposed, there has been a problem in terms of stability such as the estimated value of the secondary resistance is diverged depending on the operating conditions.

In order to solve this problem, the control apparatus of the induction motor which applied the magnetic flux observer which calculates estimated secondary magnetic flux according to the primary current, primary voltage, and motor constant of an induction motor is proposed.

For example, FIG. 13 shows a control apparatus of a conventional induction motor shown in the document "Slip frequency control equivalent to magnetic flux feedback control of an induction motor" (1992, National Institute of Electrical and Electronics Engineers Conference 110, pages 466-471). It is a block diagram shown.

First, the control principle by the control apparatus of the conventional induction motor will be described. The magnetic flux observer formed on the stationary two-axis (α-β axis) is configured by equations (1) and (2).

Also, the square matrices K ₁ and K ₂ are determined by equations (3) and (4) in order to arrange the poles of the magnetic flux observer in the conjugated complex pole or the middle pole, and k ₁ , k ₂ , k ₃ , k depending on the rotational speed. Determine ₄ .

Therefore, if Equation 1 is coordinate-converted into the stator polar coordinates on the rotation two axis (d-q axis), the equations (5) to (7) are obtained.

That is, based on equations (5) to (7), magnetic flux calculations equivalent to the magnetic flux observer on the α-β axis are possible on the d-q axis.

Here, the two square matrices K ₁ and K ₂ are designed to have two static axes. That is, square matrices K ₁ and K ₂ are determined by equations (3) and (4), and k ₁ , k ₂ , k ₃ , and k ₄ are determined according to the rotation speed. At this time, the relationship of Equation 8 is necessarily established between K ₁ and K ₂ .

In this way, it is said that the law of substituting holds for the relationship between the interchangeable matrices K ₁ and K ₂ . Equations 5 to 7 can be rewritten as in Equations 9 to 13.

Therefore, based on equations (9) to (13), the phase and amplitude of the estimated secondary magnetic flux can be obtained with the same precision as the magnetic flux observer constituted on the stationary biaxial (α-β axis). Therefore, based on this phase, if d-axis current and q-axis current are respectively controlled to a desired value, deterioration of control performance by a secondary resistance change can be suppressed.

Next, the structure of the control apparatus of the conventional induction motor shown in FIG. 13 is demonstrated. The control device of this induction motor 1 is a primary obtained from the rotational speed detector 2, the current detector 3, the magnetic flux observer 4, the control means 5, the amplification means 6 and the current detector 3. Phase of secondary flux estimated current Has a coordinate converter 7 for coordinate transformation on the rotational coordinate axis dq axis. The control means 5 also has a current controller 8, a coordinate converter 9 and a PWM inverter 10.

The amplifying means 6 has subtractors 11 and 12 and gain calculators 13 and 14. The rotational speed detector 2 detects the rotational speed ω _m of the induction motor 1, and the current detector 3 detects the primary currents i _us , i _vs of the induction motor 1.

The magnetic flux observer 4 is based on the primary voltage commands applied to the induction motor 1, V _ds ^* , V _qs ^* and the deviation signals z ₁ , z ₂ , z ₃ , z ₄ obtained from the amplifying means 6. The amplitude of the estimated secondary magnetic flux of the induction motor 1 Phase of estimated secondary flux , The d-axis component i _ds of the estimated primary current and the q-axis component i _qs of the estimated primary current are estimated.

The control means 5 comprises the phase of the estimated secondary magnetic flux obtained from the magnetic flux observer 4. Based on this, the voltage applied to the induction motor 1 is controlled so that the primary current coincides with the desired current applied on the dq axis. That is, the current controller 8 responds to the desired current applied on the dq axis (d-axis primary current command i _ds ^* , q-axis primary current command i _qs ^* ), d-axis primary current i _ds , q-axis primary. The primary voltage commands V _ds ^* and V _qs ^* on the dq axis are output so that the currents i _qs coincide with each other, and the coordinate converter 9 phases the estimated secondary magnetic flux. Based on the above, the three-phase voltage commands v _us ^* , v _vs ^* , and v _ws ^* are calculated. The PWM inverter 10 applies the three-phase voltages v _us , v _vs , v _ws to the induction motor 1 based on these three-phase voltage commands v _us ^* , v _vs ^* , v _ws ^* .

The amplifying means 6 obtains the output of the current detector 3 as the primary current on the dq axis via the coordinate transducer 7 and the estimated primary current on the dq axis obtained from the magnetic flux observer 4 and on the dq axis. The deviation of the primary current is amplified based on two square matrices K ₁ , K ₂ and output as deviation signals z ₁ , z ₂ , z ₃ , z ₄ .

That is, the subtractor 11 is the d-axis estimated primary current obtained from the magnetic flux observer 4 And the deviation between the d-axis primary current i _ds obtained from the coordinate converter 7 - Calculate The subtractor 12 is the q-axis estimated primary current obtained from the magnetic flux observer 4 And the deviation between the q-axis primary current i _qs obtained from the coordinate converter 7 - Calculate The gain calculator 13 calculates the deviation signals z ₁ and z ₂ based on the first square matrix K ₁ of Equation 9, and the gain calculator 14 deviations based on the second square matrix K ₂ of Equation 10. Calculate the signals z ₃ and z ₄ . In addition, since the first square matrix and the second square matrix are functions of the rotation speed, the gain calculators 13 and 14 are functions of the rotation speed obtained from the rotation speed detector 2.

14 is a diagram illustrating an internal configuration of the magnetic flux observer 4. The magnetic flux observer 4 includes the matrix calculators 15 to 17, the gain calculators 18 to 21, the integrators 22 to 25, the adders 26 to 30, the adder and subtracter 31, the subtractors 32 to 34, and the first. It has an air diffuser 35.

A gain computing unit 21 outputs the Pm · ω _m by Pm times the output rotational speed ω _m of the detector (2). The gain operator 16 outputs the integrator 22 And output of integrator (23) Is input, and the right side claim 1 expression (11) is performed based on the output? Of the adder 30. Gain calculator 17 outputs integrator 24 Is input, and the right side term | claim 2 of Formula (11) is computed based on the output Pm * (omega) _m of the gain calculator 21. The gain calculator 15 inputs the primary voltage commands V _ds ^* and V _qs ^* obtained from the control means 5 to perform the calculation of the right term 3 of Equation (11).

The right side of the equation (11) is calculated by the adders 26, 27, 28, 29 and the subtractors 33, 34, , It is input to the integrators 22 and 23 as the derivative of. Integrators 22 and 23 are reminded , Integrate each of the derivatives of , Outputs

Gain calculator 19 outputs integrator 24 To perform the calculation of the right side claim 1 of Equation (12). The gain operator 18 outputs the integrator 22 To perform the calculation of the right side second term of the expression (12). The adder and subtractor 31 calculates the right side of Equation 12 Is input to the integrator 24 as the derivative of. Integrator 24 reminds By integrating the derivative of Outputs

The gain operator 20, the subtractor 32, and the divider 35 perform the calculation of the right side second term in equation (13). The adder 30 calculates the right side of Equation 13 Output the derivative of, ω. The integrator 25 integrates the output ω of the adder 30 Outputs

In this way, the magnetic flux observer 4 has the rotational speed obtained from the rotational speed detector 2 and the primary voltage commands V _ds ^* , V _qs ^* of the induction motor 1 and the deviation signal z ₁ , obtained from the amplifying means 6. Estimated secondary magnetic flux of induction motor 1 based on z ₂ , z ₃ , z ₄ , And estimated primary current of the induction motor (1) , Calculate

FIG. 15 is a graph showing a relationship between a torque command and a torque error in a control apparatus of a conventional induction motor to which the magnetic flux observer shown in FIGS. 13 and 14 is applied. In Fig. 15, the horizontal axis is the torque command, and the vertical axis is the torque error (= generated torque-torque command). The upper figure of FIG. 15 shows the case where a rotational speed is 3 [rad / s], and the lower figure shows the case of 188 [rad / s]. In addition, the solid line shows the characteristic when the secondary resistance of the induction motor l is changed 1.3 times, and the broken line shows the characteristic when the secondary resistance of the induction motor 1 is changed to 1 / 1.3 times.

As can be seen by comparing FIG. 12 with FIG. 15, the control device of the conventional induction motor to which the magnetic flux observer shown in FIGS. 13 and 14 is applied is compared with the control device of the induction motor to which the sliding frequency type vector control method is applied. If the rotational frequency is 188 [rad / s] without estimating the secondary resistance during the operation of 1), the torque error can be reduced.

However, there is a problem that the effect is small in the low speed range where the rotation frequency is 3 [rad / s]. This is because, in the case of designing two square matrices K ₁ and K ₂ of the amplifying means 6, the control apparatus of the conventional induction motor is constrained that k ₁ and k ₂ are interchangeable with each other. This is because the influence of integer error cannot be suppressed properly.

In particular, electric vehicles, such as electric cars, are generally driven by torque control, and the case where torque is most needed when the electric vehicle is driven is at the time of starting and stopping. Therefore, high-precision torque control is required for both retrograde and regeneration in an area close to zero speed.

Moreover, in a printing press, it is connected to an electric motor through many gears. Therefore, at the time of starting, it accelerates gradually, rotating at the ultra low speed initially. In this case, the change in the precision of the torque control causes the speed response to change due to the speed control. If the precision of the torque control at low speed is poor, there is a problem that the reproducibility is lost at the start of the operation, the end of the operation, the summer, the winter, and the like, making it difficult to adjust.

1 is a block diagram illustrating the control contents of a control device of an induction motor when there is a resistance error;

2 is a view showing elements of the square matrix H ₁ , H ₂ of the control device of the induction motor according to the first embodiment of the present invention;

3 is a block diagram showing an overall configuration of a control device of an induction motor according to the first embodiment of the present invention;

4 is a block diagram showing the configuration of a magnetic flux observer of the control device of the induction motor according to the first embodiment of the present invention;

5 is a diagram showing the torque control accuracy of the control device of the induction motor according to the first embodiment of the present invention,

6 is a view showing elements of the square matrix H ₁ , H ₂ of the control device of the induction motor according to the second embodiment of the present invention;

7 is a block diagram showing the overall configuration of a control device of an induction motor according to a second embodiment of the present invention;

8 is a diagram showing torque control accuracy of the control device of the induction motor according to the second embodiment of the present invention;

9 is a block diagram showing an overall configuration of a control device of an induction motor according to a third embodiment of the present invention;

10 is a block diagram showing the overall configuration of a control device of an induction motor according to a fourth embodiment of the present invention;

11 is used in Example 5 of the present invention, Wow A circuit configuration for obtaining i _ds ^* from

12 is a diagram showing torque control accuracy of a control device of a conventional induction motor;

13 is a block diagram showing the overall configuration of a control device of a conventional induction motor;

14 is a block diagram showing a configuration of a magnetic flux observer used in a control apparatus of a conventional induction motor;

FIG. 15 is a graph showing the relationship between the torque command and torque error in the control apparatus of the conventional induction motor to which the magnetic flux observer shown in FIGS. 13 and 14 is applied.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and by applying two square matrices to the amplifying means 6 which appropriately suppresses the influence of motor constant errors such as resistance, torque errors resulting from motor constant errors such as resistance, etc. It is an object of the present invention to obtain a control device of an induction motor and a control method thereof capable of suppressing this.

In order to solve this problem, the control apparatus of the induction motor according to the present invention includes a rotation speed detector for detecting the rotational speed of the induction motor, a current detector for detecting the primary current of the induction motor, and an estimation 1 obtained from the magnetic flux observer. Amplification means for amplifying a deviation between the difference current and the primary current obtained from the current detector, the induction based on the rotation speed obtained from the rotation speed detector and the primary voltage of the induction motor and the deviation signal obtained from the amplification means. A magnetic flux observer for estimating estimated secondary magnetic flux and estimated primary current of the electric motor, and control means for controlling a voltage applied to the induction motor based on the estimated secondary magnetic flux obtained from the magnetic flux observer; two square matrices H _1, H _2, that is commutative rule is not satisfied, and each independent element 8 Since the deviation of the primary current is amplified based on the feedback gain, the deterioration of the torque control accuracy due to the constant parameter error of the motor is suppressed without being constrained to arrange the pole of the magnetic flux observer in the conjugated complex pole or the middle pole. can do.

In the above-described invention, the control device of the induction motor according to the present invention is such that the amplifying means determines the feedback gain based on the rotational angular velocity. Deterioration of the torque control precision resulting can be suppressed.

In the above-described invention, the control device of the induction motor according to the present invention allows the amplifying means to determine the feedback gain on the basis of the sliding angular frequency, and therefore, even when the load torque is changed, Deterioration of torque control accuracy can be suppressed.

In the above-mentioned invention, the control device of the induction motor according to the present invention makes the amplification means determine the feedback gain based on both the rotational angular velocity and the sliding angular frequency. Even if is changed, the deterioration of the torque control accuracy caused by the motor constant error can be suppressed.

In the control apparatus of the induction motor according to the following invention, in the above invention, the amplifying means is the following equation:

8-Independent Feedback Gain Since the deviation of the primary current is amplified on the basis of this, it is possible to suppress the deterioration of the torque control accuracy caused by all the motor constant errors.

In the control apparatus of the induction motor according to the following invention, in the above invention, the amplifying means is the following equation:

8-Independent Feedback Gain Since the deviation of the primary current is amplified on the basis of this, it is possible to suppress deterioration in torque control accuracy caused by the primary resistance error and the secondary resistance error.

In the control apparatus of the induction motor which concerns on the following invention, in the said invention, the said amplification means is a following formula.

8-Independent Feedback Gain Satisfies Since the deviation of the primary current is amplified on the basis of this, it is possible to suppress the deterioration of the torque control accuracy caused by the secondary resistance error.

Hereinafter, with reference to the accompanying drawings, a preferred embodiment of the control device of the induction motor according to the present invention will be described in detail.

(Example 1)

First, Embodiment 1 of the present invention will be described. Here, before explaining the specific structure of Embodiment 1 of this invention, the operation principle of the control apparatus of this induction motor is demonstrated. First, the state equation of the induction motor on the d-q axis can be expressed by equations (14) to (16).

Here, by the equations (17) to (19), the magnetic flux observer can be configured on the d-q axis.

As described above, the induction motor changes its primary resistance and secondary resistance due to heat generation during operation. For example, when R _s and R _r are changed by (1 + k) times, equations (4) and (15) change as shown in equations (20) and (21).

However, Equations 20 and 21 can be rewritten as in Equation 22.

In addition, when the vector control of the induction motor is operating correctly, the equations (23) to (25) hold in the steady state.

Thus, by substituting Equations 23 to 25 into Equation 22, Equation 26 is obtained.

In addition, when A, B ₁ , B ₂ , C, D ₂ , and w ₂ are defined by the equations 27 to 32, the induction motor formed by the equations 19 and 26 may be expressed as a block diagram shown in FIG. 1. It's okay.

Where ε is a random integer small enough, w ₁ is an arbitrary variable. In the case where the control target is written as shown in Fig. 1, in general, B ₂ · w ₂ is called state noise, and D ₂ · w ₁ is called observation noise.

State estimation error of the magnetic flux observer from the noises w ₁ and w ₂ of the system shown in the block diagram of FIG. _{s- s} , _{r- The} two square matrices H ₁ and H ₂ of the amplifying means for minimizing the energy of the impulse response up to _r ) may be given by the following equation (33). Where P is the only solution to the definition that satisfies Equation 34, called the Ricati equation.

Since A and B ₂ of Equation 34 are functions of the rotational angular velocity ω _m and the first order angular frequency ω, H ₁ and H ₂ given by Equation 33 are also functions of the rotational angular velocity ω _m and the first order angular frequency ω. do.

When the difference between ω and P _m ω _m is the sliding angular frequency ω _s , the elements of H ₁ and H ₂ obtained from equations 33 and 34, h11, h12, h21, h22, h31, h32, h41, h42 Becomes a function as shown in FIG.

In the conventional induction motor control apparatus, in order to arrange the poles of the magnetic flux observer in the conjugated complex pole or the mid pole, the square matrices K ₁ and K ₂ are interchangeable, but as shown in FIG. 2, H ₁ H _{Since 2} ? H ₂ H ₁ , the ring law does not hold between the two square matrices H ₁ and H ₂ of the first embodiment.

In addition, by converting equation (18) into stator polar coordinates, equations (17) to (19) can be rewritten as equations (35) to (39).

Here, the control apparatus of the induction motor of Embodiment 1 of this invention is demonstrated. 3 is a block diagram showing a configuration of a control device of an induction motor according to the first embodiment of the present invention.

In FIG. 3, an induction motor 1, a rotation speed detector 2, a current detector 3, a control means 5, a coordinate converter 7, a current controller 8, a coordinate converter 9, a PWM inverter ( 10) is the same as the control device of the conventional induction motor shown in FIG.

The amplifying means 6a has subtractors 11a and 12a and gain calculators 13a and 14a. In addition, the magnetic flux observer 4a is based on the primary voltage commands V _ds ^* , V _qs ^* of the induction motor 1 and the deviation signals e ₁ , e ₂ , e ₃ , e ₄ obtained from the amplifying means 6a. Estimated Secondary Magnetic Flux Amplitude of Induction Motor Phase of estimated secondary flux , The d-axis component i _ds of the estimated primary current and the q-axis component i _qs of the estimated primary current are estimated.

The amplifying means 6a obtains the output of the current detector 3 as the primary current on the dq axis via the coordinate transducer 7, and the estimated primary current on the dq axis obtained from the magnetic flux observer 4a and the 1 on the dq axis. The deviation of the difference current is amplified based on the two square matrices H ₁ and H ₂ and output as deviation signals e ₁ , e ₂ , e ₃ , and e ₄ .

That is, the subtractor 11a is the d-axis estimated primary current obtained from the magnetic flux observer 4a. And the deviation between the d-axis primary current i _ds obtained from the flux converter 7 - , And the subtractor 12a calculates the q-axis estimated primary current obtained from the magnetic flux observer 4a. And the deviation of the q-axis primary current i _qs obtained from the coordinate converter 7 - The gain operator 13a calculates the deviation signals e ₁ and e ₂ based on the first square matrix H ₁ of Equation 33, and the gain calculator 14a calculates the second square matrix H ₂ of Equation 33. Based on the calculation, the deviation signals e ₃ and e ₄ are calculated.

In addition, since the first square matrix and the second square matrix are functions of the rotational speed ω _m and the sliding angular frequency ω _s as shown in FIG. 2, the gain calculators 13a and 14a are provided from the rotational speed detector 2. Let it be a function of the rotation speed ω _m obtained and the sliding angular frequency ω _s obtained from the magnetic flux observer 4a.

4 is a block diagram showing the configuration of the magnetic flux observer 4a. The magnetic flux observer 4a has subtractors 36 to 39, adders 40 to 42, integrators 43 to 46, dividers 47, gain operators 48 and 49, and matrix operators 50 and 51. .

The matrix operator 50 performs the matrix operation of the right side claim 1 of Equation 37 based on the angular frequency ω described later. The subtractors 36 to 38 and the adders 40 and 41 perform calculations on the right side of Equation 37 based on the output of the matrix operator 50 and V _ds ^* , V _qs ^* , e _{1 to} e ₄ , The derivative of is transferred to the integrator 43, The derivative of to the integrator 44, The derivative of is input to integrator 45, respectively. Integrator 43 Integrate the derivative of, Output as. Integrator 44 is By integrating the derivative of Output as. Integrator 45 By integrating the derivative of Output as.

The subtractor 39, the gain calculator 49, and the divider 47 perform the calculation of the right side second term in equation (38), and the adder 42 adds the output of the divider 47 and the output of the gain 48. Thus, the right side of the equation (38), that is, the primary angular frequency ω is obtained. In addition, since the output of the divider 47 corresponds to the difference between the primary angular frequency ω and the rotational angular frequency Pmω _m , it is equal to the sliding angular frequency ω _s . Integrator 46 integrates the angular frequency ω Outputs The matrix operator 51 performs the calculation on the right side of the equation 39 based on the outputs of the integrators 43 to 45, , Outputs

In this way, the magnetic flux observer 4a is based on the primary voltage commands V _ds ^* , V _qs ^* , the deviation e _{1 to} e ₄ , and the rotational angular velocity ω _m . , , , , ω _s

As a result, in the first embodiment, the output torque of the induction motor can be accurately controlled regardless of the temperature change.

5 is a graph showing a relationship between a torque command and a torque error in the control device of the induction motor according to the first embodiment of the present invention. In Fig. 5, the horizontal axis represents the torque command, and the vertical axis represents the torque error (= generated torque-torque command). 5 shows when the rotational speed is 3 [rad / s], and the lower figure shows when 188 [rad / s]. In addition, the solid line shows the characteristic when the primary resistance and the secondary resistance of the induction motor 1 are changed 1.3 times, and the broken line is 1 / 1.3 times the primary resistance and the secondary resistance of the induction motor 1. The characteristic at the time of change is shown.

As can be seen by comparing FIG. 5 with FIG. 12 and FIG. 15, in the induction motor control apparatus shown in the first embodiment, even if there is a resistance error regardless of the rotational angular velocity than the control apparatus of the conventional induction motor, the torque error is reduced. It can be suppressed.

(Example 2)

Then, Example 2 of this invention is described. In Example 1 described above, the case where H ₁ and H ₂ became a function of the rotational angular velocity ω _m and the sliding angular frequency ω _s has been described, but the relation between the rotational angular velocity ω _m and the sliding angular frequency ω _s is almost clearly determined. Assuming a load to lose, H ₁ and H ₂ may be functions only of the rotational angular velocity ω _m .

For example, assuming a load in which the primary angular frequency ω is a small value Δω, the sliding angular frequency is clearly determined from the rotational angular velocity ω _m as shown in equation (40).

FIG. 6 is a diagram showing a relationship between the rotation frequency and H ₁ , H ₂ , elements h11, h12, h21, h22, h31, h32, h41, and h42 when Equation 40 is established. As shown in Fig. 6, the ring law does not hold between the two square matrices H ₁ and H ₂ of the second embodiment.

Thus, assuming a load in which the relationship between the rotational angular velocity ω _m and the sliding angular frequency ω _s is almost clearly determined, it is possible to give H ₁ , H ₂ as a function of only the rotational angular velocity ω _m , and the amount of calculation of the amplifying means. Can be reduced.

7 is a block diagram showing a configuration of a control device of an induction motor according to a second embodiment of the present invention. In FIG. 7, the induction motor 1, the rotational speed detector 2, the current detector 3, the magnetic flux observer 4a, the control means 5, the coordinate transducer 7, the current controller 8, the coordinate transducer ( 9) and the PWM inverter 10 are the same as the control device of the induction motor according to the first embodiment shown in FIG.

The amplifying means 6b has subtractors 11b and 12b and gain calculators 13b and 14b. The amplifying means 6b obtains the output of the current detector 3 as the primary current on the dq axis via the coordinate converter 7 and the estimated primary current on the dq axis obtained from the magnetic flux observer 4a and 1 on the dq axis. The deviation of the difference current is amplified based on the two square matrices H ₁ and H ₂ and output as the deviation signals e _{1 to} e ₄ . That is, the subtractor 11b is the d-axis estimated primary current obtained from the magnetic flux observer 4a. And the deviation between the d-axis primary current i _ds obtained from the coordinate converter 7 - The subtractor 12b calculates the q-axis estimated primary current obtained from the magnetic flux observer 4a. And the deviation of the q-axis estimated primary current i _qs obtained from the coordinate converter 7 - And the gain calculator 13b calculates the deviation signals e ₁ and e ₂ based on the first square matrix H ₁ . The gain calculator 14b calculates the deviation signals e ₃ and e ₄ based on the second square matrix H ₂ .

In addition, since the first square matrix and the second square matrix are functions of the rotational angular velocity as shown in FIG. 6, the gain calculators 13b and 14b are functions of the rotational speeds obtained from the rotational speed detector 2. .

8 is a graph showing a relationship between a torque command and a torque error in the control device of the induction motor according to the second embodiment of the present invention. In Fig. 8, the horizontal axis is the torque command, and the vertical axis is the torque error (= generated torque-torque command). 8 shows when the rotational speed is 3 [rad / s], and the lower figure shows when 188 [rad / s]. In addition, the solid line shows the characteristic when the primary resistance and the secondary resistance of the induction motor 1 are changed by 1.3 times, and the broken line shows the characteristic when the primary resistance and the secondary resistance of the induction motor are changed by 1 / 1.3 times. The characteristics are shown.

As can be seen by comparing FIG. 8 with FIG. 12 and FIG. 15, the control device of the induction motor according to the second embodiment has a torque error even if there is a resistance error, regardless of the rotational angular velocity, compared to the control device of the conventional induction motor. It can be suppressed.

(Example 3)

Then, Example 3 of this invention is described. In the second embodiment described above, the amplifying means (6b) of square matrices H _1, H _2, based on the the basis of the rotational angular velocity has been described as to obtain a square matrix H _1, H _2, slip angular frequency in place of the rotation angular velocity You may obtain, and also in this case, the same effect as Example 2 mentioned above can be acquired.

9 is a block diagram showing a configuration of a control device of an induction motor according to a third embodiment of the present invention. In FIG. 9, the induction motor 1, the rotational speed detector 2, the current detector 3, the magnetic flux observer 4a, the control means 5, the coordinate transducer 7, the current controller 8, the coordinate transducer ( 9) and the PWM inverter 10 are the same as the control device of the induction motor according to the first embodiment shown in FIG.

The amplifying means 6c has subtractors 11c and 12c and gain calculators 13c and 14c. The amplifying means 6c obtains the output of the current detector 3 as the primary current on the dq axis via the coordinate converter 7, and the estimated primary current on the dq axis obtained from the magnetic flux observer 4a and 1 on the dq axis. The deviation of the difference current is amplified based on the two square matrices H ₁ and H ₂ and output as the deviation signals e _{1 to} e ₄ .

That is, the subtractor 11c is the d-axis estimated primary current obtained from the magnetic flux observer 4a. And the deviation between the d-axis primary current i _ds obtained from the coordinate converter 7 - , And the subtractor 12c calculates the q-axis estimated primary current obtained from the magnetic flux observer 4a. And the deviation of the q-axis primary current i _qs obtained from the coordinate converter 7 - The gain calculator 13c calculates the deviation signals e ₁ and e ₂ based on the first square matrix H ₁ . The gain calculator 14c calculates the deviation signals e ₃ and e ₄ based on the second square matrix H ₂ .

In addition, since the first square matrix and the second square matrix are functions of the sliding angular frequency, the gain calculators 13b and 14c are said to be functions of the sliding angular frequency obtained from the magnetic flux observer 4a.

(Example 4)

Then, Example 4 of this invention is described. In the above-described square matrices H ₁ and H ₂ described in Examples 1 to 3, the case where there is an error in the primary resistance and the secondary resistance has been described, but the square matrix H ₁ , H ₂ can be determined.

In general, the temperature of the primary resistance of the induction motor can be detected by a temperature detector using a thermocouple or the like. By using this detection temperature, the primary resistance value of the induction motor can be calculated. However, in the case of a squirrel-cage induction motor, it is difficult to measure the temperature of the secondary resistance with a temperature detector such as a thermocouple.

Therefore, the case where R _s is known and R _r is changed by (1 + k) times will be described. When R _s is known and R _r is changed by (1 + k) times, equations (14) and (15) can be rewritten as in equation (41).

As described above, when the vector control of the induction motor is operating correctly, the equations 23 to 25 hold in the steady state. Thus, by substituting Equations 23 to 25 into Equation 41, Equation 42 is obtained.

If B ₂ is given by Equation 43 instead of Equation 29 and w ₂ is given by Equation 44 instead of Equation 32, the induction motor composed of Equations 19 and 42 may also be expressed as shown in FIG. It's okay.

If B ₂ is redefined by Equation 43 and w ₂ by Equation 44, and two square matrices H ₁ and H ₂ are given by Equation 33, torque error in the case where there is an error only in the secondary resistance is suppressed. can do.

In addition, similarly to the first embodiment, since A in the equation (34) is a function of the rotational angular velocity ω _m and the primary angular frequency ω, H ₁ , H ₂ given by Equation 33 is also the rotational angular velocity ω _m and the primary angular frequency. It becomes a function of ω.

In addition, as in the above-described Examples 1 to 3, the ring law does not hold between the two square matrices H ₁ and H ₂ of the fourth embodiment.

10 is a block diagram showing the configuration of a control device of an induction motor according to a fourth embodiment of the present invention. In FIG. 10, the induction motor 1, the rotational speed detector 2, the current detector 3, the control means 5, the coordinate converter 7, the current controller 8, the coordinate converter 9 and the PWM inverter ( 10) is the same as the control device of the induction motor according to the first embodiment shown in FIG.

The temperature detector 52 measures the temperature of the primary resistance of the induction motor 1. The resistance value calculator 53 outputs the primary resistance value R _s based on the temperature T of the primary resistance obtained from the temperature detector 52. The magnetic flux observer 4d is the same as the magnetic flux observer 4a except that the primary resistance R _s uses the value obtained from the resistance value calculator 53.

The amplifying means 6d has subtractors 11d and 12d and gain calculators 13d and 14d. The amplifying means 6d is the same as the amplifying means 6a except that two square matrices H ₁ and H ₂ obtained based on B ₂ and w ₂ defined by equations 43 and 44 are used.

Thereby, even if an error occurs in the secondary resistance value under the influence of temperature conversion, the torque error can be suppressed.

(Example 5)

Then, Example 5 of this invention is described. When the two square matrices H ₁ and H ₂ described above are applied to the amplifying means, the amplitude value of the estimated secondary magnetic flux calculated by the magnetic flux observer Also improves the precision.

Therefore, the d-axis current command i _ds ^* may be determined so that the amplitude of the estimated secondary magnetic flux becomes a desired secondary magnetic flux amplitude value using the circuit configuration as shown in FIG. 11. In Fig. 11, the subtractor 54 has a desired secondary magnetic flux amplitude value. ^* Amplitude value of estimated secondary magnetic flux Is calculated, and the amplifier 55 amplifies the output of the subtractor 54 and outputs it as the d-axis current command i _ds ^* .

(Example 6)

Subsequently, a sixth embodiment of the present invention will be described. In the above-described embodiment, the case where the square matrixes H ₁ and H ₂ have an error in the resistance value has been described, but the matrix is similarly applied to all the motor constant errors such as mutual inductance M, primary inductance Ls, and secondary inductance Lr. It is a matter of course that H ₁ and H ₂ for suppressing deterioration in torque control accuracy can be determined by appropriately determining both B ₁ or matrix D _2, or matrix B ₁ and matrix D ₂ .

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device of an induction motor capable of controlling the generated torque of an induction motor with high precision. In particular, the present invention relates to a control device of an induction motor suitable for a device requiring high-precision torque control at a low speed such as an electric vehicle or a printing machine. It is available.

Claims (7)

delete A rotational speed detector for detecting a rotational speed of the induction motor, A current detector for detecting a primary current of the induction motor, Amplifying means for amplifying a deviation between the estimated primary current obtained from the magnetic flux observer and the primary current obtained from the current detector; A magnetic flux observer for estimating estimated secondary magnetic flux and estimated primary current of the induction motor based on the rotational speed obtained from the rotational speed detector, the primary voltage of the induction motor, and the deviation signal obtained from the amplifying means; And control means for controlling a voltage applied to the induction motor based on the estimated secondary magnetic flux obtained from the magnetic flux observer, The amplification means determines a feedback gain comprising two square matrices which are invertible with each other based on the rotational angular velocity, and amplifies the deviation of the primary current based on this feedback gain. Induction motor control device, characterized in that. delete delete A rotational speed detector for detecting a rotational speed of the induction motor, A current detector for detecting a primary current of the induction motor, Amplifying means for amplifying a deviation between the estimated primary current obtained from the magnetic flux observer and the primary current obtained from the current detector; A magnetic flux observer for estimating estimated secondary magnetic flux and estimated primary current of the induction motor based on the rotational speed obtained from the rotational speed detector, the primary voltage of the induction motor, and the deviation signal obtained from the amplifying means; And control means for controlling a voltage applied to the induction motor based on the estimated secondary magnetic flux obtained from the magnetic flux observer, The amplification means is the following equation 8-Independent Feedback Gain The control device of the induction motor, characterized in that for amplifying the deviation of the primary current based on. delete delete