KR100283518B1 - Transconductor with wide input dynamic range and motor drive system M - Google Patents

Abstract

The present invention relates to a transconductor having a wide input dynamic range and a motor driving system having the same. The reference voltage is shifted to the amplified value and output. The voltage amplifier receives a reference voltage and outputs a voltage corresponding to the input voltage. The first resistor is connected in series between the reference voltage amplifier and the differential amplifier. The current output unit outputs a current flowing through the first resistor in proportion to a difference between the first control voltage and the second control voltage, regardless of the reference voltage. In this way, the transconductor can be operated to have a wide dynamic range without affecting the magnitude of the servo's power supply voltage.

Description

Transconductor with wide input dynamic range and motor drive system with it

The present invention relates to a transconductor having a wide input dynamic range and a motor drive system having the same, and more particularly to a transconductor used for driving a CD-ROM spindle motor.

In general, a three-phase brushless direct current (BLDC) motor is used as a CD-ROM spindle motor. The output torque of the Bildish motor is proportional to the current flowing in the motor load. In spindle motor configurations, the system is often configured to sense the output current to control the rotational speed so that this current is proportional to the input control voltage.

1 is a configuration diagram for controlling the rotational speed of the spindle motor.

The control input voltages EC and ECR are input from the servo and are the control input voltages for the rotational speed control. The spindle motor depends on the sign of the difference between the two control input voltages, that is, the EC-ECR voltage is positive or negative. The direction of rotation of is determined. This control input voltage is output as a current through the transconductance amplifier 10, and this current is irrespective of the sign of the control input voltage difference so as to supply a drive current to the load motors M and 16 even in the forward reverse direction of the spindle motor. It always passes through the absolute value circuit 11 to supply a sourcing or sinking current and then enters the positive input of the motor drive amplifier 13 via the current / voltage converter 12. The driver 13 applies the three-phase outputs U, V, W of the power transistor 14 to the motors M, 16 so that the rotation of the motors M, 16 generates maximum torque.

At this time, the currents U, V, and W supplied to the motors M and 16 are converted into voltages through the output current sensing circuit 15 to be input to the negative input terminal of the driving amplifier 13 to form a negative feedback loop. .

By this negative feedback loop, the motor driving current at the output stage is proportional to the difference between the control input voltages EC and ECR to control the rotation speed of the motors M and 16.

2A shows the ideal transfer function characteristics of this system.

As shown in FIG. 2A, the motor driving current Io is proportional to the difference between the control input voltages EC and ECR.

However, the transmission function characteristic of the actual motor drive system is shaped as shown in FIG. 2B. When the difference between the two control input voltages (EC, ECR) is zero, the offset voltages (Eoff +, Eoff-) are set so that the actual motors (M, 16) do not rotate.The difference between the control input voltages (EC, ECR) is the offset voltage (Eoff +, This is to allow the motors M and 16 to rotate by supplying current to the motors M and 16 only when the value is equal to or greater than Eoff−.

In addition, if excess current is supplied to the motor M, 16, the motor M, 16 is damaged, so that the maximum value of the output current Io is limited to Iomax as shown in FIG. do.

The voltage swing of the control input voltage (EC, ECR) input from the servo depends on the power supply voltage of the servo. Generally, since the power supply voltage of the servo is 5V or 3.3V, the input dynamic range of each of the control input voltages (EC, ECR) That is, the voltage swing width is 3.3V or 5V, which is equal to the servo's maximum supply voltage.

In order to obtain the maximum input control voltage range in the motor speed control driving system of FIG. 1, one of EC or ECR is applied as a voltage corresponding to half of the servo power voltage, and the other control input is used as the reference voltage. Control the motor speed and direction by increasing or decreasing

Therefore, the difference between the control input voltages EC and ECR is 1.675V at maximum for a 3.3V servo and 2.5V at a 5V servo.

However, in the motor driving system of FIG. 1, the difference in the control input voltages EC and ECR, that is, the motor output current, is limited by the common mode input range of the transconductance amplifier 10 or the limit of the output stage maximum output swing. The maximum effective size that can be controlled is 1.5V. At the same time, the dynamic range of the control input voltages EC and ECR is about 0.2 to 3.3V for the 3.3V servo and about 1 to 4V for the 5V servo.

Thus, the conventional transconductance amplifier 10 has a maximum effective control input of 1.5V and cannot be applied to a 3.3V servo and a 5V servo. For this reason, in the related art, different transconductance amplifiers have to be manufactured for each servo, so that there is a big disadvantage in terms of time and material damage.

Therefore, the technical problem of the present invention is to solve the disadvantages of the prior art, and to implement a transconductance amplifier that can be applied to 3.3V servo and 5V servo simultaneously by making the output voltage independent of the power supply voltage of the servo.

1 is a configuration diagram for controlling the rotational speed of the spindle motor.

2A shows ideal transfer function characteristics of this system.

2b illustrates the practical transfer function characteristics of this system.

3 is a block diagram of a transconductor according to an embodiment of the present invention;

4 illustrates one example implementation of the differential amplifier of FIG.

5 is another exemplary implementation of the differential amplifier of FIG.

FIG. 6 is a detailed circuit diagram implementing the block diagram of FIG.

Figure 7 is a block diagram of a motor drive system according to an embodiment of the present invention.

In order to accomplish this problem, in the present invention, the differential amplifier receives the first and second control voltages and the reference voltage, shifts the reference voltage to a value obtained by amplifying the difference between the first control voltage and the second control voltage, and outputs the shifted reference voltage. . The voltage amplifier receives a reference voltage and outputs a voltage corresponding to the input voltage. The first resistor is connected in series between the reference voltage amplifier and the differential amplifier. The current output unit outputs a current flowing through the first resistor in proportion to a difference between the first control voltage and the second control voltage, regardless of the reference voltage. In this way, the transconductor can be operated to have a wide dynamic range without affecting the magnitude of the servo's power supply voltage.

Then, with reference to the accompanying drawings will be described an embodiment in which the present invention can be carried out.

3 is a block diagram illustrating a transconductor according to an embodiment of the present invention.

As shown in FIG. 3, the differential amplifier 40 receives the first and second control voltages and the reference voltage Vref. The output terminal is connected to the output terminal of the voltage amplifier 41 through the resistor (R). In addition, the voltage amplifier receives a reference voltage Vref. The current output unit 45 is connected to the voltage amplifier 41 to output a current flowing through the first resistor.

Next, the operation of the transconductor having such a structure will be described in detail.

The differential amplifier 40 amplifies the difference between the control voltages EC and ECR and performs a DC level shift function in which the DC voltage of the output is equal to the reference voltage Vref which is the other input. That is, the voltage at the output node n1 becomes Vn1 = Vref + (EC-ECR). The voltage amplifier 41 is a voltage follower, and the voltage of the output node n2 of the voltage amplifier 41 is represented by the input voltage Vref. That is, Vn2 = Vref.

In addition, the output current of the voltage amplifier 41 is equal to the current flowing through the resistor R, and outputs the current Iout through the current mirrors 42 and 43.

The difference between the voltages across the nodes n1 and n2, that is, the voltage across the resistor R, becomes Vn1-Vn2 = EC-ECR, and the current flowing through the resistor becomes (EC-ECR) / R. The value of the reference voltage (Vref) is half of the power supply voltage of the two amplifiers 40 and 41 so that the maximum output swing can be obtained from the differential amplifier 40 and the voltage amplifier 41 regardless of the servo power supply voltage. For example, if the power supply voltage of these two amplifiers is 5V, it is 2.5V. Then, the output of the differential amplifier 40 can maximize the symmetrical output when the difference is negative or positive when EC-ECR = 0. Therefore, the DC level of the node n1 can be set to obtain the maximum swing irrespective of the DC level of the EC and the ECR, so that the transconductor 44 having a wide input dynamic range can be implemented regardless of the power supply voltage of the servo. .

4 is an exemplary diagram for implementing the differential amplifier 40 of FIG.

As shown in FIG. 4, the operational amplifier 50 has a rail in which the input / output voltage region reaches from the ground to the power supply voltage VCC so that the in-phase input region of the control voltages EC and ECR can satisfy the 3.3V servo and the 5V servo at the same time. It has a rail-to-rail structure and can amplify the voltage difference between two input control voltages (EC, ECR) and make the DC voltage of the output (Vout) the reference voltage (Vref).

In FIG. 4, the output voltage Vout of the operational amplifier 50 has the same resistance value between the resistor R1 and the resistor R3, and the resistance values of the resistor R2 and the resistor R4 are the same, that is, R1 = R3. And R2 = R4, Vout = (EC-ECR) + Vref. However, in the differential amplifier 40 of FIG. 4, the input impedances of the control voltages EC and ECR are different from R3 + R4 and R1, respectively, and since the servo output is a pulse width modulated signal, the two control voltages EC and ECR are different. There is a filter just before the input stage of the circuit, where the cutoff frequency of the filter can be changed by the loading effect of the input impedance.

Therefore, an embodiment that can solve this disadvantage is shown in FIG.

FIG. 5 is another exemplary implementation of the differential amplifier 40 of FIG.

In FIG. 5, the transconductance amplifiers 51 and 52 receive a differential voltage and output a current corresponding to the difference. The transimpedance amplifier 53 receives a differential current and outputs a voltage corresponding to the difference.

Two control voltages EC and ECR are input to the two input terminals Va + and Va- of the transconductance amplifier 52, respectively, and the output current Io + is equal to Gm1 (EC-ECR). The output voltage Vout of the transimpedance amplifier 53 is fed back to the positive input terminal vb + of the other transconductance amplifier 51, and the output voltage Vout is input to the other negative input terminal vb-. The reference voltage Vref for determining the DC level is applied. At this time, Io- = Gm2 (Vout-Vref). In addition, since Io + = Io- by the negative feedback loop, Gm1 (EC-ECR) = Gm2 (Vout-Vref). Here, when Gm1 and Gm2 are equal, the output voltage Vout = (EC-ECR) + Vref is obtained, the DC level of the output Vout is the reference voltage Vref, and the swing is proportional to the EC-ECR. Therefore, the same problem as in the embodiment of FIG. 4 does not occur.

FIG. 6 is a detailed circuit diagram of the block diagram of FIG.

In FIG. 6, 40 represents a part of the differential amplifier 40 of FIG. 3, 41 represents the voltage amplifier 41 of FIG. 3, and 45 represents the current output unit 45 of FIG. 3.

In Fig. 6, an emitter follower stage 62 is provided so that the in-phase input voltages of the EC and the ECR can be operated from the ground level, and the input PNP transistors QP1 and QP2 of the emitter follower are output (PNP transistors). Increase the DC level of the emitter by Vbe relative to the input.

51, 52 and 53 of FIG. 6 correspond to the transconductance amplifiers 51 and 52 and the transimpedance amplifier 53 of FIG. 5, respectively. The output currents I1 and I2 of the transconductor 52 are represented by I1 = Gm1 ((EC + Vbe)-(ECR + Vbe)) = Gm1 (EC-ECR), I2 = Gm1 (ECR-EC) = − I1. Where Gm1 = 1 / (re1 + re2 + RE), re1 = re2 = (KT / q) / Ic, and KT / q is approximately 26mV at room temperature, and Ic is the bias current of the transistor.

Similarly, the output currents I3 and I4 of the transconductor 51 are I3 = Gm2 (Vout-Vref) and I4 = Gm2 (Vref-Vout), respectively, and Gm2 = 1 / (re3 + re4 + RE). However, since the bias currents of the transistors QN1 to QN4 are all equal to I, re1 = re2 = re3 = re4, resulting in Gm1 = Gm2.

The two input currents Io + and Io- of the transimpedance amplifier 53 are represented by Io + = − (I1 + I3) and Io − = − (I2 + I4), respectively.

If the pressure current Io + of the transimpedance amplifier 53 increases and Io- decreases, the voltage of the node n3 increases and the voltage of the node n4 decreases. These two input currents Io + and Io- are input to current mirrors QN5 and QN6 through common base transistors QP3 and QP4.

Increasing Io + increases the current in QN6, which in turn increases the current in QN5. At this time, since Io- is decreasing, the current of QP3 also decreases, and eventually the potential of the QN5 collector (QN7 base) also decreases. Since the potential of QN7 is lowered, the base potential of QN10 is also lowered, thereby increasing the collector potential of QN10. The collector potential of QN10 is connected to the output Vout through the push-pull output terminals QN8, QN9, QP5 and QP6, so that the potential of Vout becomes high. As a result, when Io + increases (Io- decreases), the output voltage Vout also increases.

When the EC-ECR goes positive, that is, when the EC voltage level is greater than the ECR, I1 increases and I2 decreases by an increment of I1. As I1 increases and I2 decreases, Io + increases and Io- decreases, resulting in an increase in output voltage Vout. As the output voltage (Vout) increases, I3 current increases and I4 current decreases, resulting in a decrease in Io +, an increase in negative feedback, and finally the change by (EC-ECR) of Io + and Io-. The minute is offset by the negative feedback loop from the change in I3 and I4. In other words, the negative feedback loop causes the change of Io + and Io- to be zero, resulting in I1 = -I4 and I2 = -I3. Since I2 and I3 are I2 = Gm1 (ECR-EC) and I3 = Gm2 (Vout-Vref), respectively, and Gm1 = Gm2, eventually Vout = Vref + (EC-ECR) .The output voltage is the difference between the two inputs, EC- This is the value obtained by DC level shifting the ECR to Vref.

Since the voltage amplifier 41 is configured as a voltage follower, the emitter (QP9 emitter) voltage of QN11 becomes Vref. Therefore, the voltage across the resistor R becomes Vout-Vref = EC-ECR, and the current becomes (EC-ECR) / R. If the EC-ECR is positive, the output voltage (Vout) of the differential amplifier goes positive, so the emitter voltage of QP9 (emitter of QN11) goes positive, and the voltage amplifier lowers the base voltage of QP9, which turns on QP9. QN11 is off and the current flowing in QP9 becomes equal to the current flowing in resistor R. The current flowing through the QP9 is represented by the current mirror 43 as the output current Iout, and the current at this time becomes the sinking current and the amount is (EC-ECR) / R.

On the contrary, when the EC-ECR goes negative, QN11 is turned on, QP9 is turned off, and the current flowing in QN11 becomes the sourcing output current (Iout) through the current mirror 42, and its magnitude is (EC-ECR) / R. do. This output current is input to the drive amplifier 13 via the absolute value circuit 11 and the current / voltage QQUSGHKSRL 12 of FIG.

Since the output current of FIG. 6 is proportional to the differential amplifier output voltage Vout swing, the DC level of the output voltage must be equal to half of the power supply voltage VCC in order to maximize the output swing and EC-ECR. However, this Vref voltage can be determined regardless of the DC level of EC and ECR, that is, the power supply voltage of the servo inputting these two voltages, 3.3V or 5V, so that Vref = VCC / to have a wide input dynamic range of (EC-ECR). Set to 2 to implement a transconductance amplifier with a wide area independent of the servo's power supply voltage.

7 shows an example in which the transconductor of the present invention is applied to a system for controlling the rotational speed of the spindle motor.

As shown in FIG. 7, control voltages EC and ECR are input to the transconductor 44 from an external servo 70.

Then, the transconductor 44 outputs a current proportional to the difference between the control voltages EC and ECR using half of the power supply voltage VCC as the reference voltage Vref, and this current also applies to the forward reverse direction of the spindle motor. After passing through the absolute value circuit 11 to supply the drive current to the load motors M and 16, it enters the positive input terminal of the motor drive amplifier 13 via the current / voltage converter 12.

Next, the driver 13 applies the three-phase outputs U, V, and W of the power transistor 14 to the motors M and 16 so that the rotation of the motors M and 16 generates maximum torque.

At this time, the currents U, V, and W supplied to the motors M and 16 are converted into voltages through the output current sensing circuit 15 to be input to the negative input terminal of the driving amplifier 13 to form a negative feedback loop. .

By this negative feedback loop, the motor driving current at the output stage is proportional to the difference between the control input voltages EC and ECR to control the rotation speed of the motors M and 16.

As described above, in the embodiment of the present invention, it is possible to provide a transconductor having a maximum effective input of 1.5V and simultaneously applicable to 3.3V servo and 5V servo, and a motor driving system having the same.

Claims (13)

A differential amplifier which receives a first control voltage and a second control voltage and shifts the reference voltage to a value obtained by amplifying a difference between the first control voltage and the second control voltage using half of a power supply voltage as a reference voltage; A reference voltage amplifier receiving the reference voltage and outputting a voltage corresponding to the input voltage; A first resistor connected in series between the reference voltage amplifier and the differential amplifier; A current output unit for outputting a current flowing through the first resistor irrespective of the reference voltage Transconductor comprising a. In claim 1, And the reference voltage is half the power supply voltage of the differential amplifier and the reference voltage amplifier to increase the maximum effective swing of the output. In claim 2, The differential amplifier, A second resistor having one side connected to the first control voltage, A third resistor having one side connected to the second control voltage, A fourth resistor having one side connected to the reference voltage and the other side connected to the third resistor; An operational amplifier having the other side of the second resistor connected to an inverting input terminal and the contact point of the third resistor and the fourth resistor connected to a non-inverting terminal; And a fifth resistor connected in parallel to an output terminal and an inverting input terminal of the operational amplifier. In claim 2, The differential amplifier, A first transconductance amplifier for outputting a current proportional to a difference between the first control voltage and the second control voltage; A second transconductance amplifier for outputting a current proportional to a difference between the reference voltage and an output voltage of the differential amplifier; And a transimpedance amplifier configured to output a voltage proportional to a difference current of the first and second transconductance amplifiers. In claim 4, And a emitter follower for increasing the in-phase input area by shifting the first and second control voltages input to the first transconductance amplifier. In claim 5, The emitter follower, A first PNP transistor having a base connected to the first control voltage, an emitter connected to a current source, and a collector grounded; And a second PNP transistor having a base connected to the second control voltage, an emitter connected to the current source, and a collector grounded. In claim 6, The first transconductance amplifier, A first NNP transistor having a base connected to a contact point of the current source and the first PNP transistor, a collector connected to a power source through a second resistor, and an emitter connected to the current source; A second NPI transistor having a base connected to a contact point of the current source and a second PNP transistor, a collector connected to the power source through a third resistor, and an emitter connected to the current source; And a fourth resistor connected in series between the emitters of the first and second NPI transistors. In claim 7, The second transconductance amplifier, A base connected to the reference voltage, a collector connected to a power source through the third resistor, a collector connected to a first NPE transistor of the first transconductance amplifier, and an emitter connected to the current source. 3 nP and transistors, A fourth NPI having a base connected to the output of the differential amplifier, a collector connected to the power supply through the second resistor, a second connected to the collector of the second ENP transistor, and an emitter connected to the current source. Transistors, And a fifth resistor connected in series between the emitter of the third and fourth ENP transistors. In claim 8, The transimpedance amplifier, A third PNP transistor in which an emitter is connected to a contact point between the collectors of the first and third NPI transistors and the second resistor, and a base is connected to a bias power supply; A fourth PNP transistor in which an emitter is connected to a contact point between the collectors of the second and fourth NPI transistors and the third resistor, and a base is connected to the bias power source; A fifth NPN transistor in which a collector is connected to the collector of the third PN transistor, a base is connected to the collector of the fourth PN transistor, and an emitter is grounded; A sixth NPI transistor having a collector and a base connected to the collector of the fourth PNP transistor, and having an emitter grounded; A seventh NPI transistor in which a base is connected to a contact point of the collector of the fifth ENP transistor and the third PNP transistor, a collector is connected to the power supply, and an emitter is grounded through a constant resistance; An eighth NPI transistor having a collector and a base connected to the current source, A ninth NPI transistor having a base connected to the current source and a collector connected to the power source; A fifth PNP transistor in which an emitter is connected to an emitter of the eighth NPN transistor, and a base and a collector are connected to each other; A sixth PNP transistor in which an emitter is connected to an emitter of the ninth NPN transistor, a base is connected to a base of the fifth PN transistor, and a collector is grounded; A capacitor connected at one side to a base of the seventh NPN transistor, and at the other side to a contact point of an emitter of the eighth NPN transistor and an emitter of the fifth PN transistor; A transconductor including a tenth nPN transistor having a base connected to an emitter of the seventh NPN transistor and a contact of the predetermined resistance, a collector connected to a collector of the fifth PN transistor, and having an emitter grounded . In claim 2, The current output unit, A first PNP transistor whose emitter is connected to a power source, A second PNP transistor having an emitter connected to the power supply, a base connected to a collector, and a base connected to the base of the first PNP transistor, A first NPI transistor having a collector connected to a collector of the second PNP transistor, A second NPI transistor having a collector connected to the collector of the first PNP transistor and having an emitter grounded; A third PNP transistor having an emitter connected to an emitter of the first NP transistor, and having a base connected to a base of the first NP transistor; A transconductor comprising a third NPP transistor connected to a base and a collector, a base connected to a base of the second NPP transistor, an emitter grounded, and a collector connected to a collector of the third PNP transistor. . In claim 10, The reference voltage amplifier receives the reference voltage through a non-inverting input terminal, and an inverting terminal is connected to a contact point of an emitter of the first NPP transistor and an emitter of the third PNP transistor, and an output terminal of the reference voltage amplifier is used. And a base connected to the contact of the base of the N transistor and the base of the third PNP transistor. A transconductor receiving the first and second control voltages from an external servo and outputting a current proportional to a difference between the first and second control voltages using half of the power supply voltage as a reference voltage; An absolute value circuit unit which takes an absolute value on the current output from the transconductor, A current / voltage converter converting an output current of the absolute value circuit unit into a voltage and outputting the converted voltage; A driver which receives an output of the current / voltage converter to a non-inverting end and receives a feedback signal to an inverting end, and generates an on / off signal so that the rotation of the motor generates maximum torque; A power transistor unit which is turned on / off according to the output of the driver to apply a three-phase output current to the motor; An output current sensing unit which senses a three-phase current output from the power transistor unit and converts it into a voltage and supplies it to the driver as the feedback signal To Including motor drive system. In claim 12, The transconductor, A differential amplifier which receives a first control voltage and a second control voltage, shifts the reference voltage to a value obtained by amplifying a difference between the first control voltage and the second control voltage using half of a power supply voltage as a reference voltage; A reference voltage amplifier receiving the reference voltage and outputting a voltage corresponding to the input voltage; A first resistor connected in series between the reference voltage amplifier and the differential amplifier; A current output unit for outputting a current flowing through the first resistor irrespective of the reference voltage Motor drive system comprising a.