JPWO2018225250A1 - Radar equipment - Google Patents

Abstract

The radar apparatus (1) includes a plurality of RF receivers (8-1 to 8-Q) that receive a plurality of pulse-modulated waves reflected from a target and output a plurality of reception signals, respectively, and a plurality of reference signals. A signal processing circuit (9-1 to 9-Q) that generates a plurality of pulse compression signals by the used correlation processing and generates a band synthesis signal by synthesizing the pulse compression signals, and the band synthesis signal And a target candidate detecting unit (13) for detecting a target candidate based on the target candidate. Each of the signal processing circuits (9-1 to 9-Q) has a phase compensation amount candidate value prepared in advance for each pulse compression signal under the condition that the target relative speed is within the assumed speed range. A plurality of integral evaluation values for evaluating the signal quality of the band synthesized signal are calculated using a set of and a phase compensation amount corresponding to each pulse compression signal is selected from the set of candidate values based on the integral evaluation values To do. Each of the signal processing circuits (9-1 to 9-Q) can adjust the phase of the pulse compression signal using the phase compensation amount.

Description

  The present invention relates to radar technology, and more particularly to radar technology for detecting a target object using a plurality of transmission antennas.

  As a radar technique using a plurality of transmission antennas, for example, a MIMO radar (Multiple-Input Multiple-Output radar) technique is known. The MIMO radar technology can detect a target (target) by performing signal processing on a plurality of reception signals using a plurality of transmission antennas and a plurality of reception antennas, and can perform ranging, velocity measurement, or angle measurement. Radar technology. Such a MIMO radar technique is disclosed in Non-Patent Document 1 below, for example.

  The MIMO radar technique disclosed in Non-Patent Document 1 includes a plurality of sub-transmitters that simultaneously transmit a plurality of frequency-modulated transmission waves having different transmission frequencies, and reflection at a target. And a plurality of sub-receivers that receive the plurality of transmitted waves. This MIMO radar technique separates a plurality of band signal components having different transmission frequencies from the outputs of the sub-receivers and applies a window function to suppress side lobes caused by cross-correlation between these band signal components. .

XiZeng Dai, Jia Xu, Chumao Ye, and Ying-Ning Peng, “Low-sidelobe HRR profiling based on the FDFM-MIMO radar,” APSAR 2007, 1st Pac.

  In the MIMO radar technique disclosed in Non-Patent Document 1, when there is an influence of a target Doppler frequency, the phases of a plurality of band signal components may be shifted from each other. In this case, these band signal components cannot be coherently synthesized (integrated), and there is a problem that target detection performance deteriorates.

  In view of the above, an object of the present invention is to provide a radar device capable of improving target detection performance even when there is an influence of a target Doppler frequency.

  A radar apparatus according to an aspect of the present invention receives a plurality of pulse modulated waves reflected by a target and receives a received signal when a plurality of pulse modulated waves having different transmission frequencies are simultaneously emitted from a plurality of transmission units. A plurality of correlation units that generate a plurality of pulse compression signals by performing correlation processing on the reception signal using a plurality of reference signals respectively corresponding to the plurality of pulse modulated waves, A phase compensation amount calculation unit for calculating a plurality of phase compensation amounts for each of the plurality of pulse compression signals; and a plurality of phase compensation signals by adjusting phases of the plurality of pulse compression signals using the plurality of phase compensation amounts. And a band synthesizing unit that generates a band synthesized signal by integrating the plurality of phase compensation signals, and detects a target candidate based on the band synthesized signal A target candidate detection unit, and the phase compensation amount calculation unit is configured to provide each pulse compression signal of the plurality of pulse compression signals under a condition that the target relative speed is within a preset assumed speed range. A plurality of integral evaluation values for evaluating the signal quality of the band synthesized signal using a set of phase compensation amount candidate values prepared in advance, and the set of candidate values based on the plurality of integral evaluation values The phase compensation amount corresponding to each pulse compression signal is selected from the above.

  According to the present invention, target detection performance can be improved even when there is an influence of a target Doppler frequency.

It is a figure which shows schematic structure of the radar apparatus which is Embodiment 1 which concerns on this invention. 3 is a block diagram schematically showing a configuration example of an RF transmitter according to the first embodiment. FIG. FIG. 3 is a diagram schematically illustrating a configuration example of an RF receiver according to the first embodiment. 3 is a block diagram schematically showing a configuration example of a signal detection unit according to the first embodiment. FIG. FIG. 2 is a block diagram schematically showing a configuration example of a signal processing circuit according to the first embodiment. It is a block diagram which shows the hardware structural example which implement | achieves the function of a signal processing circuit and a signal processing part. It is a graph which shows the example of the time change of the frequency of a transmission signal. It is the schematic which shows the positional relationship between two transmission antennas and three receiving antennas, and the phase relationship between a transmission RF signal and a reception RF signal. 3 is a flowchart schematically showing an example of an operation procedure of a signal processing circuit. FIG. 3 is a schematic diagram illustrating a configuration example of a correlation unit according to the first embodiment. It is a graph which shows roughly the time change of the frequency of a received video signal. 12A to 12C are diagrams for explaining the correlation processing (pulse compression processing). 13A to 13C are diagrams for explaining the results of two types of band synthesis processing. 14A and 14B are graphs illustrating the relationship between the target relative speed and the integrated evaluation value. 15A and 15B are graphs illustrating the relationship between the target relative speed and the integrated evaluation value. 16A and 16B are graphs illustrating the relationship between the target relative speed and the maximum value distribution of the combination of the integrated evaluation values. 17A and 17B are graphs illustrating the relationship between the target relative speed and the maximum value distribution of the combination of the integrated evaluation values. 18A to 18D are graphs showing examples of various power waveforms.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In addition, the component to which the same code | symbol was attached | subjected in the whole drawing shall have the same structure and the same function.

Embodiment 1 FIG.
FIG. 1 is a diagram showing a schematic configuration of a radar apparatus 1 according to a first embodiment of the present invention. The radar apparatus 1 includes transmission units 1-1 to 1-P including a plurality of transmission antennas 2-1 to 2-P and reception units 6-1 to 6-6 including a plurality of reception antennas 7-1 to 7-Q. Q is a pulse radar that employs a MIMO radar (Multiple-Input Multiple-Output radar) technique and a digital beam forming (DBF, Digital Beam-Forming) technique. Here, the number P of transmission antennas is an integer of 2 or more, and the number Q of reception antennas is an integer of 2 or more. As shown in FIG. 1, the radar apparatus 1 includes transmission units 1-1 to 1-P that simultaneously radiate P transmission RF (high frequency) signals, which are pulse-modulated waves having different transmission frequencies, to an external space. , Receiving units 6-1 to 6-Q that receive the P transmission RF signals reflected by the target Tgt, and a signal processing unit that performs digital signal processing on parallel outputs of these receiving units 6-1 to 6-Q 10.

Each transmission unit 1-p (p is an arbitrary integer from 1 to _P ) outputs a pulse-modulated wave that is frequency-modulated with a modulation bandwidth ΔB _p centered at the frequency F _p (= f ₀ + B _p ). An RF transmitter 3-p and a transmission antenna 2-p formed of an antenna that radiates a transmission RF signal that is the pulse-modulated wave are included. Here, the frequency f ₀ is a reference center transmission frequency, and B _p is a frequency modulation amount previously assigned to the p-th transmission RF signal. The frequency modulation performed by the RF transmitter 3-p is a modulation that continuously increases or decreases the instantaneous frequency with time. In the frequency modulation of the present embodiment, the instantaneous frequency changes linearly. The P transmission antennas 2-1 to 2-P are arranged in a predetermined linear, two-dimensional or three-dimensional array, and constitute a transmission antenna array. FIG. 2 is a block diagram schematically showing a configuration example of the RF transmitter 3-p. The configuration of the RF transmitter 3-p shown in FIG. 2 will be described later.

  On the other hand, referring to FIG. 1, each receiving unit 6-q (q is an arbitrary integer from 1 to Q) includes a receiving antenna 7-q on which a plurality of transmission RF signals reflected by the target Tgt are incident, An RF receiver 8-q that receives the output of the reception antenna 7-q, and band synthesis by performing correlation processing (pulse compression processing) and band synthesis processing on the output of the RF receiver 8-q, that is, the received video signal. And a signal processing circuit 9-q for generating a signal. A superimposed signal in which P transmission RF signals reflected by the target Tgt are superimposed on each other arrives at the receiving antenna 7-q. The Q reception antennas 7-1 to 7-Q are arranged in a predetermined linear, two-dimensional or three-dimensional array, and constitute a reception antenna array.

  FIG. 3 is a diagram schematically illustrating a configuration example of the RF receiver 8-q. The RF receiver 8-q illustrated in FIG. 3 includes a reception antenna 7-q, a signal detection unit 80-q, and an A / D converter 81-q. The signal detection unit 80-q converts the reception RF signal output from the reception antenna 7-q into an analog signal in a frequency band lower than the RF band (for example, an intermediate frequency band). Next, the signal detection unit 80-q detects the phase of the analog signal and generates a detection signal including an in-phase component and a quadrature component. The A / D converter 81-q converts the detected signal into a digital signal, that is, a received video signal. FIG. 4 is a block diagram schematically showing a configuration example of the signal detection unit 80-q. The configuration of the signal detector 80-q shown in FIG. 4 will be described later.

FIG. 5 is a block diagram schematically showing a configuration example of the signal processing circuit 9-q that performs signal processing on the received video signal. The signal processing circuit 9-q shown in FIG. 5 includes a correlation unit 90-q, a phase compensation amount calculation unit 91-q, and a band synthesis unit 92-q. The correlator 90-q performs correlation processing (pulse compression processing) using a reference signal group on the input received video signal V (q, h, m), thereby performing the pulse compression signal R _PC (1, q, h, s) to R _PC (P, q, h, s) are generated. The phase compensation amount calculation unit 91-q calculates the phase compensation amounts Δφ _cor (n _cor , 1) to Δφ _cor (n _cor , P) and the reference phase compensation number N _cor based on the output of the correlation unit 90-q. . The band synthesizing unit 92-q uses the phase compensation amounts Δφ _cor (n _cor , 1) to Δφ _cor (n _cor , P) and the reference phase compensation number N _cor to generate the pulse compression signal R _PC (1, q, h , S) to R _PC (P, q, h, s) by adjusting the phase of the waveform, generating a phase compensation signal group, and synthesizing (integrating) the phase compensation signal group, thereby generating a band synthesis signal R _ΣTx. (N _cor , n _θ , q, h, s) is generated. The band synthesized signal R _ΣTx (n _cor , n _θ , q, h, s) is supplied to the signal processing unit 10 in FIG. The configuration of the signal processing circuit 9-q will be described in detail later.

  Next, referring to FIG. 1, the signal processing unit 10 includes a signal synthesis unit 11, an integration unit 12, a target candidate detection unit 13, and a target information calculation unit 14. The signal synthesis unit 11 synthesizes (integrates) the band synthesis signal group supplied from the reception units 6-1 to 6-Q. The integrating unit 12 generates a low SNR integrated signal by performing coherent integration in the hit direction on the output signal of the signal combining unit 11. The target candidate detection unit 13 can detect one or a plurality of target candidates based on the integration signal. The target information calculation unit 14 can calculate target information such as the arrival angle, relative speed, and relative distance of the input target candidate. The display unit 15 includes an information processing unit that converts the target information into image information, and a display that displays the image information. As the display, for example, a liquid crystal display or an organic EL display can be used. An audio output unit that converts target information into an acoustic signal and outputs it may be used together with the display unit 15.

  The hardware configuration of the above-described signal processing circuits 9-1 to 9-Q and the signal processing unit 10 is, for example, an LSI (Large Scale Integrate Integrated) (ASIC) such as an Application Specific Integrated Circuit (ASIC) or a Field-Programmable Gate Array (FPGA). It only has to be realized.

  FIG. 6 is a block diagram illustrating a hardware configuration example for realizing the functions of the signal processing circuits 9-1 to 9-Q and the signal processing unit 10. The signal processing device 40 shown in FIG. 6 includes a processor 41 configured by LSI, an input interface unit 43, a memory 42, a memory interface unit 44, a display interface unit 45, and a signal path 46. The signal path 46 is a bus for connecting the processor 41, the memory 42, the input interface unit 43, the memory interface unit 44, and the display interface unit 45 to each other. The input interface unit 43 has a function of transferring the received video signal group input from the RF receivers 8-1 to 8 -Q to the processor 41 via the signal path 46. The processor 41 is connected to the display unit 15 via the display interface unit 45.

  The memory 42 is, for example, a program memory that stores various programs for realizing the functions of the radar apparatus 1 of the present embodiment, a work memory that is used when the processor 41 executes signal processing, and the signal processing. Including temporary storage memory in which data used in is expanded. As the memory 42, a plurality of semiconductor memories such as a ROM (Read Only Memory) and an SDRAM (Synchronous Dynamic Random Access Memory) may be used.

  The processor 41 can access the external storage device 47 via the memory interface unit 44. The external storage device 47 is used for storing various data such as setting data and signal data for the processor 41. As the external storage device 47, for example, a volatile memory such as SDRAM, an HDD (Hard Disk Drive) or an SSD (Solid State Drive) can be used. Note that data to be stored in the memory 42 may be accumulated in the external storage device 47.

  In the example of FIG. 6, the signal processing circuits 9-1 to 9-Q and the signal processing unit 10 are realized using a single processor 41, but are not limited thereto. The signal processing circuits 9-1 to 9-Q and the signal processing unit 10 may be realized using a plurality of processors operating in cooperation with each other. Furthermore, any of the signal processing circuits 9-1 to 9-Q and the signal processing unit 10 may be configured by dedicated hardware.

Hereinafter, the configuration and operation of the radar apparatus 1 of the present embodiment will be described in detail.
First, the configuration of the RF transmitter 3-p shown in FIG. 2 will be described in detail. As shown in FIG. 2, the RF transmitter 3-p includes a local oscillator 30-p that outputs a local oscillation signal L ₀ (t) (t is time) in a high frequency band, and a local oscillation signal L ₀ (t). Modulation unit 31-p for generating a pulse signal L _pls (h, t) by performing pulse modulation on the signal, and a modulation signal generator for outputting an intra-pulse modulation signal L _chp (p, h, t) for frequency modulation The transmission RF signal Tx (p, h, t) is generated by performing frequency modulation on the pulse signal L _pls (h, t) using the 32-p and the intra-pulse modulation signal L _chp (p, h, t). A signal generation unit 33-p.

The local oscillation signal L ₀ (t) can be expressed by the following equation (1).

In Equation (1), A _L is the amplitude of the local oscillation signal L ₀ (t), φ ₀ is the initial phase of the local oscillation signal L ₀ (t), f ₀ is the center transmission frequency, and T _obs is the observation time. is there.

The pulse modulation unit 31-p performs pulse modulation on the local oscillation signal L ₀ (t) based on information indicating a preset pulse repetition period T _pri and pulse width T _0, and outputs a pulse signal L _pls ( h, t) is generated, and this pulse signal L _pls (h, t) is output to the signal generator 33-p. The pulse signal L _pls (h, t) can be expressed by the following equation (2).

In Expression (2), h is a pulse hit number that is continuous in time (hereinafter referred to as “hit number”), and h is an integer value in the range of 0, 1, 2,..., H−1. Shall be taken. H is the number of hits defined by the following equation (3).

  In Expression (3), floor (x) is an integer obtained by rounding down the numerical value after the decimal point of the variable x.

The modulation signal generator 32-p uses an intra-pulse modulation signal L _chp (used to frequency-modulate the pulse signal L _pls (h, t) using the frequency modulation amount B _p and the modulation bandwidth ΔB _p. p, t) is generated, and this intra-pulse modulation signal L _chp (p, t) is output to the signal generator 33-p. The intra-pulse modulation signal L _chp (p, h, t) can be expressed by the following equation (4).

FIG. 7 is a graph showing an example of a temporal change in the signal frequency in the transmission pulse. In the example of FIG. 7, the frequency modulation amounts (frequency differences) B ₁ , B ₂ , and B ₃ of the transmission units 1-1, 1-2, and _1-3 are B ₁ = −ΔB, B ₂ = 0, And B ₃ = + ΔB. The modulation bandwidths ΔB ₁ , ΔB ₂ , and ΔB ₃ are the same value (= ΔB). The overall modulation bandwidth B ₀ of the transmitters 1-1, 1-2, 1-3 is 3 × ΔB.

The signal generation unit 33-p in FIG. 2 uses the pulse signal L _pls (t) and the intra-pulse modulation signal L _chp (p, h, t) according to the following equation (5) to transmit the RF signal Tx (p, h , T) and outputs this transmission RF signal Tx (p, h, t) to the transmission antenna 2-p. Here, A _Tx is the amplitude of the transmission RF signal Tx (p, h, t).

Next, the operation of the receiving unit 6-q will be described. The transmission RF signals Tx (1, h, t) to Tx (P, h, t) radiated from the transmission antennas 2-1 to 2-P to the external space are reflected by the target Tgt and then received by the respective reception antennas 7−. Arrives at q. Each reception antenna 7-q outputs a reception RF signal Rx (q, h, t) expressed by the following equation (6) to the RF receiver 8-q.

In Expression (6), Rx ₀ (p, q, h, t) is radiated from the p-th transmitting antenna 2-p and then reflected by the target Tgt to reach the q-th receiving antenna 7-q. RF signal component. The received RF signal Rx (q, h, t) is a signal in which P RF signal components Rx ₀ (1, q, h, t) to Rx ₀ (P, q, h, t) are superimposed on each other. . The RF signal component Rx ₀ (p, q, h, t) can be expressed by the following equation (7).

In Equation (7), A _R is the amplitude of the RF signal reflected at the target Tgt, R ₀ is the initial target relative distance, v is the target relative speed, θ is the target angle, c is the speed of light, and t ′ is within one pulse. Is the time. Φ _Tx (p) is a reference transmission signal (for example, transmission RF signal Tx (1, h, t) of the transmission unit 1-1) and transmission RF signal Tx (p, h, t) of the transmission unit 1-p. , _Rx (p, q) is a transmission RF signal Tx (p, h, t) of the transmission unit 1-p and a reception RF signal Rx (p, q) received by the reception unit 6-q. q, h, t). The phase differences φ _Tx (p), φ _Rx (p, q) can be expressed by the following equations (8) and (9), for example.

In equations (8) and (9), d _Tx (p) is the distance between the reference point (reference position) and the transmitting antenna 2-p, and d _Tx (p) is the reference point and the receiving antenna 7-q. Is the interval. FIG. 8 shows an example of the positional relationship between two (P = 2) transmission antennas 2-1 and 2-2 and three (Q = 3) reception antennas 7-1, 7-2, and 7-3. FIG. As illustrated in FIG. 8, there is a path length difference d _Tx (2) sin θ between the first transmission antenna 2-1 and the second transmission antenna 2-2 arranged at the reference point, A path length difference d _Tx (3) sin θ exists between the first transmission antenna 2-1 and the third transmission antenna 2-3. Phase differences φ _Tx (2) and φ _Tx (3) corresponding to these path length differences d _Tx (2) sin θ and d _Tx (3) sin θ are generated. Further, there is a path length difference d _Rx (1) sin θ between the reference point (transmitting antenna 2-1) and the first receiving antenna 7-1, and the reference point and the second receiving antenna 7-2. There is a path length difference d _Rx (2) sin θ. A phase difference φ _Rx (1,1), φ _Rx (2,1), φ _Rx (3,1) corresponding to the path length difference d _Rx (1) sin θ is generated, and the path length difference d _Rx (2) sin θ. Phase differences φ _Rx (1,2), φ _Rx (2,2), φ _Rx (3,2) corresponding to

The signal detection unit 80-q in FIG. 3 has a configuration for executing phase detection processing. As shown in FIG. 4, the signal detector 80-q includes a local oscillator 800-q, a down converter (mixer) 801-q, a bandpass filter 802-q, an amplifier 803-q, and a phase detector 804-q. It is configured with. The down converter 801-q uses the local oscillation signal L ₀ (t) supplied from the local oscillator 800-q, and uses the frequency of the reception RF signal Rx (q, h, t) input from the reception antenna 7-q. Downconvert the bandwidth. That is, the down converter 801-q converts the received RF signal Rx (q, h, t) into an analog signal in a frequency band (for example, an intermediate frequency band) lower than the RF band. The band pass filter 802-q filters the analog signal and outputs a filter signal. The amplifier 803-q amplifies the filter signal and outputs an amplified signal. Then, the phase detector 804-q detects the phase of the amplified signal and generates a detection signal composed of an in-phase component and a quadrature component, that is, an analog received video signal W (q, h, t).

The analog received video signal W (q, h, t) can be expressed by the following equation (10).

In equation (10), a symbol with a superscript “*” indicates a complex conjugate. W ₀ (p, q, h, t) is a signal component generated from the RF signal component Rx ₀ (p, q, h, t). The signal component W ₀ (p, q, h, t) can be expressed by the following equation (11).

In Expression (11), _AV is the amplitude of the received video signal component W ₀ (p, q, h, t).

The A / D converter 81-q samples the analog reception video signal W (q, h, t) to generate a digital reception video signal V (q, h, m). The received video signal V (q, h, m) can be expressed by the following equation (12).

In Expression (12), V ₀ (p, q, h, m) is a digital signal component generated from the received video signal component Rx ₀ (p, q, h, t). M is a sampling number in the pulse repetition interval (PRI), and M is the number of samplings in the PRI. The digital signal component V ₀ (p, q, h, m) can be expressed by the following equation (13). Δt is a sampling interval.

  Next, the operation of the signal processing circuit 9-q in the reception unit 6-q will be described below with reference to FIGS. FIG. 9 is a flowchart schematically showing an example of the operation procedure of the signal processing circuit 9-q.

The correlation unit 90-q is separated for each transmission frequency by executing correlation processing (pulse compression processing) using a reference signal group on the input received video signal V (q, h, m). Pulse compression signals R _PC (1, q, h, s) to R _PC (P, q, h, s) are generated (step ST21). In other words, the correlation unit 90-q is input received video signal V (q, h, m) from the pulse compressed signal _R PC (1 corresponding respectively to the modulation bandwidth _{ΔB 1 ~ΔB P, q, h} , s) to R _PC (P, q, h, s) are separated.

  FIG. 10 is a schematic diagram illustrating a configuration example of the correlation unit 90-q. As shown in FIG. 10, the correlator 90-q uses the reference signals Ex (1, m) to Ex (P, m) for the input received video signal V (q, h, m). The P compression units 90-q-1 to 90-q-P that execute the P correlation processes (pulse compression processes) in parallel are provided.

Each pulse compression unit 90-q-p (p is an arbitrary integer from 1 to P) includes an intra-PRI Fourier transform unit 901, a mixer 902, an intra-PRI inverse Fourier transform unit 903, and a reference signal supply unit 904-p. A Fourier transform unit 905 and a complex conjugate calculation unit 906. Reference signal supply unit 904-p, the reference signal to generate an Ex (p, m) based on the transmission unit amount of frequency modulation used in 1-p (frequency difference) _{B p} and modulation bandwidth .DELTA.B _p. Each pulse compression unit 90-q-p performs a correlation process between the received video signal V (p, h, m) and the reference signal Ex (p, m), that is, a pulse compression process, thereby performing the pulse compression signal R. _PC (p, q, h, m) can be generated. The reference signal Ex (p, m) can be expressed by the following equation (14).

In Equation (14), A _E is the amplitude of the reference signal Ex (p, m).
The Fourier transform unit 905 performs a fast Fourier transform (FFT) on the reference signal Ex (p, m) according to the following equation (15) to calculate a frequency domain signal F _EX (p, k _r ).

In the formula (15), _{k r} is the sampling _{number, M fft} in PRI shows FFT points of the correlation process. The complex conjugate calculation unit 906 calculates a complex conjugate signal F _EX ^* (p, k _r ) corresponding to the frequency domain signal F _EX (p, k _r ).

On the other hand, the intra-PRI Fourier transform section 901 calculates the frequency domain signal F _V (q, h, k _r ) by performing FFT on the received video signal V (q, h, m) according to the following equation (16).

The mixer 902 uses the frequency domain signal F _V (q, h, k _r ) calculated by the Fourier transform unit 901 in the PRI and the complex conjugate signal F _EX ^* (p, k _r ) according to the following equation (17). Mix (multiply).

PRI in inverse Fourier transform unit 903, according to the following equation (18), the output _F V · Ex mixer 902 (p, q, h, k r) inverse fast Fourier transform to (IFFT: Inverse Fast Fourier Transform) Can be used to calculate the time domain pulse compression signal R _PC (p, q, h, s).

FIG. 11 is a graph schematically showing temporal changes in the frequency of the received video signal. In the example of FIG. 11, the received video signals V (3, h, m), V (2, h, m), V (1, h, m) are transmitted from the transmission units 1-3, 1-2, 1-1. According to the predetermined frequency modulation method used in the above, the frequency is modulated so as to increase almost linearly with time as indicated by a solid line. That is, the received video signal V (3, h, t) is modulated within the modulation bandwidth ΔB ₃ centered on the frequency f ₀ + B ₃ , and the received video signal V (2, h, t) has the frequency f _0. + _{B 2} to be modulated in the modulation bandwidth .DELTA.B ₂ around, the received video signal V (1, h, t) is modulated with a modulation bandwidth .DELTA.B within ₁ around the frequency _f 0 + _{B 1.}

12A to 12C are diagrams for explaining the correlation processing (pulse compression processing). 12A to 12C show graphs showing temporal changes in the frequencies of the reference signals Ex (3, m), Ex (2, m), and Ex (1, m). In addition, a graph of the power waveform of the pulse compression signal obtained by the correlation processing using these reference signals Ex (3, m), Ex (2, m), Ex (1, m) (the horizontal axis represents time, vertical The axis indicates power.) Is also shown. When the correlation process between the received video signal V (3, h, m) and the reference signal Ex (3, m) is executed, as shown in FIG. 12A, after the correlation, the power waveform compressed in a pulse shape is used. The pulse compression signal R _PC (3, q, h, s) having Similarly, when correlation processing between the received video signal V (2, h, m) and the reference signal Ex (2, m) is executed, the pulse compression signal R _PC (having a power waveform as shown in FIG. 2, q, h, s) is obtained, and when the correlation process between the received video signal V (1, h, m) and the reference signal Ex (1, m) is executed, the power as shown in FIG. A pulse compression signal R _PC (1, q, h, s) having a waveform is obtained. Pulse compression signals R _PC (3, q, h, s), R _PC (2, q, h, s), R _PC (1, q, h, s) having the power waveforms shown in FIGS. 12A to 12C. Are substantially the same, the pulse compression signals R _PC (1, q, h, s) to R _PC (3, q, h, s) can be coherently integrated (band synthesis).

However, when there is an influence of the Doppler frequency of the relatively moving target Tgt, the frequencies of the received video signals V (1, h, m) to V (3, h, m) are as shown by the dotted lines in FIG. Each may deviate from the appropriate range. In such a case, the prior art has a problem that if the phases of the pulse compression signals are shifted to a degree that cannot be ignored, these pulse compression signals cannot be coherently integrated (band synthesis). 13A to 13C are diagrams for explaining the results of two types of band synthesis processing. As shown in FIG. 13A, when the pulse compression signal R _PC (p, q, h, s) is integrated (band synthesis), if there is no influence of the Doppler frequency, coherent integration is possible. A composite power waveform having a sharp peak as shown in 13B is generated. However, when there is an influence of the Doppler frequency, coherent integration is not always possible, and a distorted combined power waveform as shown in FIG. 13C may be generated. The phase compensation amount calculating unit 91-q and the band synthesizing unit 92-q in the signal processing circuit 9-q of the present embodiment compensate for the phase shift between the pulse compression signals when there is an influence of such a Doppler frequency. can do.

Next, prior to detailed description of the phase compensation amount calculation unit 91-q and the band synthesis unit 92-q, the principle that coherent integration is difficult when there is an influence of the Doppler frequency will be described. In order to explain this principle, a reference signal Ex (p, m, m _τ ) having the same frequency as the transmission frequency is expressed by the following equation (19). Here, A _E is the amplitude, and m _τ Δt is an arbitrary time shift amount.

Time mΔt is, m _tau when there _Δt-T 0 / 2~m _τ in the range of Δt _{+ T} 0/2, the received video signal component _{V 0 (p, q, h} , m) and the reference signal Ex (p, m, Correlation processing in the time domain with m _τ ) is performed according to the following equation (20).

R _{V0 · Ex} (p, q, h, m _τ ) is a pulse compression signal separated for each transmission frequency. V ₀ (p, q, h, m) E ^* (p, m, m _τ ) appearing on the right side of the equation (20) can be expanded as shown in the following equation (21).

In equation (21), A is the amplitude, h is the hit number, and H is the number of hits. From the above expressions (20) and (21), the pulse compression signal R _{V0 · Ex} (p, q, h, m _τ ) can be expressed by the following approximate expression (22).

According to the above equation (22), when the following relational equation (23) holds for an arbitrary time shift amount m _τ Δt, the absolute value of the pulse compression signal R _{V0 · Ex} (p, q, h, m _τ ) It can be seen that the value can take the maximum value.

If this equation (23) is modified, an arbitrary time shift amount m _τ Δt can be obtained as shown in the following equation (24).

Therefore, when the relative speed v is non-zero (v ≠ 0), that is, when there is an influence of the Doppler frequency, the pulse compression signal R _{V0 · is obtained at} a time corresponding to the time shift amount m _τ Δt shown in the above equation (24). The absolute value of _Ex (p, q, h, m _τ ) can take the maximum value.

Here, a phase difference Δφ _PC (p, v) is defined as shown in the following equation (25).

When the above equation (23) is established and the phase difference Δφ _PC (p, v) is defined as in the equation (25), the approximate equation of the above equation (22) is expressed by the following equation (26). The

The phase difference Δφ _PC (p, v) when the absolute value of the pulse compression signal R _{V0 · Ex} (p, q, h, m _τ ) shows the maximum value is expressed by the following equations (24) and (25). It is derived as shown in Equation (27).

The approximate expression of Expression (27) is when the center transmission frequency f ₀ is sufficiently larger than the frequency modulation amount B _p and the center transmission frequency f ₀ is sufficiently larger than the modulation bandwidth ΔB _p (f ₀ >> B). _p and an approximate expression that holds for f ₀ >> ΔB _p ). In the approximate expression of formula (27), the phase difference [Delta] [phi _PC (p, v) is proportional to the target relative speed v and the frequency modulation amount _{B p,} and inversely proportional to the modulation bandwidth .DELTA.B _p.

According to the equation (27), when the target relative speed v is non-zero (v ≠ 0), that is, when the Doppler frequency is affected, the pulse compression signal R _{V0 · Ex} (p, q, h, m _τ ) Has a maximum value, a non-zero phase difference Δφ _PC (p, v) depending on the target relative speed v is generated, which may prevent coherent integration (band synthesis). I understand that there is. In this case, an integral loss that cannot be ignored occurs.

If accurate information of the target relative speed v can be obtained in advance before band synthesis, it is possible to easily calculate the phase compensation amount for compensating for the phase difference Δφ _PC (p, v). However, it is difficult to obtain accurate information on the target relative speed v in advance.

Therefore, the phase compensation amount calculation unit 91-q according to the present embodiment has the condition that the target relative speed v is within a predetermined assumed speed range (a range from the minimum speed _vmin to the maximum speed _vmax ). A reference phase compensation number N _cor and a phase compensation amount Δφ _cor (n _cor , p) satisfying a low loss condition in which the integrated loss generated during band synthesis within the predetermined speed range is within a certain range are calculated (FIG. 9). Step ST22). Here, the reference phase compensation number N _cor is an integer representing the number of phase compensation amounts Δφ _cor (n _cor , p) for each value of p. When N _cor = 1, only one phase compensation amount Δφ _cor (1, p) is calculated (n _cor = 1). When the reference phase compensation number N _cor is 2 or more, a plurality of phase compensation amounts Δφ _cor (1, p),..., Δφ _cor (N _cor , p) are calculated (n _cor = 1,..., N _cor ).

Hereinafter, a method of calculating the reference phase compensation number N _cor and the phase compensation amount Δφ _cor (n _cor , p) will be described.
The phase compensation amount calculation unit 91-q preliminarily calculates the value of each p under the condition that the target relative speed v is within a predetermined assumed speed range (range from the minimum speed _vmin to the maximum speed _vmax ). Using the set of phase compensation amount candidate values {Δφ _cor (n _cor , p)} (n _cor = 1, 2,...), The signal quality of the band synthesized signal generated by the band synthesizing unit 92-q. Integral evaluation value L _cor (n _cor , v) (n _cor = 1, 2,...) _Is calculated. The phase compensation amount calculation unit 91-q then sets a set of candidate values {Δφ _cor (n _cor , p)} (n _cor = 1, 2,...) Based on these integral evaluation values L _cor (n _cor , v). ), Phase compensation amounts Δφ _cor (1, p) to Δφ _cor (N _cor , p) satisfying the low loss condition are selected.
Here, the set of candidate values {Δφ _cor (n _cor , p)} (n _cor = 1, 2,...) Is a phase compensation amount Δφ _cor (n in a state where the reference phase compensation number N _cor is not determined. _cor , p). By determining the reference phase compensation number N _cor , phase compensation amounts Δφ _cor (1, p) to Δφ _cor (N _cor , p) satisfying the low loss condition are determined.

The integral evaluation value L _cor (n _cor , v) can be expressed by the following equation (28), for example. Here, | X | is the absolute value of the variable X.

Δφ _PC (p, v) appearing on the right side of the equation (28) is a phase difference expressed by the above-described approximate equation (27). Δφ _cor (n _cor , p) appearing on the right side of the equation (28) is a candidate value of the phase compensation amount added to cancel the phase difference Δφ _PC (p, v). According to Expression (28), when integrated without loss (when the signal to be integrated is coherent, that is, in phase), the phase difference Δφ _PC (p, v) and the candidate value Δφ _cor (n _cor , p) The difference Δφ _PC (p, v) −Δφ _cor (n _cor , p) between the two values is zero, and thus the integral evaluation value L _cor (n _cor , v) has the maximum value “1”. On the other hand, when the integral loss is maximum (when the signals to be integrated are orthogonal and cancel each other), it can be seen that the integral evaluation value L _cor (n _cor , v) can be the minimum value “0”.

Here, the integral evaluation value L _cor (n _cor , v) is not limited to the integral evaluation value expressed by the above equation (28). If the expression is configured based on the difference Δφ _PC (p, v) −Δφ _cor (n _cor , p), the integral evaluation value L _cor (n _cor , v) is defined by an expression other than the expression (28). It is possible.
The phase compensation amount calculation unit 91-q predicts the phase difference Δφ _PC (p, v) of each pulse compression signal in the assumed speed range according to the approximate expression (27), and the predicted phase difference Δφ _PC (p, v). ) And the candidate value Δφ _cor (n _cor , p), the integrated evaluation value L _cor (n _cor , v) is calculated based on the difference Δφ _PC (p, v) −Δφ _cor (n _cor , p). That's fine.

The phase compensation amount candidate value Δφ _cor (n _cor , p) is defined, for example, by the following equation (29).

Δφ _cor (n _cor , p) in the equation (29) is a reference phase compensation amount when the reference phase compensation number N _cor is given, and is defined by the following equation (30).

Here, the phase compensation amount candidates Δφ _cor (n _cor , p) obtained by the equations (29) and (30) have the modulation bandwidth ΔB _p all set to the same value ΔB as shown in the following equation (31). An amount defined when set.

For example, when the number of transmission antennas is three (P = 3), the frequency modulation amounts B _{1 to} B ₃ are expressed by the following equation (32) using the frequency modulation amount (frequency difference) B ₂ as a reference. Can be set to

As described above, by determining the reference phase compensation number N _cor , phase compensation amounts Δφ _cor (1, p) to Δφ _cor (N _cor , p) satisfying the low loss condition are determined. The phase compensation amount calculation unit 91-q calculates an integral evaluation value L _cor (n _cor , v) that can be configured from a set of candidate values {Δφ _cor (n _cor , p)} (n _cor = 1, 2,...). From the combinations, a combination that satisfies the low loss condition that the maximum value distribution L _{cor, max} (v) of the combination always exceeds the threshold value _LΣ within the assumed speed range is searched. Then, the phase compensation amount calculation unit 91-q selects a candidate value forming a combination obtained as a result of the search as a phase compensation amount that satisfies the low loss condition.

Specifically, the maximum value distribution L _{cor, max} (v) of the combination of integral evaluation values is based on the combination of the integral evaluation values L _cor (1, v) to L _cor (N _cor , v) as follows: (33) (N _cor is a provisional value).

In Expression (33), the maximum value distribution L _{cor, max} (v) is an integral evaluation value L _cor (1, v) with respect to the value of the target relative speed v within the assumed speed range (v _{min to} v _max ). ,..., L _cor (N _cor , v) is the maximum integrated evaluation value selected. Here, it should be noted that the selected integral evaluation value may differ depending on the value of the target relative speed v.

The phase compensation amount calculation unit 91-q has a maximum value distribution L _{cor, max} (v as shown in the following equation (34) over the entire assumed speed range (v _{min to} v _max ) of the target relative speed v. ) _{Exceeding the} threshold value _LΣ , the smallest integer of the set of integers {N _cor } that always satisfies the low loss condition can be selected (calculated) as the value of the reference phase compensation number N _cor .

As an example of the processing procedure for determining the definite value of the reference phase compensation number N _cor , for example, the procedure described below can be given.
That is, the phase compensation amount calculation unit 91-q sets the provisional value (temporary value) of the reference phase compensation number N _cor and the above equations (28), (29), (30 ) according to integral evaluation value _{L cor (1, v) ~L} cor (N cor, v and step B of calculating a combination of) these integral evaluation value _{L cor (1, v) ~L} cor (N cor, v ) Step C for calculating the maximum value distribution L _{cor, max} (v) using the expression (33) based on the combination of the above and the maximum value distribution L _{cor, max} (v) within the assumed speed range, the conditional expression (34 ) Can always be executed sequentially. In step D, when the maximum value distribution L _{cor, max} (v) always satisfies the conditional expression (34), the phase compensation amount calculation unit 91-q determines the provisional value as the reference phase compensation number N _cor . Select as value (step E). On the other hand, when it is determined in step D that the maximum value distribution L _{cor, max} (v) does not always satisfy the conditional expression (34), the phase compensation amount calculation unit 91-q increases the provisional value. What is necessary is just to change to an integer value and to perform step A-step D again.

14A and 14B are graphs illustrating the relationship between the relative speed v of the target Tgt and the integral evaluation value when the reference phase compensation number N _cor = 1, 2 (provisional value). 15A and 15B are graphs illustrating the relationship between the relative speed v of the target Tgt and the integral evaluation value when the reference phase compensation number N _cor = 3, 4 (provisional value). In FIG. 14A, FIG. 14B, FIG. 15A and FIG. 15B, the solid line indicates the distribution of the integral evaluation value L _cor (1, v) when the reference phase compensation number n _cor = 1, and the broken line indicates the reference phase compensation number n _cor = 2. The integrated evaluation value L _cor (2, v) in the case of (1), the one-dot chain line shows the distribution of the integrated evaluation value L _cor (3, v) in the case of the reference phase compensation number n _cor = 3, the two-dot chain line shows The distribution of the integral evaluation value L _cor (4, v) when the reference phase compensation number n _cor = 4 is shown.

FIGS. 16A and 16B are diagrams illustrating the maximum distribution L _{cor, max} (v) of the integrated evaluation value when the reference phase compensation number N _cor = 1, 2 (provisional value). FIGS. FIG. 10 is a diagram showing a maximum value distribution L _{cor, max} (v) of an integral evaluation value when the reference phase compensation number N _cor = 3, 4 (provisional value). As shown in FIGS. 16A, 16B, and 17A, in the case of the reference phase compensation number N _cor = 1, 2, 3, the maximum value distribution L _{cor, max} (v) is an estimated speed range of the target relative speed v ( v _{since min} to v _max) not always to exceed the threshold value L _sigma in, it is estimated that the integration loss exceeding the allowable range occurs. On the other hand, when the reference phase compensation number N _cor = 4 as shown in FIG. 17B, the maximum value distribution L _{cor, max} (v) is always in the assumed speed range (v _{min to} v _max ) of the target relative speed v. Since the threshold value _LΣ is exceeded, the maximum value distribution L _{cor, max} (v) is estimated to generate only an integral loss within an allowable range. Therefore, the phase compensation amount calculation unit 91-q sets the definite value of the reference phase compensation number N _cor to 4.

As described above, when the definite value of the reference phase compensation number N _cor is determined, the phase compensation amount calculation unit 91-q sets the set of phase compensation amount candidate values {Δφ _cor (n _cor , p)} ( phase compensation amounts selected from n _cor = 1, 2,... Δφ _cor (n _cor , 1) to Δφ _cor (n _cor , P) (n _cor = N _cor = 1, or n _cor = 1, .., N _cor ) are output to the band combining unit 92-q.

Phase compensation amount calculating section 91-q as described above, the phase compensation amount [Delta] [phi _cor for realizing such integration loss is within a certain range the integrator (band _{synthesis) (n cor, 1) ~Δφ} cor (n _cor , P) can be calculated (selected). For this reason, it is possible to suppress the integral loss in the band synthesizing unit 92-q. The phase compensation amount calculation unit 91-q includes the reference phase compensation number N _cor and the phase compensation amounts Δφ _cor (n _cor , 1) to Δφ _cor (n _cor , P) (n _cor = N _cor = 1, or n _cor = 1,..., N _cor ) are output to the band synthesis unit 92-q.

The band synthesizing unit 92-q performs the reference phase with respect to the pulse compression signal R _PC (p, q, h, s) separated for each transmission frequency for each reference phase compensation number n _cor according to the following equation (35). A phase compensation signal is generated by executing phase compensation processing (phase adjustment processing) using the compensation number N _cor and the phase compensation amount Δφ _cor (n _cor , p) (p = 1 to P), and these phase compensation signals _Is synthesized (integrated) to generate a synthesized band signal R _ΣTx (n _cor , p, q, h, s) (step ST23 in FIG. 9).

In Expression (35), d _Tx (p) is a phase difference in the transmission unit 1-p, θ ′ (n _θ ) is a target arrival angle candidate to which the target arrival angle candidate number n _θ is assigned, and N _θ is a target arrival angle. The number of candidates. According to Expression (35), it can be seen that the phase of the pulse compression signal R _PC (p, q, h, s) is adjusted by the phase compensation amount Δφ _cor (n _cor , p).
The target arrival angle candidate θ ′ (n _θ ) is defined by the following equation (36). Here, Δθ _samp is an assumed target angle interval.

18A to 18D are graphs showing power waveforms when the reference phase compensation number n _cor = 1, 2, 3, 4 is used. The solid line in FIGS. 18A to 18D shows the combined power waveform. As shown in FIG. 18B, when the reference phase compensation number n _cor = 2, as a result of suppressing the integral loss (integral loss is within the setting range), a band synthesized signal having a synthesized power waveform having a sharp peak is obtained. I understand that In the case of FIGS. 18A, 18C, and 18D, the integrated loss is not sufficiently suppressed (the integrated loss exceeds the set range), and a band combined signal having a distorted combined power waveform having a plurality of strong peaks is obtained. I understand that

The band synthesis unit 92-q outputs the band synthesis signal R _ΣTx (n _cor , n _θ , q, h, s) to the signal processing unit 10.

  Next, processing contents in the signal processing unit 10 will be described below.

In the signal processing unit 10, the signal synthesis unit 11 synthesizes (integrates) the band synthesis signal R _ΣTx (n _cor , n _θ , q, h, s) related to the reference phase compensation number n _cor according to the following equation (37). Thus, the combined signal R _ΣTxRx (n _cor , n _θ , h, s) is generated.

  However, when only one receiving antenna among the receiving antennas 7-1 to 7-Q is used (Q = 1), the signal combining unit 11 may not perform combining (integration).

Next, the integration unit 12 performs the coherent integration in the hit direction by Fourier transform on the synthesized signal R _ΣTxRx (n _cor , n _θ , h, s) according to the following equation (38), _{thereby obtaining} the reference phase An integrated signal f _d (n _cor , n _θ , k, s) relating to the compensation number n _cor is generated.

The integration signal f _d (n _cor , n _θ , k, s) is output to the target candidate detection unit 13. Since the integration unit 12 performs coherent integration in the hit direction, the SNR (signal-to-noise ratio) of the integrated signal f _d (n _cor , n _θ , k, s) related to the reference phase compensation number n _cor is improved, and the target Detection performance can be improved. It is also possible to obtain information on the target relative speed. However, when the SNR of the combined signal R _ΣTxRx (n _cor , n _θ , h, s) relating to the reference phase compensation number n _cor is sufficiently good, or when it is not necessary to obtain information on the target relative speed, the integration unit 12 may not be provided.

The target candidate detection unit 13 can detect one or a plurality of target candidates based on the signal strength of the integrated signal f _d (n _cor , n _θ , k, s) related to the reference phase compensation number n _cor . More specifically, for example, based on the well-known CA-CFAR (Cell Average Constant False Alarm Rate) process, the target candidate detection unit 13 may detect a target candidate. The target candidate detection unit 13 outputs the detected arrival angle candidate number n ′ _θ , velocity bin number h ′ _tgt , and sampling number s ′ _tgt in the distance direction to the target information calculation unit 14.

The target information calculation unit 14, for example, in accordance with the following equations (39), (40), (41), the input candidate angle of arrival candidate number n ′ _θ , the speed bin number h ′ _tgt, and the sampling number in the distance direction Based on s ′ _tgt , the target candidate arrival angle θ ′ _tgt , the target candidate relative speed v ′ _tgt , and the target candidate relative distance R ′ _tgt can be calculated.

In Expression (41), Δr _IFFT is a sampling interval in the distance direction after correlation. The target information calculation unit 14 displays target information including a target candidate arrival angle θ ′ (n ′ _θ ) corresponding to the arrival angle candidate number n ′ _θ , a target candidate relative speed v ′ _tgt, and a target candidate relative distance R ′ _tgt. To the unit 15. Here, the target information calculation unit 14 may supply image information that visually represents the target information to the display unit 15. The display unit 15 displays image information representing the target information supplied from the target information calculation unit 14 on the display screen.

As described above, in the radar apparatus 1 according to the present embodiment, the phase compensation amount calculation unit 91-q compensates for the influence of the Doppler frequency of the moving target that moves relative to the radar apparatus 1. , Phase compensation amounts Δφ _cor (n _cor , 1) to Δφ _cor (n _cor , P) are calculated. The band synthesizing unit 92-q executes a phase compensation process and a band synthesizing process using the reference phase compensation number N _cor and the phase compensation amounts Δφ _cor (n _cor , 1) to Δφ _cor (n _cor , P). For this reason, the integral loss in the band composition unit 92-q can be suppressed. Therefore, even when there is an influence of the Doppler frequency, it is possible to perform the band synthesis in which the integral loss is suppressed and to provide the radar apparatus 1 with improved target detection performance.

  Further, since the signal synthesis unit 11 is provided, the radar apparatus 1 with further improved target detection performance can be provided. In addition, since the integration unit 12 performs the coherent integration in the hit direction, an integration signal with a good SNR can be calculated, thereby further improving the target detection performance.

  Although various embodiments according to the present invention have been described above with reference to the drawings, these embodiments are examples of the present invention, and various forms other than these embodiments can be adopted. In the above embodiment, a target object is detected by the MIMO radar technique using a plurality of transmission units 1-1 to 1-P and a plurality of reception units 6-1 to 6-Q. Instead, the above-described implementation is performed so that the target object is detected by MISO radar (Multiple-Input Single-Output radar) technology using a plurality of transmission units 1-1 to 1-P and one reception unit 6-q. The configuration of the form may be changed. For example, in the radar apparatus 1 shown in FIG. 1, the signal processing unit 10 is configured such that the signal synthesis unit 11 is not used and only the output of one reception unit 6-q is directly input to the integration unit 12. Can be changed.

  It should be noted that within the scope of the present invention, any combination of the constituent elements of the above-described embodiment, modification of any constituent element of the above-described embodiment, or omission of any constituent element of the above-described embodiment is possible.

  Since the radar apparatus according to the present invention has a configuration capable of improving the target detection performance even when there is an influence of the target Doppler frequency, the moving target moves relatively with respect to the radar apparatus. It is preferable to be used in a radar system that detects Moreover, the radar apparatus according to the present invention can be used in a state where it is installed on the ground or in a state where it is mounted on a moving body such as an aircraft, an artificial satellite, a vehicle or a ship.

  DESCRIPTION OF SYMBOLS 1 Radar apparatus, 1-1 to 1-P transmission unit, 2-1 to 2-P transmission antenna (antenna), 3-1 to 3-P RF transmitter, 30-p local oscillator, 31-p pulse modulation unit , 32-p modulation signal generator, 33-p signal generator, 6-1 to 6-Q receiver, 7-1 to 7-Q reception antenna, 8-1 to 8-Q RF receiver, 80-q Signal detector, 800-q local oscillator, 801-q down converter, 802-q bandpass filter, 803-q amplifier, 804-q phase detector, 81-q A / D converter, 9-1 to 9- Q signal processing circuit, 90-q correlation unit, 90-q-1 to 90-q-P pulse compression unit, 901 Fourier transform unit within PRI, 902 mixer, 903 inverse Fourier transform unit within PRI, 904-1-904 -P reference signal supply unit, 905 full D conversion unit, 906 complex conjugate calculation unit, 91-q phase compensation amount calculation unit, 92-q band synthesis unit, 10 signal processing unit, 11 signal synthesis unit, 12 integration unit, 13 target candidate detection unit, 14 target information calculation Unit, 15 display unit, 40 signal processing device, 41 processor, 42 memory, 43 input interface unit, 44 memory interface unit, 45 display interface unit, 46 signal path, 47 external storage device, Tgt target.

A radar apparatus according to an aspect of the present invention receives a plurality of pulse modulated waves reflected by a target and receives a received signal when a plurality of pulse modulated waves having different transmission frequencies are simultaneously emitted from a plurality of transmission units. an RF receiver for outputting, by performing correlation processing on the received signal using a plurality of reference signals corresponding to the plurality of pulse modulation wave, a correlation unit that generates a plurality of pulse compression signal, A phase compensation amount calculation unit for calculating a plurality of phase compensation amounts for each of the plurality of pulse compression signals; and a plurality of phase compensation signals by adjusting phases of the plurality of pulse compression signals using the plurality of phase compensation amounts. And a band synthesis unit that generates a band synthesized signal by integrating the plurality of phase compensation signals, and a target that detects a target candidate based on the band synthesized signal The phase compensation amount calculation unit for each pulse compression signal of the plurality of pulse compression signals under the condition that the target relative speed is within a preset assumed speed range. Calculating a plurality of integral evaluation values for evaluating the signal quality of the band synthesized signal using a set of candidate values for the phase compensation amount prepared in advance, and based on the plurality of integral evaluation values, A phase compensation amount corresponding to each pulse compression signal is selected from among them.

Claims (12)

An RF receiver for receiving a plurality of pulse modulated waves reflected by a target and outputting a received signal when a plurality of pulse modulated waves having different transmission frequencies are simultaneously radiated from a plurality of transmission units;
A plurality of correlation units that generate a plurality of pulse compressed signals by performing correlation processing on the received signal using a plurality of reference signals respectively corresponding to the plurality of pulse modulated waves;
A phase compensation amount calculating unit for calculating a plurality of phase compensation amounts for each of the plurality of pulse compression signals;
Bands that generate a plurality of phase compensation signals by adjusting the phases of the plurality of pulse compression signals using the plurality of phase compensation amounts, and generate a band synthesized signal by integrating the plurality of phase compensation signals A synthesis unit;
A target candidate detection unit for detecting a target candidate based on the band synthesized signal,
The phase compensation amount calculating unit prepares phase compensation prepared in advance for each pulse compression signal of the plurality of pulse compression signals under a condition that the target relative speed is within a preset assumed speed range. Calculating a plurality of integral evaluation values for evaluating the signal quality of the band synthesized signal using a set of candidate values of the quantity, and compressing each pulse from the set of candidate values based on the plurality of integral evaluation values A radar apparatus, wherein a phase compensation amount corresponding to a signal is selected.   The radar apparatus according to claim 1, wherein the phase compensation amount calculation unit predicts a phase difference between the pulse compression signals in the assumed speed range, and determines between the predicted phase difference and the candidate value. A radar apparatus that calculates the plurality of integral evaluation values based on a difference. The radar apparatus according to claim 2, wherein
The plurality of transmission units generate the plurality of pulse modulated waves using a preset frequency modulation amount and modulation bandwidth,
The radar apparatus according to claim 1, wherein the phase compensation amount calculation unit calculates a value proportional to the target relative speed and the frequency modulation amount and inversely proportional to the modulation bandwidth as the predicted phase difference.   2. The radar apparatus according to claim 1, wherein the phase compensation amount calculation unit is configured such that a maximum value distribution of the combination always exceeds a threshold value within the assumed speed range among the plurality of integration evaluation value combinations. A combination satisfying a condition is searched, and a candidate value forming a combination obtained as a result of the search is selected from the set of candidate values as a phase compensation amount corresponding to each pulse compression signal. Radar device. The radar apparatus according to claim 1,
Further comprising the plurality of transmission units;
The plurality of transmitters perform frequency modulation on a plurality of pulse signals in a high frequency band, respectively, to generate the plurality of pulse modulated waves.   The radar apparatus according to claim 1, further comprising a target information calculation unit that calculates a relative speed, a relative distance, and an arrival angle of the target candidate. When a plurality of pulse modulated waves having different transmission frequencies are simultaneously radiated from a plurality of transmitters, a plurality of RF receivers receive the plurality of pulse modulated waves reflected by the target and output a plurality of received signals, respectively. And
A plurality of reference signals respectively corresponding to the plurality of pulse modulated waves are used to perform correlation processing on each of the plurality of reception signals, thereby generating a plurality of pulse compression signals for each of the plurality of reception signals. A correlation section;
A plurality of phase compensation amount calculation units for calculating a plurality of phase compensation amounts for the plurality of pulse compression signals for each of the plurality of received signals;
Each of the plurality of phase compensation signals is adjusted by using the plurality of phase compensation amounts to adjust the phase of the waveform of the plurality of pulse compression signals, and each of the plurality of phase compensation signals is integrated to generate a band. A plurality of band synthesizers for generating a synthesized signal;
A target candidate detection unit for detecting a target candidate based on the band synthesized signal,
Each of the plurality of phase compensation amount calculation units prepares in advance each pulse compression signal of the plurality of pulse compression signals under a condition that the target relative speed is within a preset assumed speed range. Calculate a plurality of integral evaluation values for evaluating the signal quality of the band synthesized signal using a set of phase compensation amount candidate values, and based on the plurality of integral evaluation values, from among the set of candidate values, A radar apparatus, wherein a phase compensation amount corresponding to each pulse compression signal is selected.   The radar apparatus according to claim 7, wherein each of the plurality of phase compensation amount calculation units predicts a phase difference between the pulse compression signals in the assumed speed range, and the predicted phase difference and the candidate value A radar apparatus, wherein the plurality of integral evaluation values are calculated based on a difference between them. The radar apparatus according to claim 8, wherein
The plurality of transmission units generate the plurality of pulse modulated waves using a preset frequency modulation amount and modulation bandwidth,
Each of the plurality of phase compensation amount calculation units calculates a value proportional to the target relative speed and the frequency modulation amount and inversely proportional to the modulation bandwidth as the predicted phase difference. Radar device.   8. The radar apparatus according to claim 7, wherein each of the plurality of phase compensation amount calculation units has a threshold value distribution that is always within a range within the assumed speed range among the plurality of integration evaluation value combinations. A combination satisfying a condition that exceeds the value is searched, and a candidate value forming a combination obtained as a result of the search is selected as a phase compensation amount corresponding to each pulse compression signal from the set of candidate values. A radar device characterized by the above. The radar apparatus according to claim 7, further comprising a signal synthesis unit that integrates a plurality of the band synthesis signals respectively generated by the plurality of band synthesis units,
The radar apparatus according to claim 1, wherein the target candidate detection unit detects the target candidate based on an output of the signal synthesis unit. The radar apparatus according to claim 11, wherein
An integration unit that generates an integrated signal by performing coherent integration by Fourier transform on the output of the signal synthesis unit;
The radar apparatus according to claim 1, wherein the target candidate detecting unit detects the target candidate based on the integrated signal.

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