JPWO2014207858A1 - Rotating machine and rotating machine drive system - Google Patents

Abstract

In a 6-phase rotating machine drive in which a 3-level inverter is arranged in parallel, a rotating machine and a rotating machine drive system capable of stable rotation at high speed and low noise are provided. A rotating machine drive system for storing a winding in a plurality of slots to form a stator and driving a rotating machine having two sets of three-phase windings that are Y-connected. The neutral points of the three-phase windings are insulated from each other, and the windings of each phase of each set are wound from the first slot position, and then wound from the second slot position via a crossover wire. The two sets of windings before and after the crossover are connected to each other at the axial target position, the rectifier circuit connected to the AC power source, the rectifier circuit and the AC Two sets of three-level inverters that provide three-level AC output to each phase are provided, and two sets of three-level inverters supply power to each of the three-phase windings of each set of the rotating machine and are Y-connected 2 It is characterized in that currents of opposite phases are applied to the same phase winding of the three-phase windings of the pair .

Description

  The present invention relates to a rotating machine and a rotating machine drive system driven by a plurality of inverters, and in particular, vibrations of a motor caused by different currents flowing between parallel circuits due to parallelization when the capacity is increased by a plurality of inverters. The present invention relates to a rotating machine and a rotating machine drive system that can suppress the rotation.

  With the recent increase in capacity of semiconductor switching elements such as IGBTs, MW class rotating machine drives using inverters are possible.

  Further, as a method for further increasing the capacity, there is parallelization of inverters. This is intended to increase the capacity of a rotating machine drive system by partially connecting a plurality of inverters in parallel to supply power that cannot be supplemented by a single inverter.

  As a parallel circuit system, Patent Documents 1 and 2 disclose a rotating machine drive configuration in which DC units are connected in parallel. Patent Document 3 discloses a method of making the windings of the rotating machine symmetrical in order to balance the electromagnetic force in the rotating machine at the time of abnormality.

JP-A-3-15273 Japanese Patent No. 4380755 Japanese Patent No. 4691897

  As a method for increasing the capacity by parallelization, the rotating machine drive system described in Patent Document 1 includes two three-level inverters in which DC circuits are connected in parallel, and the two three-level inverters include windings different in phase by 60 degrees. The electric motor is driven. According to Patent Document 1, there is a feature capable of suppressing the third-order voltage pulsation in the DC circuit portion. Further, the AC voltage generated from this inverter is stabilized. Further, the capacitor capacity for smoothing the DC circuit portion can be reduced.

  The technique of Patent Document 1 is based on the premise that exactly the same current flows in the parallel circuit of the rotating machine. However, in reality, since there are impedances of the inverter power supply and the rotating machine in the two independent circuits, the same current cannot be caused to flow due to a difference in performance of the power supply or a slight difference in impedance of the rotating machine. For example, the harmonic component of the current also changes depending on the switching timing of the inverter. Along with this, there is a problem in that the same electric current flows and the inside of the rotating machine where the electromagnetic force should be balanced becomes unbalanced and cannot be rotated stably. In particular, in the case of a large-capacity high-speed rotating machine, balance is important for stable rotation, and vibration of the rotating machine leads to a major accident. Or problems, such as noise, arise.

  In Patent Document 2, harmonics are described, but two sets of three-phase windings connected in parallel are connected at a neutral point, and the purpose is to reduce electromagnetic noise. When the two sets of three-phase neutral points float at different timings, current flows between them, and current that does not contribute to the rotating machine output increases. Large-capacity rotating machines and inverter circuits have a problem that cooling due to such increased loss due to current becomes difficult and cannot be reduced in size.

  Patent Document 3 describes a difference in rotating machine current between parallel circuits that occurs during an abnormality, and does not consider the difference in current due to switching.

  The present invention has been made in view of the above, and provides a low-noise rotating machine and rotating machine drive system that can stably rotate at a high speed in a six-phase rotating machine drive in which three-level inverters are arranged in parallel. Objective.

In order to achieve the above object, the rotating machine drive system of the present invention drives a rotating machine including two sets of three-phase windings that are Y-connected, by forming windings in a plurality of slots to form a stator. The neutral point of two sets of three-phase windings connected in a Y-direction is insulated from each other, and the windings of each phase of each set are wound from the first slot position, A rotating machine configured to wind from the second slot position via the crossover to reach the neutral point, and the two sets of windings before and after the crossover are placed at the axial target position, and an AC A rectifier circuit connected to a power source, and two sets of three-level inverters connected to the rectifier circuit and providing a three-level AC output to each AC phase are provided. Power is supplied to each of the phase windings, and two sets of three-phase windings that are Y-connected Wherein the opposite phase current is applied to each other in the winding phase.

  With this configuration, the electromagnetic force in the motor generated by the current harmonics output by the two single inverters is always 180 degrees rotationally symmetric, so no eccentric force acts on the rotor, so even a high-speed, large-capacity motor can be used. It can be driven stably.

The figure which shows the coil | winding structure of the synchronous machine (motor) of this invention. The figure which shows the whole structure of the rotary machine drive system which concerns on this invention. The figure which shows the structural example of a 3 level inverter. The figure which shows the arrangement | positioning of the coil of the motor of FIG. 1, and the connection method to an inverter. The figure which shows the coil arrangement | positioning which expand | deployed the winding of the motor of FIG. The figure which shows the specific circuit structure of a controller. The figure which shows the relationship between the carrier signal which concerns on this invention, a voltage command value, and a phase voltage. The figure which shows the coil | winding structure example which does not employ | adopt this invention. The figure which shows a ring primary mode. The figure which shows the example from which a coil current waveform differs. The figure which shows the annular | circular secondary mode which arises in Fig.10 (a). The figure which shows the example from which a coil current waveform differs. The figure which shows the annular | circular quartic mode which arises in Fig.11 (a). The figure which shows the relationship of a carrier signal, voltage command value, and phase voltage when not reversing a carrier. The figure which shows the expansion | deployment example of the other rotary machine coil | winding employable by this invention. The figure which shows the example of the other rotary machine coil | winding employable by this invention. The figure which shows the example which has arrange | positioned the coil of each pole 90 degree | times rotation symmetry. The figure which shows the example of the 6 pole rotating machine coil | winding employable by this invention.

  Embodiments of the present invention will be described below with reference to the drawings.

  FIG. 2 shows the overall configuration of the rotating machine drive system according to the present invention. However, FIG. 2 shows the rotating machine drive system 2 when the rotating machine is a synchronous motor. The rotating machine in this case shows an example of a synchronous motor, but it may be a synchronous generator or a synchronous generator motor.

  The components of the rotary machine drive system 2 in FIG. 2 are roughly classified into a rectifier 30, an inverter system 21, and a synchronous machine 10. In addition, a power system 50 is linked to the rotating machine drive system 2, and torque is generated in the rotor (shaft) 7 of the synchronous machine 10 by the inverter system 30 of the rotating machine drive system 2, and torque is transmitted to the load 6. .

  In the case of FIG. 2, since a synchronous motor is intended, the rectifier 30 is installed because electric power is supplied from the electric power system 50 to the synchronous rotating machine 10. Therefore, when a synchronous generator is used as a synchronous machine, an inverter is provided instead of the rectifier 30 and power may be sent from the generator side to the power system side.

  Hereinafter, with the synchronous motor in mind, the configuration and schematic functions of the components of the rotating machine drive system 2 will be described in order from the side closer to the power system 50.

  First, the rectifier 30 includes a transformer 320 provided for the purpose of electrical insulation and voltage conversion with respect to the power system 50, and a diode rectifier 310 that is connected to the transformer 320 and converts input AC power into DC power. . The rectifier 30 rectifies AC power received from the power system 50 into DC and supplies DC power to the inverter system 20.

  Next, the inverter system 21 will be described. The inverter system 21 includes a power main circuit and its control circuit portion. First, the power main circuit includes three-level inverters 230 and 235, filter reactors 240 and 250, and DC capacitors 400P and 400N. Among these, the DC input terminals P, M, and N (P: positive terminal, M: neutral point terminal, N: negative terminal) are connected in parallel to the three-level inverters 230 and 235. Further, two DC capacitors 400P and 400N having the same capacity are connected to the DC circuit connecting the rectifier 310 and the three-level inverters 230 and 235, and each terminal of the DC capacitors 400P and 400N is connected to the three-level inverter as shown in FIG. 230 and 235 are connected to DC terminals P, M, and N. The DC capacitor 400 is installed for the purpose of reducing the pulsation of the DC power output from the rectifier 30.

  The synchronous machine 10 includes two sets of three-phase stator windings that have power ground terminals PE connected to a ground point E and are insulated. Each set of three-phase stator windings is connected to inverters 230 and 235 for power supply. A rotor (shaft) 7 of the synchronous machine 10 is connected to a load 6. The connection relationship between the two sets of three-phase stator windings of the synchronous machine 10 will be described later with reference to FIGS. 4 and 5.

  The control circuit portion of the inverter system 21 includes a controller 101 and its input / output circuit portion, and the inverters 230 and 235 are controlled by a gate signal G provided by the controller 101.

  The controller 101 detects the output current i of the inverters 230 and 235 by the current sensors 81A, 81C, 81X and 81Z, and detects the output voltage v of the inverters 230 and 235 by the voltage sensors 82AB, 82BC, 82XY and 82YZ. The detected value is input to the controller 101.

  In the following illustrations and explanations, the promises for assigning symbols are as follows. First, the AC side terminals of the three-level inverters 230 and 235 in which the DC input terminals P, M, and N are connected in parallel are all expressed in U, V, and W phases. However, there are cases where it is better to explain the amount of electricity handled by the AC side terminals U, V, W of the 3-level inverter 230 and the AC side terminals U, V, W of the 3-level inverter 235. For this reason, in the following description, the current i and voltage v on the inverter 230 side are given symbols A, B, and C or a, b, and c to distinguish between phases and lines, and the current i on the inverter 235 side. The voltage v is denoted by the symbols X, Y, Z or x, y, z to distinguish between phases and lines.

  In FIG. 2, only two currents out of the three phases of current i (in the figure, ia and ic on the inverter 230 side, ix and iz on the inverter 235 side) are input to the controller 101, Only two line voltages among the lines (in the drawing, vab and vbc for the inverter 230 side, vxy and vyz for the inverter 235 side) are input to the controller 101. This is because the remaining one-phase currents ib and iy and the remaining one-line voltages vca and vzx can be obtained by synthesis, and the configuration of the input circuit portion is simplified. Treated as a quantity.

  The detailed calculation contents of the controller 101 will be described later with reference to FIG. 6. In short, here, the gate signals G (G-ABC, G-ABC,...) Of the three-level inverters 230 and 235 so that the torque command τref and the output torque to the synchronous machine 10 coincide. G-XYZ) and the gate signal G is output to the three-level inverters 230 and 235, respectively. The gate signal G-ABC supplied to the three-level inverter 230 and the gate signal G-XYZ supplied to the three-level inverter 235 are distinguished by the symbols A, B, C or the symbols X, Y, Z described above. Is displayed. The controller 101 calculates gate signals for controlling the three-level inverters 230 and 235. The number of gate signals is 12 points per inverter.

  FIG. 3 shows a configuration example of a three-level inverter as an example 230. Since the three-level inverter 235 has the same configuration as the three-level inverter 230, the description thereof is omitted.

  The three-level inverter 230 is an inverter composed of three arms for each AC side terminal. Each arm includes four IGBT elements connected in series and two diodes connected to a terminal M at a DC neutral point.

  In this figure, the arm connected to the U-phase AC terminal U is 230 U, the arm connected to the V-phase AC terminal is 230 V, and the arm connected to the W-phase AC terminal is 230 W. Since each arm has basically the same configuration, an example of the arm 230U will be described here.

  In the arm 230U, four sets of IGBT elements (230m, 230n, 230s, 230v) are connected in series, and both ends of the four series-connected IGBT element arrays are connected to the positive terminal P and the negative terminal N of the DC circuit, respectively. Has been. In the illustrated example, the cathode side of the IGBT element 230m is connected to the positive terminal P of the DC circuit, and the emitter side of the IGBT element 230v is connected to the negative terminal N of the DC circuit. Each IGBT element is composed of an IGBT and a diode connected in reverse parallel to the IGBT.

  In the series arm composed of four IGBT elements connected in series, the upper two arms 230m and 230n constitute the upper arm, and the lower two 230s and 230v constitute the lower arm. And the connection point of the upper and lower arms is connected to the U-phase AC terminal.

  Further, a diode 230a, 230d is connected in series between the connection point of the two IGBT elements 230m, 230n of the upper arm of the series arm and the connection point of the two IGBT elements 230s, 230v of the lower arm of the series arm. The circuit is connected. The series circuit connection point of the diodes 230a and 230d is connected to the DC neutral point terminal M. As shown in FIG. 8, the DC neutral point terminal M is a connection part of a series circuit of two DC capacitors 400P and 400N having the same capacity.

  Although not described, the other arms 230V and 230W have the same configuration as 230U. In such a configuration, the branch current iMabc of the DC current iM flows from the DC neutral point terminal M of the three-level inverter 230, and this branches to each arm. The sum of the output currents of the arms whose AC output voltage is the same potential as that of the DC neutral point terminal M is iMabc.

  The IGBT element is driven by inputting the gate signal output from the controller 101 to the gate terminal which is the control electrode of each IGBT element of the three-level inverter 230 having such a configuration. A specific control method of the IGBT element will be described using the arm 230U as an example.

The gate signal G in the three-level inverter is output in the following three patterns.
Pattern 1: IGBT elements 230m and 230n are on, 230s and 230v are off Pattern 2: IGBT elements 230n and 230s are on, 230m and 230v are off Pattern 3: IGBT elements 230s and 230v are on, 230m and 230n are off Pattern 1 The AC terminal U is the potential of the positive terminal P of the DC circuit, the pattern 2 is the AC terminal U is the potential of the intermediate terminal M of the DC circuit, and the pattern 3 is the negative terminal of the DC circuit. Since it becomes the potential of the terminal N, it is possible to output three voltages of + Vdc / 2, 0, and 1 Vdc / 2 when viewed from the DC midpoint potential. This relationship is the same for other AC terminals.

  The schematic configuration of the rotating machine drive system 2 configured with two sets of three-level inverters has been described above with reference to FIGS. 2 and 3. The characteristic part of the present invention in this system 2 is the winding structure of the synchronous machine 10 and the control method based on this winding structure. Therefore, in the following description, the winding structure of the synchronous machine 10 will be described first, and then the control method will be described sequentially.

  First, the winding configuration of the synchronous machine (motor) 10 of the present invention is shown in FIG. The motor 10 includes two sets of three-phase windings as described with reference to FIG. As shown in FIG. 1, the ABC phase and the XYZ phase constitute three phases, respectively. This is a six-phase motor in which one set of three phases is constituted by coils 10A, 10B and 10C, and another set of three phases is constituted by 10X, 10Y and 10Z. Each of the three-phase coils is Y-connected, and has neutral points N1 and N2, respectively, and the neutral points are not connected.

  Further, as shown in FIG. 2, the U-phase is expressed as A-phase on the 3-level inverter 230 side and X-phase on the 3-level inverter 235 side, but the A-phase and X-phase representing the U-phase are fundamental waves. The coils are arranged so that their current phases differ by 180 degrees. Thereby, the U-phase coils 10A and 10X are arranged in the same direction position, the V-phase coils 10B and 10Y are arranged in the same direction position, and the W-phase coils 10C and 10Z are arranged in the same direction position. It is supposed to be configured.

  The arrangement of the motor coil in FIG. 1 and a method of connecting to the inverter will be further described with reference to FIG. The V-phase (B, C-phase) and W-phase (Y, Z-phase) are arranged with the U-phase (A, X-phase) coils electrically shifted by 120 degrees. Therefore, only U-phase (A-phase, X-phase) coils are shown. Here, an example in which a four-pole motor is configured will be described.

  As shown in FIG. 4, the U-phase (A) of the three-level inverter 230 is connected to the winding 10A of the motor 10, and the U-phase (X) of the three-level inverter 235 is connected to the winding 10X of the motor 10. ing. An example of winding will be described with an example of 24 slots.

  Winding 10A consists of one turn between slot numbers 1 and 6, and one turn between slot numbers 2 and 7 to form two windings. 1 turn, and then 1 turn between slot numbers 14 and 19 to form 2 windings. Note that the two sets of windings before and after the crossing line 10k are placed at positions that are 180 degrees out of phase with each other.

  Similarly, the winding 10X has one turn between the slot numbers 7 and 12 and one turn between the slot numbers 8 and 13 to form two windings. , 24 and one turn between slot numbers 20 and 1 to form two windings. Note that the two sets of windings before and after the crossing line 10k are placed at positions that are 180 degrees out of phase with each other. That is, the windings constituting each phase are placed at positions that are axially targeted before and after the crossing line 10k.

  In this way, the coils 10A and 10X are each composed of four coil windings, and are arranged in 180-degree rotational symmetry so that two of them are opposed to each other, and these four coil windings are all connected in series. Opposing coil windings are electrically connected by a crossover wire 10K. Electrically, the coil 10A forms an N pole, and the coil 10X forms an S pole.

  FIG. 5 shows a coil arrangement in which the windings of the motor are developed. This motor has four poles and 24 stator slots, and each coil has two layers of distributed winding wound with a pitch of slot numbers 1-6. According to the winding of FIG. 5, the solid line current flowing from the terminal A flows to the neutral point N1 via the slot number 1-6-2-7-13-18-14-19.

  In the present invention, since the terminal A and the terminal X are operated so as to input alternating currents having the same magnitude in the opposite directions of the fundamental wave, at the timing described above when the current flows from the terminal A, the neutral point N2 The flowing dotted line current flows to the terminal X via the slot number 1-20-24-19-13-8-12-7. By doing so, a magnetic field is generated alternately in N and S.

  According to the combination of the inverter and the motor shown in FIG. 4, since the same current always flows through the coils wound in series, the electromagnetic force (mainly attractive force) generated by the opposing N-pole coils is equal, and the rotor Eccentricity does not work. Further, the electromagnetic force generated by the opposing S-pole coils is also equal, and this also does not act to decenter the rotor. This is true even if the currents flowing in 10A and 10X differ, and by using such a winding, a 6-phase motor driven by two parallel inverters can be smoothly rotated without eccentric force. The noise of the motor can also be suppressed.

  The characteristics of the winding of the motor 10 of the present invention have been described above. According to this, as shown in FIG. 1, it was set as the motor provided with two sets of three-phase windings. Of these, the two coils of the same phase are arranged in a straight line. When the current flowing through one coil is in the direction from the input terminal toward the neutral point, the current flowing through the other coil is in the direction from the neutral point toward the input terminal. As a result, the currents flowing through the two coils in the same phase of the two sets of three-phase windings flow in the same direction.

  On the premise of such winding arrangement, in the control method of the present invention, the currents flowing through the two coils in the same phase of the two sets of three-phase windings are controlled so that the fundamental wave phases are opposite to each other. Hereinafter, a control method for realizing this control will be described. This control is realized by the controller 101.

  Next, the controller 101 will be described with reference to FIG. The controller 101 includes a three-level inverter 230 controller 111ABC, a three-level inverter 235 controller 111XYZ, a torque distributor 1100, and a PWM carrier generator 1101. The two sets of controllers 111ABC and 111XYZ control the three-phase currents of each set supplied from the three-level inverters 230 and 235 to the rotating machine 10.

  The input given to the controllers 111ABC and 111XYZ will be described. Since the original function of the controllers 111ABC and 111XYZ is in torque control, one of the inputs is a torque command τref. The torque command τref is divided by ½ by the torque distributor 1100 and input to the controllers 111ABC and 111XYZ in common. Accordingly, the two controllers 111ABC and 111XYZ operate with the torque command τref having the same value.

  Further, since it is necessary to derive an actual torque corresponding to the torque command τref, the detection values of the voltage sensor and the current sensor are obtained. As shown in FIG. 2, the controller 111ABC uses ia, ic, and the controller 111XYZ uses ix, iz, and the voltage v uses the controller 111ABC using vab, vbc, and the controller 111XYZ uses vxy as shown in FIG. , Vyz.

  As other inputs, a d-axis current command value Idref for vector control in the controllers 111ABC and 111XYZ and a carrier signal Tri for determining the ignition timing are used.

  Based on these signals, the controllers 111ABC and 111XYZ are operated so that the torque finally given to the rotor of the synchronous machine 10 by the corresponding inverters 230 and 235 coincides with the divided torque command τref / 2. AC output voltage command values varef, vbref, and vcref of the inverter are obtained from the 2-phase / 3-phase converter 111lABC.

  The calculated AC output voltage command values varef, vbref, vcref are compared in magnitude with carrier signals TriP, TriN, which are outputs of the carrier generator 1101, in the PWM calculator 1112ABC. As a result, the gate signals G 1 ABC and G 1 XYZ are calculated so that the instantaneous average value of the inverter output voltage matches the voltage command value.

  A specific calculation method will be described with reference to the drawings. However, since the controller 111ABC and the controller 111XYZ have the same circuit configuration and are provided with the same arithmetic unit, an example of the controller 111ABC will be described.

  First, the voltage detection values vab and vbc are input to the two-phase three-phase calculator 1101ABC and converted into three-phase phase voltages Vanl, Vbnl, and Vcnl. Since the calculation in the two-phase / three-phase calculator can be performed by a known method, a detailed description is omitted. Similarly, the current detection values ia and ib are input to the two-phase three-phase calculator 1102ABC and converted into three-phase currents ia, ib and ic.

  The three-phase voltage detection values Vanl, Vbnl, Vcnl, and current detection values ia, ib, ic are input to the phase calculator 1103ABC, and the stator winding induced voltage phase of the synchronous machine 10 is calculated. Specifically, the phase calculation is realized by estimating the induced voltage from the input current of the synchronous machine 10, the synchronous machine impedance, which is a known value, and the input terminal voltage, and calculating the phase θabc of the voltage. Since this calculation is also a phase detection method widely known in this field, detailed description is omitted.

  In the torque calculator 1105ABC, the induced voltage phase calculated value θabc and the terminal voltages vanl, vbnl, vcnl and input currents ia, ib, ic of the synchronous machine 10 converted into three phases are input to the synchronous machine 10 by the three-level inverter 230. Torque τabc applied to the rotors of is calculated.

  In the torque compensator 1107ABC, the output of the torque distributor 1100 (torque command τref / 2) and the actual torque τabc are input, and the torque current command value (vector control q-axis current command value) iqref is reduced so as to reduce the deviation between the two. Calculate and output to the current controller 1108ABC. In the vector control, the d-axis current is also adjusted at the same time, and the d-axis current command value Idref is given to the current controller 1109ABC from the outside.

  The q-axis current iq-abc, which is a torque current used in these current controllers 1108ABC, 1109ABC, and the d-axis current id-abc, which is a reactive current, are calculated as follows. First, input currents ia, ib, and ic are input to an α-1β calculator 1104ABC. The α-1β calculator 1104ABC performs α-β conversion on the input three-phase quantity and outputs the converted value to the dq converter 1106ABC.

  In the dq converter 1106ABC, the α-β conversion value of the input current is dq converted using the phase calculation value θ, and the torque current iq-abc and the invalid current id-abc are obtained. Output to the controllers 1108ABC and 1109ABC. Current controller 1108ABC calculates q-axis output voltage command value vqref of two-level inverter 230 so that torque current command iqref-abc and iq-abc match. The current controller 1109ABC calculates the d-axis output voltage command value vdref of the three-level inverter 230 so that the reactive current command value idref-abc that is zero and id-abc that is the reactive current coincide.

  The voltage command values vdref and vqref are subjected to α-β conversion by the dq inverse converter 1110ABC and then three-phased by the two-phase / three-phase converter 111lABC, and the three-phase voltage command values varef, vbref and vcref are PWM It is output to the calculator 1112ABC.

  As described above, the three-level inverter 230 is controlled to generate a torque that matches the torque command. On the other hand, the controller 111XYZ performs the same calculation as the controller 111ABC. That is, using the voltage detection values vxy, vyz, and current detection values ix, iy, the output of the torque command distributor 1100 and the torque applied by the three-level inverter 235 to the rotor of the synchronous machine 10 coincide with each other. The gate signal is calculated. Note that the PWM computing units in the controller 11lABC and the controller 11lXYZ have the same configuration.

  The above-described configuration of FIG. 6 in the controller 101 has been conventionally used and is well known, and the controller 101 of the present invention has the following features. This feature is that in the controller 111ABC and the controller 111XYZ, the carriers TriP and TriN input to the PWM calculator 1112ABC are phase-inverted by the multiplier 1115, and thus the X phase, Y phase, and Z of the synchronous machine 10 are obtained. The voltage phase induced in the phase is inverted from the voltage phase induced in the A phase, B phase, and C phase of the synchronous machine 10.

  In FIG. 7, carrier signals TriP and TriN, which are outputs of the carrier generator 1101, carrier signals TriP2 and TriN2 phase-inverted by a multiplier 1115, and voltage command values varef, vxref, In addition, a time-series change of the phase voltage obtained as a result is shown. However, these waveforms represent the a-phase waveform of the controller 11lABC in the upper row and the x-phase waveform of the controller 11lXYZ in the lower row.

  FIG. 7 shows the carriers TriP and TriN of the three-level inverter 230, the sinusoidal AC output voltage command value varef, and the a-phase potential Va in order from the top, and then the carriers TriP2 and TriN2 and the voltage command of the three-level inverter 235. The waveform of the value vxref and the phase voltage Vx of the x phase is shown. The horizontal axis represents time. When Varef and Vxref are positive, the point where TriP intersects is shown, and when Varef and Vxref are negative, the time of the time when Trif intersects with TriN is shown by t1 to t22.

  According to the waveform of FIG. 7, the triangular waves TriP and TriN given to the PWM calculator 1112ABC of the controller 11lABC are triangular waves having the same phase and amplitude but different direct current biases. Due to the DC bias, the triangular wave TriN is set to 0 when the triangular wave TriP reaches the maximum value T, and the triangular wave TriN is set to the negative maximum value (−T) when the triangular wave TriP reaches the minimum value 0. Accordingly, the gate signal G-ABC is calculated from the re-controller 11lABC using the carrier, and the a-phase potential Va is generated.

  Triangle waves TriP2 and TriN2 in which the phases of TriP and TriN are inverted by a multiplier 1115 are input to the controller 11XYZ as carriers, and the gate signal G-XYZ is calculated using the carriers to generate an x-phase potential Vx.

  Triangular waves TriP2 and TriN2 given to the controller 11lXYZ side are also triangular waves having the same phase and amplitude and different direct current biases. This is obtained by inverting the phases of the triangular waves TriP and TriN given to the controller 11lABC side by the multiplier 1115. is there. Accordingly, the triangular wave TriP2 is set to 0 when the triangular wave TriN2 reaches the maximum value T, and the triangular wave TriP2 is set to the negative maximum value (−T) when the triangular wave TriN2 reaches the minimum value 0. Thus, the gate signal G-XYZ is calculated from the re-controller 11lXYZ2 using the carrier, and the x-phase potential Vx is generated.

  The a-phase potential Va and the x-phase potential Vx obtained as described above are signals of three levels: positive, zero, and negative. In the period in which the a-phase potential Va is positive or negative, x This is a period in which the phase potential Vx is negative or positive, and the polarity is inverted. However, in the period in which the a-phase potential Va is zero, the x-phase potential Vx is also zero.

  As described above, the voltage command values Varef and Vxref of the three-level inverters 230 and 235 are symmetrical waveforms. The carriers TriP and TriN, and TriP2 and TriN2 have the same phase and are shifted from each other by 180 degrees. When the phase voltages Va and Vx in which the inverter voltage is controlled on / off at the timings t1 to t22 are seen, as shown in FIG. 7, the change timings of the phase voltages Va and Vx are the same, the waveforms are the same, and It becomes symmetric.

  When the above control result is shown in FIG. 5 showing the coil arrangement in which the windings of the motor are developed, for example, a positive phase voltage is generated and applied to the coil 10A between times t1 and t2 in FIG. In this case, a negative phase voltage is generated and applied to the coil 10X. This means that when a current is flowing from the terminal A toward the neutral point N1, a current is flowing from the neutral point N2 toward the terminal X. Further, this is equivalent to flowing the same current i from the terminal A to the terminal X via the neutral point N1 and the neutral point N2 in FIG.

  The present invention is configured and controlled as described above, and the meaning of doing so will be described. Specifically, what will happen when such a configuration and control method are not employed will be described in comparison with the events in the present invention.

  First, regarding the winding configuration, instead of the coil arrangement of the present invention, for example, a coil is arranged as shown in FIG. In the assumed case, two windings 10A (indicated by white coils in FIG. 8) connected to the terminal A are arranged adjacent to each other, and two windings 10X (hatched in FIG. 8) connected to the terminal X are arranged. N poles and S poles are alternately arranged adjacent to each other.

  In this case, an eccentric force acts on the rotor 10 due to the difference between the coil currents of 10A and 10X. Such an excitation force mode is called an annular primary mode. FIG. 9 shows an annular primary mode. An eccentric force acts in a predetermined direction with respect to the rotating machine shaft. When this circular primary mode occurs, the rotor cannot be rotated stably.

  Furthermore, the ring mode of the excitation force changes depending on the difference in the current waveform. This will be described with reference to FIGS. FIG. 10A and FIG. 11A show examples in which the waveform of the coil current flowing through the coils 10A and 10X is different. Here, the thick solid line indicates the coil current waveform iA flowing through the coil 10A, the thick dotted line indicates the coil current waveform ix flowing through the coil 10X, and the thin solid line indicates the basic waveform i.

  The waveform of FIG. 10A is a harmonic superimposed on the fundamental wave i. However, when a positive harmonic is placed on iA as exemplified in the section T1, a negative harmonic is applied to ix. It is a waveform with waves. Conversely, as illustrated in the section T2, when a negative harmonic is on iA, a positive harmonic is on ix. On the other hand, the waveform of FIG. 11A is also a harmonic superimposed on the fundamental wave, but as shown in the section T1, when a positive harmonic is on iA, ix is also positive. It is a waveform with harmonics of values. Conversely, as illustrated in the section T2, when a negative harmonic is on iA, a negative harmonic is also on ix.

  In contrast to such a difference in current waveform, first, if the current flowing in the A phase and the X phase is different as shown in the waveform shown in FIG. 10A, the forces of the opposing north and south poles are different. As shown in FIG. 10B, an excitation force in the circular secondary mode is generated. An eccentric force is generated in two directions whose phases are 90 degrees different from each other in the excitation force of the circular secondary mode.

  However, in the current waveform as shown in FIG. 11A, the electromagnetic force exerted on the rotor by the opposite N pole and the opposite S pole is the same. This is because the electromagnetic force (mainly the attractive force) between the stator and the rotor depends only on the strength of the magnetic poles regardless of the direction of the magnetic poles. For this reason, when the current waveform of FIG. 11 (a) flows through the motor of FIG. 4, the circular secondary mode of the excitation force does not appear, but instead the circular quadratic mode shown in FIG. 11 (b). Will appear. Higher modes other than the primary ring are not forces that decenter the rotor, but noise is generated by deforming the stator and vibrating the motor outer diameter.

  In general, in the ring mode of the exciting force, noise becomes a problem in the low order. In the case of the second or higher order, the number of deformation increases as the order increases, and the amount of displacement with respect to the electromagnetic force in the same radial direction decreases. For this reason, in order to reduce the noise of the motor, it is necessary to devise a high-order mode so that the low-order mode of the ring is not output. In the present invention, the low noise of the motor is realized by combining with inverter switching. it can.

  The above mainly describes measures against imbalance in the spatial distribution of electromagnetic force. The main causes of noise were deformation of the motor outer diameter and eccentric force acting on the shaft. The present invention pays attention to the excitation force of this annular mode.

  In the present invention, as a device in the control method, the carrier of the three-level inverter is reversed. The problem of not reversing the meaning of this reversal will be described. FIG. 12 shows a case where the carrier phases are the same. In FIG. 12, the waveform of the upper half controller 11lABC is the same as that of FIG. 7, but the waveform of the lower half controller 11lXYZ is the case where the carriers TriP2 and TriN2 are in phase with the upper half TriP and TriN.

  The voltage command value Vxref in this case is the same as in FIG. 7, but since the carrier to be compared is different from that in FIG. 7, the timing of the ignition timings t1 to t22 is shifted, and the waveforms of the phase voltages Va and Vx are vertically symmetrical. It will not be. The fundamental voltages of the phase voltages Va and Vx are the same, but since the harmonic voltage components are different, the currents also have different harmonic components. An image of current at this time is shown in FIG.

  If the current also reflects the voltage waveform and the carrier phase is 180 degrees different, the A phase and the X phase are completely symmetrical in the vertical direction as shown in FIG. 11A, and if the carriers are in phase, as shown in FIG. It becomes a waveform. In this assumption, even if the circuit impedances of the A phase and the X phase are exactly the same and the performance of the inverter is exactly the same, the coil current waveforms of the A phase and the X phase differ depending on the carrier phase. The present invention paid attention to this point.

  An example of rotating machine winding that can be used in the present invention will be further described. FIG. 13 shows the same coil as FIG. 5, but the connection method for connecting the coils in series is different. This motor is also a 4-pole motor similar to FIG. 5 and the stator slot is 24 motors, and each coil is a two-layer distributed winding wound at a pitch of slot numbers 1-6. According to the winding in FIG. 13, the solid line current flowing in from the terminal A flows to the neutral point N1 via the slot number 1-6-13-8-13-18-1-20.

  In the present invention, since the terminal A and the terminal X are operated so as to input alternating currents having the same magnitude in the opposite directions of the fundamental wave, at the timing described above when the current flows from the terminal A, the neutral point N2 The flowing dotted line current flows to the terminal X via the slot number 2-7-12-7-14-19-24-19. By doing so, a magnetic field is generated alternately in N and S. Even in this case, although the winding order and the like are different, the windings constituting each phase are placed at positions which are axially targeted before and after the crossing line 10k.

  In this way, in the winding example of FIG. 13, there are two coils that constitute one pole, and one pole is connected in series to form a two parallel circuit. At this time, the coils are connected so that the sum of the voltage vectors of the series coils is the same in the two parallel circuits, and the opposing coils in the series coils are exactly 180 degrees rotationally symmetric.

  As shown in FIG. 14, the opposing series coils are 180 degrees symmetrical, but they are not 90 degrees rotationally symmetric with the adjacent pole coils and are offset by one slot. The reason for this is that, for example, when the coils of each pole are arranged 90 degrees rotationally symmetric as shown in FIG. 15, the induced voltage differs between the coils 10A and 10X because the phases of magnetic flux received from the rotor are different. This is because even if 235 generates the same voltage, the currents are different.

  The windings in FIG. 14 have two phase-coiled coils and two delayed coils in 10A and 10X, respectively, so that the voltage vectors match. The deformation mode of the electromagnetic force (attraction force) in this case is as shown in FIG. 11B, and an annular quaternary mode is generated. Strictly speaking, the circular secondary is not generated in FIG. 15, but the circular secondary is slightly generated in FIG. 14 because the 4-fold rotational symmetry is slightly broken. However, the effectiveness of the present invention does not change because the secondary mode is kept small.

  Moreover, although the case of 4 poles was given as an example in the winding examples of FIGS. 4 and 5, the same applies to the case of 6 poles as shown in FIG. 16. However, in this case, the electromagnetic force balance can be obtained when the coils having the same polarity are connected in series instead of the opposing coils in series. In this case, if the A phase is N polarity, the X phase is S polarity. As a result, the third-order annular mode is structurally generated. However, by switching the switching timing to the opposite phase, it is possible to increase the order to the sixth-order annular mode, thereby reducing the noise. In the case of the embodiment of FIG. 13, the same concept can be used regardless of the number of poles.

  In the above explanation, the coil position is strictly rotationally symmetric. However, even if the coil arrangement is shifted by several slots, the electromagnetic force is dispersed and the eccentric force is reduced or the excitation force is reduced. Needless to say, this has the effect of suppressing the lower-order modes of the ring.

  In the case of FIG. 13, the case where there are two coils constituting one pole is shown, but it goes without saying that it can be similarly configured if it is an even number.

  In this embodiment, the carriers of the two inverters are triangular waves having a phase difference of 180 degrees. However, even if the phase is not 180 degrees, the output currents of the two inverters are within 90 to 270 degrees. This waveform has almost the same effect because it has almost the opposite phase. However, since the symmetry of the current increases as the phase difference is closer to 180 degrees, it is desirable to set the phase difference to 180 degrees.

  As described above in detail, even when different currents flow between two parallel circuits, the present invention can provide an axially symmetric electromagnetic force by arranging the series coil group constituting one parallel circuit at an axially symmetric position. By this, an attempt is made to suppress the eccentric force acting on the rotor and the noise.

1: motor drive system, 6: load, 7: shaft, 10: synchronous machine, 20, 21: inverter system, 30: rectifier, 230, 235: 3-level inverter, 210m, 210o, 210p, 210q, 210r, 230m, 230n, 230o, 230p, 230q, 230r, 230s, 230t, 230u, 230v, 230w, 230x: IGBT elements, 230a, 230b, 230c, 230d, 230e, 230f: diodes, 50: power system, 101: controller, 10A, 10X: Winding, 10K: Crossover, 81A, 81C, 81X, 81Y: Current sensor, 82AB, 82BC, 82XY, 82YZ: Voltage sensor, 240, 250: Reactor, 400P, 400N: DC capacitor

Claims (6)

A rotating machine that includes two sets of three-phase windings that are Y-connected, with windings housed in a plurality of slots to form a stator,
The neutral points of the two sets of three-phase windings connected in the Y-direction are insulated from each other, and the windings of the respective phases of each set are wound from the first slot position and then connected to the second through the crossovers. While winding from the slot position to the neutral point, the two sets of windings before and after the crossover are placed at the axial target positions and the same phase of the two sets of three-phase windings connected in Y-direction. A rotating machine characterized in that currents of opposite phases are applied to the windings. The rotating machine according to claim 1,
The rotating machine is characterized in that the number of poles is 4N (N is a natural number), and the winding group constituting the circuit of each phase is such that windings at opposite positions are connected in series. . The rotating machine according to claim 1,
The rotating machine has 2N poles (N is an odd number greater than or equal to 3), and the winding group constituting the circuit of each phase is characterized in that the windings at opposite positions are connected in series. Rotating machine. A rotating machine drive system that drives a rotating machine having two sets of three-phase windings that are Y-connected, with a winding formed in a plurality of slots to form a stator,
The neutral points of the two sets of three-phase windings connected in the Y-direction are insulated from each other, and the windings of the respective phases of each set are wound from the first slot position and then connected to the second through the crossovers. A rectifier connected to an AC power source and a rotating machine configured to wind from the slot position to the neutral point, and the two sets of windings before and after the crossover are placed at the axial target position. And two sets of three-level inverters connected to the rectifier circuit and providing a three-level AC output to each AC phase,
The two sets of three-level inverters supply electric power to each of the three-phase windings of each set of the rotating machine, and the same-phase windings of the two sets of three-phase windings connected in the Y-phase are opposite in phase. A rotating machine drive system characterized in that a current of The rotating machine drive system according to claim 4,
Each of the three level inverters is operated by a PWM calculator that determines a starting point by comparing a sine-wave voltage command value and two sets of carrier signals, and is supplied to the PWM calculator of one of the three level inverters. Two sets of carrier signals are supplied to the PWM calculator of the other three-level inverter as the inverted signals. The rotating machine drive system according to claim 4 or 5,
A rotating machine drive system, wherein a phase difference between PWM control carriers of the two sets of three-level inverters is 90 to 270 degrees.

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