JPS6359112B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION (Field of Industrial Application) The present invention relates to a method for measuring frequency-shifted waves suitable for use in rotational unevenness measurement, FM modulation depth measurement, and the like. Furthermore, the present invention has little phase delay and can detect even small deviations even when the input frequency becomes high, and can also measure rotational unevenness for each revolution of a rotating body and remove abnormal phenomena during rotational unevenness measurement, such as dropouts. An object of the present invention is to provide a method for measuring a frequency-shifted wave that can be carried out.

(Prior art) A conventional method for measuring frequency-shifted waves is the method used in Japanese Patent Application Laid-Open No. 51-73370, which was developed by the applicant.
−16911) method.

As shown in FIG.
The flutter component is displayed on the flutter indicator B.

Among these, the system that displays the flutter component is a frequency-voltage conversion circuit (F/V conversion circuit) C, a frequency counter D, a variable attenuator E, and a high-pass filter F.
, an amplifier G, and a rectifier circuit H.

If the characteristic is a straight line, the F/V conversion circuit C has a DC voltage that is proportional to the height of the input measurement frequency and an AC voltage that increases and decreases around the DC voltage in proportion to the magnitude of wow and flutter. Both are multiplied and output, and the frequency counter D detects the frequency shifted wave (e.g. wow/flatter signal).
The variable attenuator E has an attenuation amount scaled by the frequency value and is designed to attenuate the conversion output in proportion to the input frequency, and the high-pass filter F has a wah filter. - Only the flutter frequency component is allowed to pass, and the average frequency of the frequency-shifted wave S of the flutter indicator B is calibrated with the output of the reference frequency.

In Fig. 1, the frequency counter D and the frequency -
A frequency-shifted wave (for example, a wow/flatter signal) S is input to the voltage conversion circuit C, the frequency counter D calculates the average frequency of the frequency-shifted wave S, and the F/V conversion circuit C calculates the average frequency of the frequency-shifted wave S. Frequency-shifted wave S
The output of the F/V conversion circuit C is controlled according to the average value of the frequency of the frequency-shifted wave S by the variable attenuator E, and the F/V conversion is performed by the high-pass filter F. The alternating current component contained in the output voltage of circuit C is extracted, it is amplified by amplifier G, rectified by rectifier circuit H, this rectified output is supplied to indicator B, and the flutter value is displayed on flutter indicator B. It is displayed as a percentage.

Here, the reason why the output of the F/V conversion circuit C is controlled by the variable attenuator E in accordance with the average value of the frequency of the frequency-shifted wave S is as follows.

Wow/flatter value W&F and F/V conversion circuit C
The relationship between the output and the output is as follows.

W&F=Δf/f×100%=ΔV/V×100% where f: Measurement frequency Δf: Deviation frequency (wow/flutter) V: Converted DC voltage ΔV: Converted wow/flutter voltage (AC) In other words, F- The output level V and ΔV of the V conversion circuit C are determined by the input measurement frequency f and its f
There is a relationship in which the magnitude varies depending on the instantaneous height Δf. Therefore, it is necessary to convert this relationship into a relationship between constant f and Δf at that time. As a result, the relationship between constant V and ΔV at that time can be determined.

Therefore, the device shown in Fig. 1 uses a variable attenuator E, and by adjusting it according to the average value of the frequency of the frequency-shifted wave S measured by a frequency counter D, the output of the variable attenuator E is , the converted DC voltage is always at a constant level and the wow/flutter voltage is proportional to the shift frequency correlated with the DC voltage. As a result, regardless of the input frequency, it is possible to extract the wow and flutter components in the FM component of that frequency in proportion to the input frequency.

(Problems with the prior art) In the conventional system shown in Figure 1, a digital system as shown in Figure 2 or an analog system as shown in Figure 3 is used as the F/V conversion circuit C. There are some difficulties as follows.

The digital system in Figure 2 consists of a limiter a, a clock pulse generation circuit b, an AND circuit c, a counter d,
It consists of a subtraction circuit e and a digital-to-analog converter f.

As shown in Figure 4, this method inverts the input pulse i output from limiter a to obtain gate signal j, and measures the period of pulse signal i by counting clock pulses k between gate signals j. Therefore, when the input frequency becomes high, it is difficult to increase the clock pulse k accordingly, resulting in a drawback that high resolution cannot be obtained.

The analog system shown in Figure 3 consists of a monostable multivibrator g and an integrator h that combines a resistor and a capacitor, so the demodulated signal has a phase lag and the frequency characteristics of the reproduced signal change. It has the disadvantage of being bad.

Next, the conventional rotational unevenness meter mainly measures the amount of frequency deviation of the input signal continuously, and the rotational unevenness per revolution is not measured very much. If this is a problem, use a rotational unevenness meter as shown in Figure 5.
The wow/flatter waveform output m is measured using an oscilloscope n.
The waveform is observed by synchronizing with the synchronization signal O of one pulse per rotation. In addition, the size of wow and flatta is measured by humans visually observing the wave height value, which has the disadvantage that the reading accuracy is poor and it cannot be used for automatic measurement.

(Object of the Invention) The present invention uses the F/V conversion circuit newly developed by the applicant as shown in FIG. This is to eliminate the problem.

(Example) The present invention will be explained in detail based on the drawings. 6th
The figure is a block diagram of the present invention, including a frequency counter 1 and a frequency-voltage conversion circuit (F/V conversion circuit).
2, a variable attenuator 3, a high-pass filter 4, an amplifier 5, a rectifier circuit 6, and a flutter indicator 7.

Of these, the components other than the F/V conversion circuit 2 are the same as those shown in FIG. Furthermore, in the present invention, instead of connecting the variable attenuator 3 to the output terminal of the F/V conversion circuit C to control the output of the circuit as shown in FIG. /V
It is connected before the output end of the conversion circuit 2 to control the operating frequency of the F/V conversion circuit 2 itself.

That is, in the present invention, the frequency-shifted wave S is supplied to the frequency counter 1 and the F/V conversion circuit 2,
The frequency counter 1 calculates the average frequency of the frequency-shifted wave, and the variable attenuator 3 controls the operating frequency of the F/V conversion circuit 2 according to this average value. A voltage proportional to the frequency of the frequency-shifted wave S is taken out, an alternating current component included in it is taken out from the output voltage of the F/V conversion circuit 2 by a high-pass filter 4, it is amplified by an amplifier 5, and then a rectifier 6 The rectified output is supplied to an indicator 7 to display the frequency deviation amount of the frequency-shifted wave in %.

In the present invention, the F/V conversion circuit 2 shown in FIG. 7 is used. The F/V conversion circuit 2 in FIG. 7 includes a limiter 8, monostable multi 9, 10, constant current source circuit 11, buffers a ₁ , a ₂ ,
Electronic switches S ₁ , S ₂ , S ₃ , capacitor C ₁ ,
It is composed of C ₂ and C ₃ .

The limiter 8 forms the waveform of the input frequency-shifted wave S to obtain an input pulse A (FIG. 8).

As shown in FIG. 8, the monostable multi 9 generates a pulse signal B at the falling edge of the input pulse A.
The monostable multi 10 is designed to generate the pulse signal C at the falling edge of the pulse signal B.

Switch S ₁ is set when pulse signal B is “1”
ON, OFF when input pulse A is “0”, switch _S2 is ON when input pulse A is “1”, and OFF when input pulse A is “0”.
OFF, _S3 is ON when pulse signal C is “1”,
It is set to be OFF when it is "0".

In the F/V conversion circuit 2 shown in FIG. 7, a constant current continues to flow from the constant current source circuit 11 to the capacitor _C1 , and the switch _S1 is turned off and the switch _S2 is turned off.
is ON and switch _S3 is OFF, the capacitor
The charging voltage of C ₁ increases linearly as shown in Figure 8 D, and when the input pulse A becomes "0", switch S ₂ turns OFF at the falling edge, and the rise in the charging voltage of capacitor C ₁ stops. At the same time, the voltage level of the sawtooth wave D at the time of stop is maintained in the capacitor _C2 .

Also, at this time, pulse signal B is generated at the falling edge of input pulse A, turning on switch _S1 , and the voltage charged in capacitor _C1 is discharged, dropping to 0 voltage as shown in Figure 8D. When the pulse signal B becomes 0 voltage, charging of the capacitor _C1 from the constant current source circuit 11 is restarted, and as a result, the charging voltage increases and a sawtooth wave D is generated every cycle of the input pulse A. It is set as.

Furthermore, when the pulse signal B becomes "0", the pulse signal C is generated at the falling edge, turning on the switch _S3 , and the voltage charged in the capacitor _C2 is charged to the capacitor _C3 and outputted. (Figure 8F).

In Fig. 7, the switch _S2 is not directly controlled by the input pulse A, but the input pulse A is input to the control section 12 to obtain the synchronizing signal J shown in Fig. 8, and when this synchronizing signal J is "1", When the switch _S2 is turned on, the switch _S2 is turned off when the synchronizing signal J is "0".

The control section 12 is composed of an inverting gate 13, a flip-flop 14, and an AND circuit 15.
And input pulse A from limiter 8 (Fig. 8)
is input to the AND circuit 15, inverted by the inverting gate 13 as shown in FIG. 5H, and input to the CK input terminal of the flip-flop 14. When the external trigger signal shown in FIG. 5G is input to , the output signal of the flip-flop 14 becomes as shown in FIG. 9I,
The signal J in FIG. 9 is output from the AND circuit 15, and this output signal J is sent to the switch _S2 .
This becomes the synchronization signal J that controls ON and OFF.

When the switch _S2 is turned OFF by this synchronization signal J, the voltage rise of E shown in FIG. 8 is stopped at that point, and that voltage level is maintained while the synchronization signal J is "0", so as shown in FIG. The output signal of the switch _S3 is as shown in FIG. 8K. Therefore, the output waveforms at points A, B, C, D, E, G, H, I, J, and K in FIG. 7 are as shown in FIG. 8.

In the present invention, the current from the constant current source circuit 11 is controlled by the variable attenuator 3 according to the average value of the frequency of the frequency-shifted wave S determined by the frequency counter C. The operating frequency is adjusted.

In the F/V conversion circuit 2 shown in FIG. 7, there is some discharge time T from when the level of the sawtooth wave D stops increasing at the falling edge of each input pulse A until the sawtooth wave D becomes 0. This discharge time T does not result in a measurement error for the following reason.

When the constant current source circuit 11 and the capacitor C are combined as shown in FIG. 9, the relationship is as follows: i∝Vcont...Vout=(1/C)∫idt=(i/C)t...

From the formula, V ₁ = (i/C) T ₁ , V ₂ = (i/C) T ₂ (see Figure 10) Here, when calculating the change ΔV in the output voltage due to the period T, ΔV = V ₂ - V ₁ = (i/C) T ₂ - (i/C) T ₁ = (i/C) (T ₂ - T ₁ )... In the formula, the period change ΔT is given by ΔT = T ₂ - _{T 1} , so V=(i/C)ΔT... i/C=K, ΔV=KΔT... As is clear from the equation, the change in output voltage ΔV is proportional to the change in period ΔT.

Next, assuming that there is a constant discharge time Dt within the period T as shown in Figure 11, the formula is V'out = (i/C) (t - Dt) = (i/C) t - (i/C )Dt (i/C)Dt is constant, so (i/C)Dt=A
Then, V'out=(i/C)t-A... From the formula, V ₃ = (i/C)T ₃ -A V ₄ = (i/C)T ₄ -A (See Figure 12) Here Calculating the change ΔV' in output voltage due to period T, ΔV'=V ₄ −V ₃ = {(i/C)T ₄ −A}−{(i/C)T
₃ −A} = {(i/C)T ₄ }−A−{(i/C)T ₃ }+A=(
i/C) (T ₄ − T ₃ )… In the formula, the period change ΔT′ is ΔT′=T ₄ −
Since it is given by T ₃ , ΔV'=(i/C)ΔT'... If i/C=K, ΔV'=KΔT'... Similarly to the equation, as is clear from the equation, if the discharge time Dt is constant, However, the output voltage change ΔV' is proportional to the period variation ΔT'.

Therefore, if the slope of the sawtooth wave D is constant, the discharge time will not result in a measurement error.

Now, assuming that ΔT is 1% of T, ΔT×100=T ∴ΔT=0.01×T ΔVout=ΔTi/C=0.01×T×i/C Here, f=1/T ∴ΔVout=0.01×(1/ f)×(i/C
)=(0.01/C)×(i/f) In order to keep the output voltage constant regardless of the frequency per 1% change in frequency, if C is fixed, i/f = N (N is a constant). Must be.

∴i=Nf... According to the formula, the current i is proportional to the control voltage Vout, so Vcont∝f, and by controlling Vcont, the wow and flutter output voltage can be made constant at any frequency.

(Effect of the invention) The present invention has the following various effects.

(1) Due to the relationship of frequency f∝(1/N)i,
By varying the current of the constant current source circuit 11 according to the average value of the frequency of the frequency-shifted wave determined by a frequency counter, the wow and flutter components of the frequency-shifted wave can be suppressed regardless of the changes in the input frequency. can be measured.

(2) Since the F/V conversion circuit 2 measures each cycle of the input frequency, the demodulated wave has a delay of only one cycle of the input frequency. Therefore, the frequency characteristics are excellent, and the frequency-shifted wave can be measured with high precision.

(3) High resolution is obtained because the sawtooth wave D is held using an analog method, and resolution does not deteriorate even if the input frequency becomes high. Therefore, it is possible to measure frequency-shifted waves with high accuracy.

(4) Since the discharge time of the sawtooth wave D does not cause an error, highly accurate frequency-shifted wave measurement can be performed.

(5) As the F/V conversion circuit 2, the input frequency-shifted wave S is waveform-shaped into a pulse signal, and this input pulse A is controlled by an external trigger signal G to obtain a synchronization signal J, and this synchronization signal J stops the voltage rise in Figure 8E, holds the voltage level, and extracts a voltage proportional to the period of the input pulse A, so one rotation of the rotating body It is also possible to measure rotational unevenness at each rotation.

(6) It is possible to obtain the desired synchronization signal J by appropriately selecting the input of the external trigger signal G, and remove unnecessary components included in the input signal only during this synchronization signal J, so that it is possible to remove unnecessary components contained in the input signal during rotational unevenness measurement. It is also possible to eliminate abnormal phenomena such as dropouts.

(7) Since the output signal K in Fig. 8 is normally input to the subsequent circuit through a capacitor, if the voltage level of the removal section L of the output signal K is suddenly cut off to 0 level, the DC component d at the time of cut-off is is discharged from the above capacitor. Therefore, the subsequent circuit cannot perform the next operation until the charging voltage of the capacitor is discharged, and the response time is delayed accordingly. Also, even if the next measurement starts,
Until the capacitor is charged to the DC voltage level of the output signal, a shock occurs at the subsequent stage due to a sudden voltage change, making it impossible to perform measurement, and the response time is delayed accordingly. However, in the present invention, the voltage level of the removal section L of the output signal K is not set to 0, but the DC voltage level at the time of stop is held, so there is no such drawback. Therefore, the response time is fast, and the frequency-shifted wave can be measured with high accuracy.

Moreover, since the DC voltage that is held is almost equal to the voltage that is output for the frequency of the input signal, there will be no sudden change in DC voltage even when the next measurement starts, and therefore there will be no sudden change in the DC voltage when the next measurement starts. There is no shock due to voltage changes, and virtually no noise due to shock occurs, making it possible to measure frequency-shifted waves with high precision in a short time.

[Brief explanation of drawings]

Figure 1 is an explanatory diagram of a conventional frequency-shifted wave measurement method, and Figures 2 and 3 are explanations showing various examples of F/V converters used in the conventional frequency-shifted wave measurement method. Fig. 4 is an explanatory diagram of the operation of the F/V converter shown in Fig. 2, Fig. 5 is an explanatory diagram of the measurement of the conventional rotation unevenness meter, and Fig. 6 is an explanation of the frequency-shifted wave measurement method of the present invention. 7 is a block diagram of an F/V conversion circuit used in the frequency-shifted wave measuring method of the present invention,
FIG. 8 is a waveform diagram explaining the operation of the block in FIG. 7, and FIGS. 9, 10, and 11 are diagrams explaining the operation of the F/V conversion circuit in FIG. 7. 1 is a frequency counter, 2 is an F/V conversion circuit, 3
is a variable attenuator, 4 is a high-pass filter, 5 is an amplifier, 6
is a rectifier circuit, 7 is an indicator, 8 is a limiter, 9,1
0 is a monostable multi, 11 is a constant current source circuit.

Claims (1)

[Claims] 1 Frequency-shifted wave S is divided into frequency counter 1 and F/
The frequency counter 1 calculates the average value of the frequency of the frequency-shifted wave, and the variable attenuator 3 controls the operating frequency of the F/V conversion circuit 2 according to this average value. F/V conversion circuit 2
A voltage proportional to the frequency of the frequency-shifted wave S is extracted from the output voltage, the alternating current component contained in it is extracted and rectified, and this rectified output is calibrated with the output whose average frequency of the frequency-shifted wave is the reference frequency. In the method for measuring a frequency-shifted wave in which the amount of frequency deviation of the frequency-shifted wave is displayed by supplying the frequency-shifted wave to an indicator 7, the F/V conversion circuit 2 is configured to supply the input frequency-shifted wave to The waveform of the shifted wave is shaped into a pulse signal, this input pulse is controlled by an external trigger signal to obtain a synchronization signal, and this synchronization signal stops the voltage rise of the sawtooth wave and holds the voltage level of the sawtooth wave at the time of stop. A method for measuring a frequency-shifted wave, characterized in that a voltage at a level proportional to the period of an input pulse is extracted.