JPS6346621B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a receiving device that receives orthogonally multiplexed multi-channel amplitude modulated (hereinafter abbreviated as PAM) signals that have been orthogonally multiplexed and then reach a receiving end via a transmission path. multiple complex basebands
Orthogonal multiplexing in which the PAM signal undergoes orthogonal amplitude modulation (hereinafter abbreviated as QAM) process and is then orthogonally multiplexed.
The present invention relates to a receiving device that receives a QAM signal and reproduces a plurality of complex baseband PAM signals through digital processing.

As a receiving apparatus of this type, a "digital processing type receiving apparatus for orthogonal multiplexed signals" has already been proposed in Patent Application No. 150238 of 1972 (Japanese Unexamined Patent Application Publication No. 81712/1982). However, although this digital processing receiver for orthogonal multiplexed signals only requires N/2-point complex data as output, it is necessary to perform N-point Fourier transform every T/2 seconds (where T is for each PAM). (N/T is the clock period of the signal, and N/T is the sampling frequency.) Furthermore, it is necessary to provide a switching control section every T/2 seconds at various locations, which has the disadvantage of complicating the hardware scale.

SUMMARY OF THE INVENTION In view of the above-mentioned drawbacks, it is an object of the present invention to provide a digital processing receiver for orthogonal multiplexed signals, which allows the use of an N/2-point Fourier transformer and minimizes the number of switching control sections every T/2 seconds. It is.

The present invention will be explained below using the drawings.

Figure 1 is a block diagram showing the transmission/reception system of orthogonal multiplexed QAM signals. 1 ₀ , 1 ₁ , ..., 1 _2L-1 is an input to which 2L PAM signals synchronized with each other with a clock period of T seconds are input. At the end, 2 ₀ , 2 ₁ ,..., 2 _L-1 are L T/2 second delay circuits, 3 ₀ , 3 ₁ ,..., 3 _2L-1
are 2L transmit baseband filters, 4 ₀ , 4 ₁ ,
..., 4 _2L-1 is a 2L modulator, 5 is a multiplex circuit that adds all the outputs of each modulator and sends it to the transmission path, 6 is a transmission path, 7 ₀ , 7 ₁ , ..., 7 _2L-1 is 2L demodulators, 8
₀ , 8 ₁ ,..., 8 _2L-1 are 2L receiving baseband filters, 9 ₀ , 9 ₁ ,..., 9 _L-1 are L T/2 second delay circuits, 10 ₀ , 10 ₂ ,... , 10 _L-1 are L T-second delay circuits 11 ₀ , 11 ₁ , . . . , 11 _2L-1 is an output terminal from which 2L PAM signals are output. In Figure 1, the Rth (R is less than or equal to L) input terminal 1 _R
and (L+R)th input terminal 1 _L+R respectively.
Assume that a PAM signal is input. Rth
The PAM signal is delayed by T/2 seconds in a T/2 second delay circuit _2R , and then band-limited and waveform-shaped by a transmission baseband filter _3R before reaching a modulator _4R . On the other hand, the (L+R)th PAM signal is band-limited and waveform-shaped by the transmission baseband filter 3 _L+R as it is, and then reaches the modulator 4 _L+R . In the modulators 4 _R and 4 _L+R , the in-phase carrier cos2πf _R t and the orthogonal carrier sin2πf _R t of the frequency f _R are respectively input as modulation carriers, and both modulation outputs are added in the multiplexing circuit 5 to obtain the center frequency f. An Rth QAM signal _R is formed. Here, the frequency of the carrier used in each modulator is 1R
f _R -f _R-1 = T ^-1 for R L-1, and the carriers in modulators 4 ₀ , 4 ₁ , ..., 4 _L-1 are alternately cosine waves and sine waves. It is located. It is already well known that orthogonally multiplexed QAM signals are output from the multiplexing circuit 5 by such modulation operations on the transmitting side. The orthogonal multiplexed QAM signal output from the multiplexing circuit is transmitted to the receiving side via the transmission line 6.

On the receiving side, a conversion that is completely opposite to that on the transmitting side is performed. That is, in demodulators 7 _R and 7 _L+R , the in-phase carrier cos2πf _R t and the orthogonal carrier
sin2πf _R t is input as a demodulation carrier, and the Rth demodulation output and (L+R)th demodulation output are band-limited and waveform-shaped by the receiving baseband filters 8 _R and 8 _L+R , respectively, for T/2 seconds. After delay-related correction is performed in the delay circuit 9 _R and the T-second delay circuit 10 _R , the signals reach the output terminals 11 _R and 11 _L+R , respectively. Here, the transmitting baseband filters 3 ₀ , 3 ₁ , ..., 3 _2L-1 and the receiving baseband filters 8 ₀ , 8 ₁ , ..., 8 _2L-1 all have the same frequency response G (ω) and a 3 dB reduction band ( It is assumed that it is a 1/2 T hertz resistance pass filter (hereinafter referred to as the effective band), and G ² (ω) is 6 dB at 1/2 T hertz.
If it is a normal Nyquist filter characteristic that decreases, the receiving side output terminals 11 ₀ , 11 ₁ , ..., 11 _2L-1 will receive a PAM signal that has no inter-symbol interference and no inter-channel interference at an appropriate sampling point. known to be obtained.

Next, consider performing all signal processing operations in the receiving section shown in FIG. 1 by digital processing. Here T/2
The second delay circuits 9 ₀ , 9 ₁ , ..., 9 _L-1 and the T second delay circuits 10 ₀ , 10 ₁ , ..., 10 _L-1 are simply provided to match the delay relationships of the reproduced PAM signals. Therefore, consider excluding this.

First, let the sampling frequency f _S of the system be N/T (however, N is an even number and N = L), and Z = e ^j2 〓 ^f/fS
shall be. An example of the received signal sample value system on the receiving side is Y
(z), then the output X _R (z) of the receiving baseband filter 8 _R in FIG. 1 is X _R (z) = 1/2 [Y(α _R Z )+Y(α _R ^-1 Z)]G(
Z)(1) X _R+L (Z)=j/2 [Y(α _R Z)−Y(α _R ^-1 Z)]G
(Z) When R is an odd number, X _R (Z) = j/2 [Y (α _R Z) - Y (α ^-1 _R Z)] G (
Z) X _R+L (Z)=1/2 [Y(α _R Z)+Y(α ^-1 _R Z)]G
(Z) (2) becomes. However, G(Z) is the result of Z-transforming the reception baseband filter, and α _R is α _R =e ^j2 〓 ^fR/fS = e ^j2 〓 ^/N(R+f ₀ ^T) .

Here, β _R (Z) is defined as follows. That is, β _R (Z)=X _R (Z)+jX _R+L (Z) X _R+L (Z)+jX _R (Z)...When R is an even number...When R is an odd number...(3) (1) , (2) and (3), β _R (Z) = Y (α ^-1 _R Z) G (Z) (4). Furthermore, Y(Z) and G(Z) are demultiplexed as follows. That is, Y(Z)= _N-1 〓 ⁿ⁼⁰ Z ^-n Y _o (Z ^N ) (5) G(Z)= _N-1 〓 ^m=0 Z ^m G _n (Z ^N ) (6) (4 ), (5), and (6), β _R (Z)= _N-1 〓 ⁿ⁼⁰ α ⁿ _R Y _o (α ^-N ₀ Z ^N ) _N-1 〓 ^m=0 Z ^mn G _n (Z ^N ) (7) Here, β _R (Z) is a high-speed sample value sequence,
The original PAM signal is a data sequence every T seconds, and the transmitting side selectively splits multiple PAM signals into T/2
Considering that a second delay is added, it can be seen that it is sufficient to extract sample values every T/2 seconds from the high-speed sample value series of β _R (Z). Now β _R
If the sample value taken every T/2 seconds from (Z) is β〓 _R (Z ^N/2 ), then from equation (7), β〓 _R (Z ^N/2 ) = _N/2-1 〓 ⁿ⁼⁰ α ⁿ _R Y _o (α ^-n _O Z ^N ) [G _o (Z ^N ) + Z ^N/2 G _o+N/2 (Z ^N )] + _N-1 〓 〓 ^n=N/2 α ⁿ _R Y _o (α ^-N ₀ Z ^N ) [Z ^-N/2 G _NN/2 (Z ^N ) + G _o (Z ^N
)] (8) Expression (8) as β〓 _R (Z ^N/2 )= _N-1 〓 ⁿ⁼⁰ α ₀ W ^Rn _N η〓 _o (Z ^N/2 ) (9) However, W _N = e ^j2 〓 ^/N , If rewritten as, the desired complex sample value sequence {β _R
(Z ^N/2 )} ^L _R=0 is {η〓 _o (Z ^N/2 )} ^N-1 _o=0 is subjected to N-point discrete Fourier transform and then multiplied by the frequency offset term α ⁿ ₀ (hereinafter, this operation is (referred to as offset Fourier transform) can be obtained as an output.

The conventional digital processing receiver for orthogonal multiplexed signals disclosed in Patent Application No. 150238 of 1972 is based on the above principle.

The disadvantages of the above method are as follows. First, as can be seen from equation (10), to obtain the input for the N-point Fourier transform, {Y _o (α ^-N ₀ Z ^N )} ^N/2-1 _o=0 is filtered by a low-pass filter {G _o (Z _N
)+
Z ^N/2 G _o+N/2 (Z ^N )} ^N/2-1 The output passed through _o=0 is the input for the first N/2 points, and {Y _o (α ^-N ₀ Z ^N )} ^{N /2-1} _o=0 is delayed by T/2 seconds and the output is passed through {G _o (Z ^N ) + Z ^N/2 G _o+N/2 (Z ^N )} ^N/2-1 _o=0 It must be input at N/2 points in the second half, and requires a T/2 second switching control circuit.

Furthermore, as can be seen from equation (9), an N-point offset Fourier transformer is required even though originally only outputs of N/2 points or less are required.

The principle of the present invention will be explained below.

Equation (8) can be expressed as follows.

β〓 _R (Z ^N/2 )= _N/2-1 〓 ⁿ⁼⁰ α ⁿ _R Y _o (α ^-N/2 _R Z ^N/2 ) G _o (Z ^N/2 ) (11) However, Y _o (Z ^N/2 )=Y _o (Z ^N )+Z ^-N/2 Y _o+N/2 (Z _N ) (12) G _o (Z ^N/2 )=G _o (Z ^N )+Z ^N/2 G _o+N/2 (Z _N ) (13) Here, it is expressed as α ₀ = W ^m+ 〓 _N ). However, m is a non-negative integer, and γ is a number in the range 0γ<1.

At this time, β〓 _R (Z ^N/2 )= _N/2-1 〓 〓 ⁿ⁼⁰ W ^(R+m+ 〓 ⁾ⁿ _N Y _o (W ^-Rm- 〓 ₂ Z ^N/2 ) G _o (Z ^{N /2} ) (14)
Therefore, for R where 0R<N/4, β〓 _2R-n (Z ^N/2 ) = _N/2-1 〓 ⁿ⁼⁰ W ^Rn _N/2 W〓 ⁿ _N Y _o (W ^- 〓 ₂ Z ^{N /2} ) G _o (Z ^N/2 ) (15) β〓 ^* _2R+1-n (Z ^N/2 )= _N/2-1 〓 〓 ^q=0 W ^(N-2R-2)n _N W ^(1- 〓 ⁾ⁿ _N Y _o (W ^-1+ 〓 ₂ Z ^N/2 ) G _o (Z ^N/2 )
(16) However, the symbol * means taking the complex conjugate of each response. further here If we define a _R (Z ^N/2 ), then (all signals except β ₀ , β ₁ , ..., β _L-1 are dummy signals), when 0R<N/4, a _R (Z ^N/2 ) = _{N/ 2-1} 〓 ⁿ⁼⁰ W ^Rn _N/2 W ^1/2n _N ξ _o (Z ^N/2 ) G _o (jZ ^N/2 ) (18) When N/4R<N/2 a _R (Z ^{N /2} ) = _N/2-1 〓 ⁿ⁼⁰ W ^Rn _N/2 W ^1/2n _N ξ ^* _o (Z ^N/2 ) G _o (jZ ^N/2 ) (19) However, ξ _o (Z ^{N/ 2} )=W ⁽ 〓 ^-1/2)n _N Y _o (W ^1/2- 〓 ₂ Z ^N/2 ) (20) Here {G _o (jZ ^N/2 )} ^N/2-1 _{o= 0} is N/ where the effective bandwidth on one side is 1/2T and only the linear phase gradient is different in steps.
Since the two low-pass filters {G _o (Z ^N/2 )} ^N/2-1 _o=0 are frequency-shifted by W ^1/2 _N , it is a complex filter with an effective bandwidth of 1/T. It can be seen that this is a band filter group. From equations (18), (19), and (20), {a _R (Z ^N/2 )} ^N/4-
1 _R=0 is {ξ _o (Z ^N/2 )} ^N/2-1 _o=0 respectively complex band filter {G _o
(jZ ^N/2 )} ^N/2-1 N/2 point output obtained through _o=0
Obtained as the first N/2 point output when point offset Fourier transform is performed, {a _R (Z ^N/2 )} ^N/2-1 _R=N/4 is {ξ ^* _o (Z ^N/2 )} ^N/2-1 _o=0
N obtained by passing each through a complex band filter {G _o (jZ ^N/2 )} ^N/2-1 _o=0
/
It can be seen that when the two-point output is subjected to N/2-point offset Fourier transform, the latter half N/2-point output is obtained.

In particular, when γ = 1/2, equations (18), (19), and (20) are It can be expressed as follows.

a _R (Z ^N/2 ) = _N/2-1 〓 ⁿ⁼⁰ W ^Rn _N/2 W ^1/2n _N Y _o (Z ^N/2 ) G _o (jZ ^N/2 )
...(21) However, 0R<N/2-1 Therefore, in this case, the signal processing process is significantly simplified.

FIG. 2 is a block diagram showing a specific embodiment of the digital processing type reception system for orthogonal multiplexed signals according to the present invention. In Fig. 2, 12 is the input terminal, 1
3 is a sampler, 14 is a demultiplexing circuit, 15 ₀ ,
15 ₁ ,..., 15 _N/2-1 is the input terminal of the preprocessing circuit 16,
17 ₀ , 17 ₁ , ..., 17 _N/2-1 and 18 ₀ , 18 ₁ ,
..., 18 _N/2-1 is the output terminal of the preprocessing circuit 16, 19 is the first polyphase circuit, 20 is the second polyphase circuit, 21 ₀ , 21 ₁ , ..., 21 _N/2-1 is the first The output terminals of the polyphase circuit 19, 22 ₀ , 22 ₁ ,
..., 22 _N/2-1 is the output terminal of the second polyphase circuit 20, 23 is the first offset Fourier transformer,
24 is a second offset Fourier transformer, 25 ₀ ,
25 ₁ ,..., 25 _N/4-1 is the first N/4 point output terminal of the first offset Fourier transformer 23, 26 ₀ , 26
₁ ,...,26 _N/4-1 is the output terminal of the second half N/4 point of the second offset Fourier transformer 24, 27 is the post-processing circuit, 28 ₀ , 28 ₁ ,..., 28 _L-1 is the output terminal It is.

First, the received signal input to the input terminal 12 is sampled by the sampler 13 at a sampling frequency of 1/NT hertz, and becomes an actual sample value sequence Y(Z). This actual sample value series Y(Z) is converted to T/2 by the demultiplexing circuit 14 according to the above equations (5) and (12).
The sample value series every second {Y _o (Z ^N/2 )} is decomposed into ^N/2-1 _o=0, and the nth sample value series Y _o (Z ^N/2 ) is the input terminal 1
5 Entered in _o . (0n<N/2-1) In the preprocessing circuit 16, each Y _o (Z ^N/2 ) is subjected to a frequency shift of W ^1/2- 〓 ₂ , and then W ⁽ 〓 ^-1/2)n _N A sample value series ξ _o (Z ^N/2 ) expressed by equation (20) is generated. The sample value series ξ ₀ (Z ^N/2 ), ξ ₁ (Z N/2) obtained in this way are output to the output terminals 17 ₀ , 17 ₁ , ..., 17 N ^/ _2-1.
,
..., ξ _N/2-1 (Z ^N/2 ₎ is output, and at the same time, _the complex conjugate sample _value of each sample value ξ ^* ₀ (Z ^N/2 ), ξ ^* ₁ (Z ^N/2 )
,
..., ξ ^* _N/2-1 (Z ^N/2 ) is output. First polyphase circuit 19 and second polyphase circuit 20
are both N/2 complex band filters {G _o (jZ ^N/2 )}
^N/2-1 _o=0 , and an R _- ^th complex band filter _{G R (} jZ _{N /} ² ) is connected. The output of the first polyphase circuit 19 is input to a first offset Fourier transformer 23. The first offset Fourier transformer 23 performs N/2 point offset Fourier transform on the N/2 point input according to equation (18), and then converts the first half N/4 points (for convenience of explanation, N is ) The output is output to the output terminals 25 ₀ , 25 ₁ , ..., 25 _N/4-1 . On the other hand, the output of the second polyphase circuit 20 is input to the second offset Fourier transformer 24, and the second offset Fourier transformer 24 converts the output of the second polyphase circuit 20 into N/2 according to equation (19) for the N/2 point input. After performing point offset Fourier transform, the second half N/4 point output is sent to the output terminal 26 ₀ ,
Output to 26 ₁ ,...,26 _N/4-1 .

Therefore, the output terminals 25 ₀ , 25 ₁ , ..., 25 _N/4-1 have the sample value series a ₀ (Z ^N/2 ) expressed by equation (17),
a ₁ (Z ^N/2 )..., a _N/4-1 (Z ^N/2 ) are obtained, and _{a N / 4 (} ^ZN/2 ),
a _N/4+1 (Z ^N/2 ) and a _N/2-1 (Z ^N/2 ) are obtained, respectively. The post-processing circuit 27 uses the thus obtained a ₀ (Z ^N/2 ), a ₁
(Z ^N/2 ), …, a _N/2-1 (Z ^N/2 ) to T/2 according to the following
Complex PAM signal per second β〓 ₀ (Z ^N/2 ), β〓 (Z ^N/2 ), ...,
β〓 _L-1 (Z ^N/2 ) is generated at the output terminals 28 ₀ , 28 ₁ , ..., 2
8 Output each to _L-1 . {a _o (Z ^N/2 )} ^N/2-1 _o=0 {
β〓 _o
(Z ^N/2 )} The process of obtaining ^L-1 _o=0 is expressed as follows from equation (17).

For 0RN/4-1, β〓 _2R-n (Z ^N/2 ) = a _R (-jZ ^N/2 ) ... (22) β〓 _2R-1-n (Z ^N/2 ) = [a _N/2-R (−jZ ^N/2 )〕 ^* ……(23
) That is, the Pth sample value of β〓 _2R-n (Z ^N/2 ) is a _R
It is obtained by multiplying the Pth sample value of (Z ^N/2 ) by (−j) ^p , and the Pth sample value of β〓 _2R-1-n (Z ^N/2 ) is a _N/2-R (−j) ^p for the Pth sample value of (Z ^N/2 )
It is obtained by multiplying by , and then taking its complex conjugate.

As described above, in the digital processing receiver for orthogonal multiplexed signals according to the present invention, two N/2-point Fourier transforms are performed instead of one N-point Fourier transform. The total amount of multiplication required for one N-point complex Fourier transform is 4N ² , and the total amount of multiplication required for two N/2-point complex Fourier transforms is 4×(N/2) ² ×2=2N ² . Therefore, the amount of digital calculation in the digital processing receiving apparatus for orthogonal multiplexed signals according to the present invention is significantly reduced compared to the conventional one. Furthermore, as can be seen from FIG. 2, the digital processing receiver for orthogonal multiplexed signals according to the present invention does not require a T/2 second switching control circuit. Therefore, it is possible to avoid complicating the device.

In addition, in the special case where γ=1/2, ξ _o (z ^N/2 ) is the demultiplexed input real sequence from equation (20).
Since it is equal to Yn(z ^N/2 ), the preprocessing circuit 16 shown in FIG. 2 becomes unnecessary. Also, equations (18) and (19) are reduced to equation (21), and {a _R (Z ^N/2 )} ^N/2 _R=0 is the second
N/2 of the first offset Fourier transformer 23 in the figure
Since it will be obtained all at once as a point output,
The second polyphase circuit 20 and second offset Fourier transformer 24 in FIG. 2 are also eliminated. FIG. 3 shows another embodiment of the digital processing receiving apparatus for orthogonal multiplexed signals according to the present invention obtained in this manner, and the meanings of each reference number correspond to those in FIG. 2.
However, in FIG. 3, all N/2 point outputs of the offset Fourier transformer 23 are effectively used, so the output terminals are expanded to 25 ₀ , 25 ₁ , ..., 25 _N/2-1. .

In the above description, all Fourier transforms are forward Fourier transforms, but this can be replaced by inverse Fourier transforms by taking the complex conjugate of input and output.

As described above, according to the present invention, it is possible to obtain a digital processing receiving apparatus for orthogonal multiplexed signals with a small amount of digital calculations and a simplified circuit scale, and its practical use is extremely large.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing a transmitting and receiving system for orthogonally multiplexed QAM signals, and FIGS. 2 and 3 are block diagrams showing a specific embodiment of a digital processing receiving apparatus for orthogonally multiplexed signals according to the present invention. In the figure, 2 ₀ , 2 ₁ , ..., 2 _L-1 are L T/2 second delay circuits, 3 ₀ , 3 ₁ , ..., 3 _2L-1 are 2L transmission baseband filters, 4 ₀ , 4 ₁ ,...,4
_2L-1 is 2L converters, 5 is a multiplexing circuit, 6 is a transmission line, 7 ₀ , 7 ₁ , ..., 7 _2L-1 is 2L demodulators, 8 ₀ ,
8 ₁ ,..., 8 _2L-1 are 2L receiving baseband filters, 9 ₀ , 9 ₁ ,..., 9 _L-1 are L T/2 second delay circuits, 10 ₀ , 10 ₁ ,..., 10 _L-1 is L number of T
seconds delay circuit, 13 is a sampler, 14 is a demultiplexing circuit, 16 is a preprocessing circuit, 19 is a first polyphase circuit, 20 is a second polyphase circuit, 23
24 is a first offset Fourier transformer, 24 is a second offset Fourier transformer, and 27 is a post-processing circuit.

Claims (1)

[Claims] 1. 2Ls synchronized with each other with a clock cycle of T seconds
A signal processing process in which L complex carriers with frequencies f ₀ , f ₁ , ..., f _L-1 are orthogonally modulated using baseband data (where L is a positive integer) and the outputs are orthogonally multiplexed is N/ T hertz (N is an even number,
When performing sampling at a sampling rate of N≧2L),
In an orthogonal multiplexed signal digital processing receiving device that receives an orthogonal multiplexed signal generated by setting the parameter γ, which is the fractional part of the product f _{0 T} of T and the minimum carrier frequency f 0 , to 0.5, the input terminal The received signal obtained in T/N seconds (however, N=
2L) and outputs N/2 point actual sample values every T/2 seconds;
Input two actual sample values and effective bandwidth is 1/T
Only the linear phase gradient differs stepwise in Hertz N/
A polyphase circuit consisting of two complex band filters and a T/2 filter connected to the polyphase circuit.
It includes an offset Fourier transformer that performs an N/2 point offset Fourier transform every second, and a post-processing circuit that performs a constant frequency offset correction on the N/2 point outputs of the offset Fourier transformer, and an orthogonal multiplexer. 1. A digitally processed receiver for orthogonal multiplexed signals, characterized in that signal demodulation is performed by digitally processed signals. 2 2L synchronized with each other with a clock period of T seconds
Reception in which orthogonal multiplexed signals obtained by orthogonally modulating L complex carriers with frequencies f ₀ , f ₁ ..., f _L-1 using baseband data (where L is a positive integer) are digitally processed. In the device, the received signal obtained at the input end is sampled every T/N seconds (N is an even number), and the signal is sampled every T/2 seconds by N/2.
A desired frequency offset is applied to the point actual sample values to output a first complex signal group a ₁ , a ₂ , ..., a _N/2 , and a second complex signal group b which is the conjugate complex signal of the complex signal. ₁ , b ₂ , ..., b _N/2 and a preprocessing circuit that inputs the first complex signal group a ₁ , a ₂ , ..., a _N/2 and has an effective bandwidth of 1/T hertz. A first polyphase circuit composed of N/2 complex band filters whose only linear phase gradients differ stepwise, and the second complex signal group b ₁ , b ₂ , ..., b _N/2 are input. a second polyphase circuit having the same configuration as the first polyphase circuit; and a first offset Fourier transformer connected to the first polyphase circuit and performing an N/2 point offset Fourier transform every T/2 seconds. , a second offset Fourier transformer connected to the second polyphase circuit and having the same configuration as the first offset Fourier transformer, and a first N/4 point output of the first offset Fourier transformer; The second half N/4 of the second offset Fourier transformer
A digitally processed receiving device for an orthogonal multiplexed signal, characterized in that it includes a post-processing circuit that performs a certain frequency offset correction on a total of N/2 point outputs consisting of point outputs, and performs demodulation of the orthogonal multiplexed signal by digital processing. .