JPS6335170A - Power source device - Google Patents

Abstract

PURPOSE:To permit the coincidence of the waveform of an input current with the waveform of an input voltage substantially with a simple constitution and stabilize an output voltage, by a method wherein the closing period of a switching element is determined by the value of a D.C. voltage or a load current while the opening period of the same is determined in accordance with the value of a full-wave rectifying voltage. CONSTITUTION:A commercial power source 1 is rectified by a full-wave rectifier 2 and the voltage of the commercial power source 1 is converted into a D.C. voltage by a chopper circuit 9 equipped with a switching element 7, thereafter, the voltage is supplied to a load 4. An ON period determining circuit 11, setting the closing period of the switching element 7, determines the closing period by a D.C. voltage Vo or a load current and an OFF period determining circuit 12 determines the opening period of the switching element 7 by a full-wave rectifying voltage Vi. When the D.C. voltage increases, the closing period is controlled so as to be shortened and when the instantaneous value of an input voltage increases, the opening period is controlled so as to be shortened.

Description

[Detailed description of the invention] 【Technical field】

The present invention relates to a power supply device using a chopper device that inputs commercial power and outputs a constant DC voltage.

[Background technology]

There are many ways to input a commercial power supply and obtain a constant DC voltage, but in particular, a voltage comparable to the effective value of the commercial power supply, for example 50
A series regulator as shown in FIG. 6 is available as a method for outputting a voltage of V to several hundred volts. That is, in FIG. 6, the commercial power supply 1 is rectified by the full-wave rectifier 2,
A DC voltage is supplied to the load 4 via. However, in this case, the loss of the transistor as a voltage adjustment element is large, and it is not used most of the time.Therefore, with the recent increase in the speed of switching elements, a high-speed chopper type power supply as shown in Fig. 7 has been developed. The device has become widely used. In FIG. 7, a commercial power supply 1 is rectified by a full-wave rectifier 2, connected to a load 4 via an inductance element 5 and a reverse current prevention diode 6, and a switching element 7 is connected to a control circuit 8.
controlled by. However, since the commercial power supply 1 is once accumulated in the smoothing capacitor C1, the waveform of the input current Is becomes as shown in Figure 8, which has many harmonic components, a low input power factor, and Since the peak value is very high, the current withstand capacity of the full-wave rectifier 2 must also be made larger than necessary. Therefore, there is a device shown in FIG. 9 that achieves low distortion and high input power factor by eliminating the smoothing capacitor and making the input current waveform as close as possible to the input voltage waveform. In FIG. 9, a commercial power source 1 is full-wave rectified by a full-wave rectifier 2 to obtain a pulsating voltage Vi. A chopper device 9 is configured by an inductance element 5, a reverse current prevention diode 6, a smoothing capacitor C2, and a switching element 7, and a DC stabilized voltage V0
is converted into and supplied to load 4. DC stabilization voltage■. is detected and the well-known PWM control circuit 10 adjusts the on/off duty cycle of the switching element 7 to obtain a stabilized DC voltage V. becomes stable. C0 is a capacitor that integrates high frequency components. The principle of the chopper device 9 is as shown in FIG. That is, when the switching element 7 is on, the equivalent circuit becomes as shown in FIG. After the switching element 7 is turned on for a time ton, the peak value of IQ during the time when it is turned off is Ip, which is accumulated in the inductance element 5. Here, Vi is the rectified Vs, so Vi=Vplsinωltl
((ALL is the angular frequency of Vs) and changes over time. On the other hand, the switching frequency f of the switching element 7
, is sufficiently high compared to ωL/2π, and IL is integrated by the smoothing capacitor C1, so the waveform of the input current Is eventually corresponds to the envelope of Ip. These waveforms are shown in Figure 11 (a
) to (d). The PWM control circuit 10 is ■. The on-time ton of the switching element 7 is varied depending on the magnitude of Vi, but since Vo is stable, ton can be considered to be approximately constant over the entire range of Vi. Therefore, since Ip is proportional to Vi and proportional to l sinωLtl, Is approximately matches sinωLt. Energy in the inductance element 5 is accumulated when the switching element 7 is on, and when the switching element 7 is off, the energy in the inductance element 5 is accumulated in the load 4 and the capacitor C2
However, the off-time tof of the switching element 7
If f is insufficient, the residual energy of the inductance element 5 will be added when the next switching element 7 is turned on, and 1. , Is become as shown in FIG. 11 (elf) and do not match sinωlt, so it is necessary to consider tof f. The equivalent circuit when the switching element 7 is off is as shown in FIG. 10B, and the following IQ flows through the inductance element 5. In order for the energy of the inductance element 5 to be completely released during the off-period tof of the switching element 7, it is sufficient to satisfy I □(toff)≦0 in the above equation, so that the following equation is obtained. Therefore, at any moment when tof f is Vi (1
) The PWM control circuit 10 may be set to satisfy the equation (2), but if the step-up ratio is low in (2). When approaches Vp, tOf
f has to be increased. Therefore, the above-mentioned conventional example has the following drawbacks. That is, it is necessary to make toff sufficiently large with respect to ton, and when Vi is small, 1. As a result, the integrative ability of capacitor C1 decreases and IL
is less likely to become a sine wave, and in order to transmit sufficient energy, it is necessary to reduce L of the inductance element 5, so Ip increases, the withstand capacity of the switching element 7 increases, and the speed of the switching element 7 increases. Due to the relationship, the absolute value of ton cannot be made small, and in the end fH will be lowered, and in extreme cases, fH may fall into the audible range.
Furthermore, in this countermeasure, if a fast switching element 7 is used, there is a drawback that the cost increases. As a method of compensating for the above-mentioned drawbacks, there is a method shown in FIG. 12 (see British Patent No. 2022943). According to this method, the current of the inductance element 5 is detected, and the voltage of this and Vi is divided. Since the off timing of the switching element 7 is determined by comparing the obtained value, the input current Is can be a sine wave. Furthermore ■. In order to stabilize
■. is detected, and based on this value, the partial pressure ratio of the previous Vi is made variable to control the machining. However, Vi, ■. It is necessary to detect not only the machining, but also the machining, which has the drawback that its control is quite complicated.In addition, since the envelope line of 19 matches Vl, the machining s is a sine wave in any case, but switching Since the on and off frequencies of the element 7 are kept constant, there is a drawback that energy may remain in the inductance element 5.

[Purpose of the invention]

An object of the present invention is to provide an inductance element current that has a simple configuration, allows the input current waveform to substantially match the input voltage waveform, stabilizes the output voltage, and avoids energy accumulation in the inductance element. To provide a control means for a power supply device that does not require an unnecessarily long suspension.

[Disclosure of the invention]

Figure 1 is a basic block circuit diagram of the present invention.
is full-wave rectified by a full-wave rectifier 2 , converted into a DC voltage by a chopper circuit 9 equipped with a switching element 7 , and supplied to a load 4 . The on-period determining circuit 11 that sets the closing period of the switching element 7 determines the closing period based on the DC voltage V0 or the load current, and the off-period determining circuit 12 determines the closing period of the switching element 7 using the full-wave rectified voltage ■i. The period is determined, and if the DC voltage further increases, the closing period is shortened, and if the instantaneous input voltage value increases, the closing period is shortened or controlled in the opposite direction. This will be explained in detail below using examples. X11L In FIG. 2, a commercial power supply 1 is full-wave rectified by a full-wave rectifier 2 to obtain a pulsating voltage Vi. A chopper device 9 is configured by the inductance element 5, the reverse current prevention diode 6, the smoothing capacitor C2, and the switching element 7, and supplies DC voltage to the load 4. The on-period determining circuit 11 determines the closing period of the switching element 7, and the off-period determining circuit 12 determines the closing period. The waveforms of each part in FIG. 2 are as shown in FIGS. 3(a) to 3(h). The operating principle of the chopper device 9 is the same as that of the conventional example. now,
When the output Q of the RS flip-flop FF is High and Q is Low, the switching element 7 is turned on and the current I
Q begins to play. At the same time, the charge/discharge control switch S is turned on, the capacitor C2 is discharged via the resistor R3, and the output of the voltage comparison element CPI becomes Low.At the same time, the charge/discharge control switch S2 is turned off, and the capacitor C3 is connected to the power supply Vcc.
Charging is then started via resistor R2. The DC voltage v0 is divided by resistors R3 and R3, and the amplifier A2, resistors R,
, the power supply ■"2 constitutes an inverting amplifier, and the VA2 output is -■
. Voltage Vc4 of capacitor C4, which outputs a voltage proportional to
The switching element 7 remains on until the voltage reaches VA2. When Vc4≧VA2, the output of the voltage comparison element CP2 becomes High, the RS flip-flop FF is reset, the output Q becomes Low and ζ becomes High, the switching element 7 is turned off, and the current ID begins to flow. At the same time, charge/discharge control switch S2 is turned on, and capacitor C4
is discharged via the resistor R1°, the output of the voltage comparison element CP2 becomes Low, and at the same time, the charge/discharge control switch SI is turned off, and the capacitor C2 starts charging from the power supply Vcc via the resistor RI. The pulsating voltage Vi is divided by resistors R5 and R4, and the amplifier A1, resistors Rf, R7', R7'', and power supply VrI constitute a non-inverting amplifier, and the VAI output outputs a voltage proportional to Vi, and the capacitor C3 The switching element 7 remains off until the voltage Vc3 reaches VAI. When Vc3≧VA, the output of the voltage comparison element CP+ becomes High, the RS flip-flop FF is set, and the switching element 7 is turned on. The on/off state of the switching element 7 is controlled by repeating the above operation over 0. That is, ① is detected and converted to VA2, and to is determined by this and the charging curve of the capacitor C1.
Determine n, detect Vi and convert it to vAI, and determine tof by this and the charging curve of capacitor C. Also, in both AI and VA□, Vi and V are non-invertingly amplified and invertively amplified by amplifiers A5 and A2, respectively, so that when Vi goes up, VAI goes up, and when Vo goes up, VA2 goes down. Furthermore, since the charging curves of capacitors C1 and C1 are constant, as Vi increases, VAI increases, and since the intersection with Vc3 becomes later, toff increases, and (2). As VA2 increases, the intersection with Vc4 becomes earlier and ton
becomes smaller. According to equation (1) in the prior art example, when Vi is high, toff is increased and Vi
This shows that when tof is low, it is sufficient to make tof f small, and the first embodiment satisfies this, so it is possible to avoid reducing unnecessary IL pause periods as in the conventional example, and moreover,
ton without affecting the Is waveform. Since it is controlled by ■. The output is stable and the configuration is simple. Figure 4 shows another embodiment of the present invention, and the waveforms at various parts are as shown in Figures 5(a) to 5(e). Assuming that the output Q of the RS flip-flop FF is High, the switching element 7 is turned on and the current flows. begins to flow. At the same time, in the current mirror circuit composed of current mirror transistors ST, Sr1, resistor R1, and capacitor C1, the voltage V e 3 of capacitor C1 begins to rise at a constant slope. DC voltage v0 is divided by resistors R,,R2, and the output VA of the inverting amplifier composed of amplifier A, power supply Vr, and resistor R3 outputs a voltage proportional to -Vo.VA and Vc3 are connected to voltage comparison element CP2. The switching element 7 is turned on until Vc, , reaches VA. Vc, -VA
Then, the output of the voltage comparison element CP2 becomes High, the RSS flip-flop 1F is reset, the output Q becomes Low, the switching element 7 is turned off, and ID begins to flow. V c 3 and VAI at voltage comparison element CP +
The switching element 7 is turned off until vc3 reaches VAIV, and when it reaches Vc3-VA+■T, the output of the voltage comparison element CP becomes High, the RS flip-flop FF is set, and the output Q is At the same time, the charge/discharge control switch S is turned on to discharge the capacitor C1, and the voltage comparison element CP1 immediately becomes Low, so the charge/discharge control switch S3 is also turned off and the above operation is repeated. Embodiment 2 is a buck-boost chopper, where I Q(t)=Vi/L-t IO(t)=I p-V. /L-tI p=Vi/L-ton toff≧Vi/Vo·ton If toff is proportional to Vi, it is possible to completely eliminate IL pauses. In this example, since Vc3 is a perfect straight line, it is a suitable example in which toffocVi can be easily realized.

【Effect of the invention】

As described above, the present invention determines the closing period of the switching element based on the DC voltage value or the load current so that the input voltage waveform substantially matches the commercial power supply waveform, and determines the closing period according to the full-wave rectified voltage value. Since the input current waveform can be made to substantially match the commercial power supply waveform with a simple configuration, the output voltage can be stabilized, and the inductance element current stopper can be used to avoid energy accumulation in the inductance element. There is no need to unnecessarily lengthen the DC voltage value, and since the closing period of the switching element is shortened when the DC voltage value increases and lengthened when the DC voltage value decreases, the DC voltage value can be made more stable. be.

[Brief explanation of the drawing]

Fig. 1 is a basic block circuit diagram of the present invention, Fig. 2 is a circuit diagram of one embodiment of the invention, Fig. 3 is an operation time chart of the same as above, and Fig. 4 is a circuit diagram of another embodiment of the invention. , Fig. 5 is an operation time chart of the same as above, Fig. 6 is a circuit diagram of the conventional example, Fig. 7 is a circuit diagram of another example of the same as above, Fig. 8 is a voltage and current waveform diagram of the same as above, and Fig. 9 is a diagram of the same as above. FIG. 10 is a circuit diagram of another example of the conventional example, FIG. 10 is an operation explanatory diagram of the same as above, FIG. 11 is a voltage and current waveform diagram of the main part of the same as above, and FIG. 12 is a circuit diagram of still another example of the conventional example. In the figure, 1 is a commercial power supply, 2 is a full-wave rectifier, 4 is a load, 5 is an inductance element, 6 is a backflow prevention diode, 7 is a switching element, 9 is a chopper device, 11 is an off-period determining circuit, and 12 is an off-period It is a decision circuit. Agent Patent attorney Ishi 1) Chief 7th 1st figure 2nd figure 6th figure 7th figure 8th figure 9th figure IU figure 10th figure 11th figure 12th figure

Claims (2)

[Claims] (1) A commercial AC power source is full-wave rectified, converted to DC voltage by a chopper device consisting of a switching element, an inductance element, and a backflow prevention diode, and the input current waveform is The closing period of the switching element is determined by the DC voltage value or the load current so as to substantially match the commercial power supply waveform, and the closing period is determined according to the full-wave rectified voltage value. power supply. (2) The power supply device according to claim 1, wherein the closing period of the switching element is shortened when the DC voltage value increases and lengthened when the DC voltage value decreases.