JPS6321433B2 - - Google Patents

Description

[Detailed Description of the Invention] The present invention relates to a thyristor motor control device,
In particular, the present invention relates to a control device that performs constant output control using control advance angle β and field current control.

The output P of a thyristor motor can be expressed as the product of the DC machine brush-to-brush equivalent voltage V _S and the main circuit DC equivalent current I _d , as shown in equation (1), ignoring the loss inside the motor. can.

P=V _S ×I _d ………(1) However, E _t : Motor terminal voltage β : Effective control advance angle u : Load side commutation overlap angle I _M : Motor current In addition, the equivalent voltage between DC machine brushes (hereinafter referred to as _VS ) is shown in equation (2). It is expressed as a function of the terminal voltage E _t and the control advance angle β, and the terminal voltage E _t is a function of the angular velocity ω and the field current I _f , as shown in the following equation (4), and E _t = K (β, I _d ) × ω × I _f (4) where K ( _β , I _d ): constant ω: motor rotational angular speed If: field current.

Therefore, as a conventional constant output control method using a thyristor motor, a control device as shown in FIG. 1 has been proposed. This is to control the control advance angle β and the field current If so that V _S remains constant regardless of the angular velocity _ω .

In FIG. 1, the main circuit of the thyristor motor is composed of a thyristor forward converter 1, an inverse converter 2, a motor 3, a current transformer 4, and a voltage detection transformer 8. Thyristor converter 1
9. Consists of a field current detector 20. The control circuit includes a current detector 5 that controls the main circuit current;
A current control system on the power supply side consisting of a current control circuit 6, an automatic pulse phase shifter 7 on the power supply side, and a V _S command setting device 2 that controls the control advance angle β so that V _S is constant.
1. A load-side control system consisting of a V _S detector 9, a V _S control circuit 10, a basic β _O command circuit 11, and a load-side automatic pulse phase shifter 12, and a field so that the terminal voltage E _t is constant. A field control system consisting of an _Et command setting device 22 for controlling the magnetic current, an _Et detector 15, a terminal voltage _Et control circuit 16, a field current control circuit 17, and an automatic pulse phase shifter 18 for the field. It is configured. Second
The figure illustrates the relationship between V _S , E _t , β _O , and If with respect to the motor rotational angular velocity ω when constant output control is performed using the conventional control method shown in _FIG . 1. As is clear from Fig. 2, when the angular velocity ω is higher than the base angular velocity ω _B , as ω increases,
I _f is weakly controlled so that E _t is constant, and the set control advance angle β _O is largely controlled so that V _S is constant. Therefore, between ω _B and ω _T , V _S is constant and I _d is constant regardless of the angular velocity, resulting in a constant output. However, in the vicinity of zero angular velocity, that is, in the initial startup region, there is no back electromotive force, so β _O = 0 degrees is fixed, and commutation margin angle is gradually secured from a predetermined speed after startup (usually about 5 to 10% speed). Therefore, between zero speed and base low speed ω _B , the DC machine _brush voltage V _S does not become a voltage proportional to the speed, but the second
It has the disadvantage of being stepped, as shown by V _S in the figure.
Here, the base speed is the rated speed without field weakening.

By the way, in the case of a thyristor motor, the armature reaction and the commutation overlap angle are large, and these change depending on the main circuit current and field current. The situation will be explained with reference to FIGS. 3 and 4.

FIG. 3 shows changes in the armature reaction angle δ and the commutation overlap angle u with respect to the main circuit current.
The solid line in the figure indicates that the field current I _f is strong (I _f −full), and the dotted line indicates I _f
The case of weak (I _f −week) is shown. As the main circuit current I _d increases and as the field current I _f decreases, δ and u become larger.

In addition, in the case of a system that uses the back electromotive force of the motor to cause commutation, such as the thyristor motor that is the object of the present invention, as is well known, the commutation margin angle (β-u)
There is a problem in that the turn-off time of the thyristor must be ensured. Therefore, it is necessary to determine the setting control advance angle β _O such that β−u maintains a predetermined value under the conditions of the maximum current used and the minimum field current.

FIG. 4 is a vector diagram explaining the change in δ due to the main circuit current and field current. FIG. 4a shows a case where the main circuit current changes. Under the conditions of no-load magnetic flux Φ _O and no-load induced voltage E _O , if the main circuit current I _M1 is advanced by β _O - u/2 relative to E _O , the magnetic flux will decrease due to demagnetization due to armature reaction. Φ ₁ and the terminal voltage drops to E _t1 . When the main circuit current is further increased to I _M2 , the armature reaction angle changes from δ ₁ to δ ₂
The magnetic flux Φ ₂ terminal voltage E _t2 becomes even smaller. FIG. 4b shows the effect of δ and terminal voltage E _t as the field current changes. No-load magnetic flux Φ ₀ ,
The main circuit current I _M1 is applied to the no-load induced voltage E _O in an advanced phase of β _O -u/2, and an armature reaction angle δ ₁ is generated, resulting in a state of magnetic flux Φ ₁ and terminal voltage E _t1 . If the field current is weakened (week) from this state, the no-load magnetic flux will become Φ 0 ' and the no-load terminal voltage will become Φ ₀ ', as shown by the dotted line.
Even if the amount of _{demagnetization} is the same for the same main circuit current I _M1 , the armature reaction angle becomes large, δ ₃ , and the magnetic flux Φ ₃ and terminal voltage E _t3 become small.

Therefore, in order to perform constant output control in a thyristor motor, field weakening control is necessary, but in this case, as mentioned above, the influence of armature reaction is large, and these must be taken into consideration.

However, in the conventional control method shown in Fig. 1,
The effects of changes in the main circuit current and field current on u, δ and terminal voltage _Et are not taken into consideration. β _O is controlled only by changing u, δ, and E _t and, as a result, changing V _S . This is β-u
The disadvantage is that the response is slow when it comes to controlling β _O to ensure . In particular, in the field weakening region, there is a possibility of commutation failure due to insufficient commutation margin angle β-u.

In addition, as shown in equation (2), V _S varies depending on the elements of E _t and β, that is, u and δ, but in the conventional method shown in Figure 1, the variation of E _t due to armature reaction is taken into consideration. Because this is not the case, it may work in the opposite direction as a control for β _O. In order to secure β-u by increasing the current, β _O must be increased, but as E _t decreases,
Detects a decrease in V _S and as a result reduces β _O to reduce V _S
It operates to keep the value at a predetermined value. As described above, such a control method cannot cope with a sudden increase or decrease in current.

Furthermore, if an attempt is made to deal with the above problem by increasing the setting β _O , the power factor and efficiency will deteriorate and the converter capacity will increase. Furthermore, if an attempt is made to manufacture an electric motor in which the effects of u and δ are small, the size and weight of the electric motor will become extremely large, which is uneconomical.

The purpose of the present invention is to eliminate the drawbacks of the conventional method, compensate for the fluctuations of u and δ due to the main circuit current and field current, and the fluctuations of _Et due to armature reaction, and compensate for the power factor and
It is an object of the present invention to provide a thyristor motor control device capable of efficient and economical constant output control.

The present invention focuses on the fact that in order to create an economical device with the best power factor and efficiency, considering equation (2), the commutation margin angle β-u must always be minimized, and the main circuit current , field current, and motor angular velocity ω to predict u and δ, and determine the optimal setting control advance angle β _O and terminal voltage E _t to ensure that β−u is at a minimum and to keep V _S constant.
is calculated in advance and given as a command value.
The variation of V _S is characterized in that β _O is corrected by a closed loop control system of V _S to keep V _S constant.

The armature reaction angle δ is obtained from the vector diagram in Fig. 4 using equation (5), and the commutation overlap angle u is obtained using equation (6).
It can be obtained from the formula.

δ＝tan ^-1 X _q・I _M cos(β _O −u/2)/E _O −X _d・I _M sin(
β _O −u/2)……(5) However, X _d : d-axis synchronous reactance X _q : Q-axis synchronous reactance X _C : Commutation reactance Therefore, in advance,
Determine the change pattern of V _S and terminal voltage E _t , then select u and δ such that β-u is minimum (2), (5),
The β _O command value can be determined by repeatedly calculating equation (6).

If E _t and β _O obtained in the above calculation are given as command values, V _S becomes almost constant, but manufacturing errors and
A highly accurate constant output control system can be realized by providing a closed loop control system that controls V _S to a constant value in consideration of setting errors, and correcting the β _O command using the output of this V _S control circuit.

Next, one embodiment of the present invention will be described. FIG. 5 shows an embodiment of the present invention.

In FIG. 5, main circuit components and current control circuit 6, power supply side automatic pulse phase shifter 7, load side automatic pulse phase shifter 12, V _S detector 9, E _t detector 15,
The E _t control circuit 16, the field current control circuit 17, and the field automatic pulse phase shifter 18 are the same as those shown in FIG. 1 described above, and will not be particularly described here.

The circuit configuration that characterizes the present invention is such that the main circuit current command, field current I _f , and angular velocity ω signals are input, and the optimum β _O command obtained by the above calculation is output _.
The command calculation circuit 23 outputs the terminal voltage _Et command according to the angular velocity ω and the main circuit current command. The _Et command calculation circuit 24 takes the angular velocity ω as input and outputs the V _S command.
It is composed of a V _S command circuit 25 and a V _S control circuit 26 that corrects the β _O command so that the V _S detected value becomes the V _S command. Specifically, the β _O command calculation circuit 23 is configured as shown in FIG. 6, for example. This figure shows an example of a configuration including function generators 29, 29', 29'' and a multiplier 30.

The output of the β _O command arithmetic circuit 23, that is, the β _O command, is generated by the no-load β _O command generator 27 and the loaded β _O command generator 2.
Output as the sum of 8. The no-load β _O command generation unit 27 inputs the signals of the angular velocity ω and the field current I _f ,
A reference β _O command value that provides a predetermined V _S at no-load is output according to ω and _If . Under load β _O command generation section 28
is determined by the main circuit current command and field current I _f signals,
A β _O command correction value that compensates for the increase in u and δ due to armature current and maintains a predetermined V _S is output. The functions of the function generators 29, 29', 29'' are the terminal voltage E _t and the field current I _f with respect to the main circuit current and angular velocity ω.
After determining the amount of change in , β, u, and δ are obtained in advance by repeatedly calculating equations (2), (5), and (6). The function of β _O can be determined from β, u, and δ obtained in the above calculation.

The E _t command calculation circuit 24 receives the motor angular velocity ω and the main circuit current command as input signals. A no-load terminal voltage E _t command as shown in FIG. 7a is output in accordance with the angular velocity ω. The dotted line in the figure shows the characteristics when the main circuit current I _d is increased. That is, the voltage drop due to armature reaction in response to an increase in main circuit current is predicted, and the terminal voltage E _t command is lowered in accordance with the current command. The terminal voltage E _t command in Fig. 7a is increased in proportion to the motor angular velocity ω between ω _B and ω _T , and between 0 and ω _B.
Considering changes in β _O and If, _{an E t command} is given as a function of ω so that V _S is proportional to ω.

The V _S command circuit 25, as shown in FIG. 7b,
Between 0 and _ωB , the command is proportional to ω, and between _ωB and _ωT , a constant rated V _S command is output.

The characteristic curves of I _f and β _O shown in Fig. 7a and b are as follows:
The β _O command calculation circuit 23 and the E _t command calculation circuit 2
4. Changes in I _f and β _O when the output of the E _t control circuit 16 is controlled at constant output by each command signal.

Next, FIG. 6 shows the operation of the control device shown in FIG.
This will be explained using FIGS. 7a and 7b. Since β _O is set to 0 degrees at the initial stage of starting the motor, that is, between 0 and ω ₁ speeds, a constant field current _If flows. Speed ω ₁ ~
During ω _B , β _O is gradually increased to a predetermined value required for load commutation, but during this period, the E _t command is created such that V _S changes linearly, so the E _t control is A signal for increasing I _f is output from the circuit 16, and I _f increases. Between the speeds _ωB and _ωT , the rate of change of the _Et command with respect to the speed decreases, so that _If is weakened as the speed increases. (The weakening command is also the same as E _t
It is output from the control circuit 16. ) By weakening the field current, the function generators 29' and 29'' outputs in the β _O command calculation circuit 23 operate to increase the β _O command, thereby compensating for the armature reaction at the time of field weakening. ing.

The above explanation is about the operation under light load, but the case where the load increases will be explained below. The main circuit current command (I _d command) increases due to the increase in load, but due to the increase in the I _d command, the output of the function generator 29 in the β _O command calculation circuit 23 increases, making the β _O command larger. This operation ensures that the commutation margin angle is always maintained. Furthermore, as the load current increases, the armature reaction increases and the terminal voltage E _t decreases. This effect is particularly large in the field weakening region, making it difficult to keep V _S constant. Therefore, in this device,
The change in β _O due to the increase in load current and the change in δ accompanying it are calculated in advance, and the E _t command is corrected by the current command so that the terminal voltage E _t at that time is obtained. E _t as the corrected E _t command
Control circuit 16 results in an increase in _If . In this way, in the ω _B to ω _T speed region (field weakening region), by increasing the field current in accordance with the increase in load current, it is necessary to reduce the influence of armature reaction and make the control advance angle too large. There isn't. Therefore, stable operation with good power factor and efficiency is possible even during overload.

Note that the above explanation assumes that the settings of β _O and E _t are ideally matched. In reality, V _S does not reach the desired value due to setting errors in β _O and E _t and differences in u and δ due to motor manufacturing errors. Therefore, the fifth
In the present invention shown in the figure, the β _O command is corrected by the V _S control circuit 26.

In one embodiment of FIG. 5, the β _O command calculation circuit 23 inputs and determines the main circuit current command, field current, and angular velocity, but instead of the field current, the terminal voltage
The purpose of the present invention remains the same even if the E _t signal is input and the β _O command is generated.

In addition, in the embodiment shown in Fig. 5, a DC type thyristor motor consisting of a forward converter and an inverse converter is used as the main circuit converter, but an AC type thyristor motor using a cycloconverter as the converter is used. However, the purpose of the present invention can be similarly achieved.

As described above, according to the present invention, even when the current suddenly increases, it is possible to stably control the phase to the minimum required level without commutation failure. Therefore, it is possible to realize an economical constant output control device with good power factor and efficiency. Furthermore, since the control advance angle β can be minimized, the torque ripple generated by the electric motor is also reduced, and the present invention can be applied to various drive machines that require constant output control.

[Brief explanation of drawings]

Figure 1 is a configuration diagram of a conventional constant output control circuit.
Fig. 2 is a diagram showing the output characteristics of each part with respect to the motor angular speed when controlled by the conventional method, Fig. 3 is a diagram for explaining changes in β, u, and δ with respect to the main circuit current and field current, and Fig. 4 Figures a and b are vector diagrams for explaining changes in δ and E _t due to armature reaction, Figure 5 is a circuit configuration diagram showing one embodiment of the present invention, and Figure 6 is the inside of the β _O command calculation circuit. A diagram showing an example of the configuration, FIG. 7a shows an _Et command, which is a feature of the present invention
FIG. 7b, which is a diagram showing the β _O command, is a diagram showing output characteristics of each part with respect to angular velocity when controlled according to the present invention. 1... Thyristor forward converter, 2... Thyristor inverse converter, 3... Motor, 4... Current transformer, 5...
Current detector, 6... Current control circuit, 7... Power supply side automatic pulse phase shifter, 8... Transformer, 9... DC machine brush-to-brush equivalent voltage (V _S ) detector, 12... Load side automatic Pulse phaser, 13...Position detector, 14
...Pilot generator, 15...Terminal voltage (E _t )
Detector, 16...Terminal voltage control circuit, 17...Field current control circuit, 18...Automatic pulse phase shifter for field, 19...Thyristor for field, 20...Field current detector, 23 ... Setting advance phase angle (β _O ) command calculation circuit, 24 ... Terminal voltage (E _t ) command calculation circuit,
25... DC machine brush-to-brush equivalent voltage (V _S ) command circuit, 26... V _S control circuit.

Claims (1)

[Claims] 1. A thyristor converter that converts power supplied from an AC power supply, a motor that rotates upon receiving power from the thyristor converter, and a field that supplies field current to the field windings of the motor. a current control means for controlling a power supply side firing phase angle of the thyristor converter according to a motor current command and a current deviation of the motor current; a phase control means for controlling a motor-side firing phase angle; a field current control means for controlling a field current by controlling a firing phase angle of the field thyristor converter according to a field current command; Inputting the motor current command, field current, and motor speed, predicting the armature reaction and commutation overlap angle, calculating a control advance angle that can secure a commutation margin angle, and applying the control advance angle command to the phase control means. means for inputting the motor current command and motor speed, and terminal voltage command means for outputting a terminal voltage command of the motor according to the input amount; A terminal voltage control means that outputs a magnetic current command and applies it to the field current control means, and a DC machine brush-to-brush equivalent voltage that increases in proportion to the motor speed up to the base speed and remains constant from the base speed to the maximum speed. DC machine brush-to-brush equivalent voltage command means for outputting a command; and DC machine brush-to-brush equivalent voltage command means for correcting the set control advance angle applied to the phase control means in accordance with the deviation between the command value and the detected value of the DC machine brush-to-brush equivalent voltage. A thyristor motor control device comprising voltage control means.