JPS61156914A - Variable phase shift circuit - Google Patents

Abstract

PURPOSE:To set a relative phase difference in a phase output signal to a prescribed phase difference by synthesizing outputs signals in terms of vectors obtained in response to voltage drops across a resistor and a capacitor at output terminals of the 1st and 2nd differential amplifier circuits. CONSTITUTION:The relative phase between voltages e31 and e32 outputted from polyphase circuits 22A, 22B is controlled variably from the initial state of a phase difference of 90 deg. by a drive control current Icon in a direction increasing or decreasing the phase difference by applying phase control to the combined voltages e31, e32 by a common drive control current Icon in reverse directions to each other. The relative phase between detection signals SA, SB is adjusted by an amount as required by controlling a phase shift amount setting element 34 of a drive current source control circuit 31. The absolute value of the output voltages e31, e32 of the phase shift circuits 22A, 22B is changed respectively when voltages eOR1, eOR2 are changed accordingly, but the ratio of the absolute values is kept always to a constant value. Thus, the variable phase shift circuit with simple constitution is realized easily.

Description

[Detailed Description of the Invention] [Industrial Application Field] The present invention relates to a variable phase shift circuit that measures displacement, such as length and angle, using a magnetic scale (referred to as a magnetic scale). This is suitable for application to a displacement measurement device.

[Conventional technology]

As this type of displacement measuring device, the conventional
The method described in Publication No. 17 has been proposed. As shown in FIG. 9, this displacement measuring device measures wavelength λ (for example, λ=
A scale signal consisting of a rectangular wave or a sine wave of 200 μm is recorded in advance, and the magnetic scale 1 is attached to the object to be measured. A pair of magnetic heads 2A and 2B are arranged on this magnetic scale 1 with an interval expressed by the following equation.

1, = fi + □λ ・・・・・・(1)
However, n represents the wave number of the scale signal recorded on the magnetic scale 1 between the heads 2A and 2B.

Here, the magnetic heads 2A and 2B are moved by the relative displacement of the magnetic scale 1.
A so-called magnetic field-responsive magnetic head is used that can generate detection signals SA and SB that respond to the strength of the magnetic field of the scale signal recorded at each recording position when the recording position of the scale signal facing B is displaced. It will be done. As described in Japanese Patent Publication No. 50-25817, this type of magnetic head has a method in which a carrier signal of a predetermined frequency is passed through an excitation coil wound around the saturable core portion of the magnetic core. A configuration may be used in which the detection signals SA and SB are obtained by modulating the generated magnetic flux with the magnetic field picked up from the magnetic scale 1 by a detection coil.

The detection signals SA and SB of the heads 2A and 2B are converted into detection voltage signals e1 and e= expressed by the following equations in the output amplifier circuits 3A and 3B of the phase detection circuit 5, respectively.

2π e + = E, sin -x cosωotλ (2) 2π e, = E, cos -x sinωotλ (3) The detected voltage signals e and e2 are a composite circuit 4, and is sent out as a detection output signal e3 expressed by the following equation.

In addition, in equations (2)2 to (4), E1 to E3 are amplitudes and ω. is the carrier angular frequency, and X is the relative displacement amount of the magnetic scale l with respect to the helads 2A and 2B.

In this way, detection signals el and e2 having a phase difference of 90'' are generated from the pair of heads 2A and 2B, and by combining these signals, a carrier frequency having a phase determined by the relative displacement ilx is detected. By determining the phase of the detected output signal e based on equation (4), the relative displacement X (therefore the displacement of the length, angle, etc. of the object to be measured ) can be measured.

By the way, in the displacement measuring device having this configuration, the relative positional interval between the pair of magnetic heads 2A and 2B is determined by the scale signal 9.
If positioning cannot be performed with high accuracy at a position with a phase difference of 0@ minutes, the detection voltage signal e with a phase difference of 90 degrees will be incorrect.
, and e2 cannot be obtained, so in principle, it is inevitable that an error will be mixed into the detection output signal e obtained based on equation (4). Therefore, a pair of heads 2A and 2B
The distance between them must be positioned with high precision at a position corresponding to a phase difference of 90°.

However, since the spacing between the heads 2A and 2B is determined by mechanical positioning accuracy, it is not possible to make the accuracy that high. To compensate for this, conventionally, a CR circuit is used as shown in FIG. A method is used in which a variable phase shift circuit 11 is provided. That is, one output terminal of the output amplifier circuits 3A and 3B is connected in common and connected to the output terminal T1, and a series circuit of a capacitor CI and a variable resistor R1 is connected between the other output terminals, and the common connection point is connected to the output terminal T1. is connected to the second output terminal T2.

In this way, the outputs of the output amplification circuits 3A and 3B are combined via the capacitor CIO and the variable resistor RIO, respectively, and the detection output signal e3, which is the vector sum thereof, is sent between the terminals T1 and T2. Ru. Therefore, by adjusting the variable resistor R1, the phase of the detection output signal e3 can be adjusted.

[Problem that the invention seeks to solve]

However, according to the configuration shown in FIG. 10, if the impedance of the voltage source constituted by the output amplifier circuits 3A and 3B is not suppressed to a sufficiently low value, the phase of the detected output signal e will be rotated by the variable phase shift circuit 11. As a result, there is a problem that the signal level of the detection output signal e3 fluctuates.

In order to solve this problem, it is considered that the impedance of the voltage source, that is, the output impedance of the output amplifier circuits 3A and 3B, should be selected to a low value. A new problem arises in that a sufficient output voltage cannot be obtained.

Therefore, when adjusting the amount of phase shift using the configuration shown in FIG. 10, in practice, when the output level of the voltage detection signal e fluctuates due to the adjustment of the phase shift, a complicated operation is required to readjust the output level each time. By doing this, the necessary voltage gain was ensured. For example, as shown in FIG. 11, the output amplifier circuits 3A and 3B are connected to a common IC,
2, the variable phase shift circuit 11 is provided as an external component, and a variable resistor 13A for DC balance of the output amplifier circuits 3A and 3B is provided as another external component.
and 13B, and a variable resistor 14 for gain adjustment of the voltage detection signal e1 are provided, and if the output level changes when the variable resistor RIO of the variable phase shift circuit 11 is adjusted, the DC balance variable resistors 13A, 13B and By adjusting the gain adjustment variable resistor 14, the fluctuation of the output level is suppressed as much as possible.

However, when this adjustment is performed, the phase of the voltage detection signal e3 shifts, so it is necessary to adjust the variable resistor RIO of the variable phase shift circuit 11 again. The need arises.

As described above, in practice, in the past, correction conditions that were practically satisfactory were found by trial and error through complicated adjustment work.

The present invention has been made in consideration of the above points, and by making it possible to adjust the phase of the voltage detection signal without changing the output level, it is possible to easily move the voltage detection signal without the need for complicated adjustment work. This paper attempts to propose a variable phase shift circuit with a simple configuration that allows phase adjustment.

[Means for solving the problem] In order to solve the problem, in the present invention, a two-phase output signal having a predetermined phase difference is produced by adjusting the relative phase difference between the two-phase input signals SA and SB. In a variable phase shift circuit that sends out e3 and e3□, two-phase input signals SA and S
First and second phase shift circuits 22A, 2 each receiving B
2B, and a drive current source control circuit 31 that generates a drive control signal ICon that controls the drive currents of the first and second phase shift circuits 22A and 22B. , 22B each have a first transistor Q2, C21 whose base is supplied with an input signal, and is driven by a drive current supplied from a controllable current source Q8, C27. 25 and a first transistor Q5, C whose base is supplied with a reference voltage
24, driven by constant current #Q9, C28 drive current, resistors and capacitors (RZI and CZI), (RZ2 and C20
) are connected in series, and one end of the series circuit is connected to the first transistors Q2 and C of the first differential amplifier circuit 23.25.
21, and the other end is connected to the bases of the first transistors Q5 and C24 of the second differential amplifier circuit 24.26, and the resistors RZ1 and RZ2 and the capacitor CZ
I, CZ2 connection midpoint is connected to the first and second differential amplifier circuits (
23 and 24), the second transistors (Q3 and Q4) (25 and 26), and a phase shift element connected to the bases of (C22 and C23), the drive current source control circuit 3
1 to a drive control signal I obtained in response to the phase shift amount setting output of the phase shift amount setting element 34. 9 is supplied to the controllable current sources Q8, C27 of the first and second phase shift circuits 22A, 22B, and in the first and second phase shift circuits 22A, 22B,
First and second differential amplifier circuits (23 and 24), (25
By vector-synthesizing the output signals obtained corresponding to the voltage drops of the resistors RZI, RZ2 and the capacitors CZI, C20 at the output terminals of 26) and 26), the relative phase difference of the phase-shifted output signals e and e3t is set to a predetermined value. Set it to phase difference.

[Effect]

In the first and second phase shift circuits 22A and 22B, two-phase input signals SA and SB are transmitted through resistors RZI and RZ2 and capacitors CZI and C20 as phase shift elements, respectively.
Flowing currents il and 12 corresponding to , an output signal is generated at the output terminal of the differential amplifier circuit 23.25, which is a vector on the real axis corresponding to the voltage drop V, 11, (eORI% - eORI), (
eo, IZ, eORI2) are obtained, and a capacitor CZI,
Voltage drop V of C20. Output voltage (eOcl, -〇〇CI) represented by a vector on the imaginary axis corresponding to VCt
, (eOC2, e.ct) is obtained. The outputs of the first and second differential amplifier circuits represented by vectors on the real axis and vectors on the imaginary axis obtained in this way are selected as necessary and vector-combined with each other to obtain phase-shifted outputs e31 and e. , g.

The second differential amplifier circuit 24.26 is a constant current source Q9, C28
The first differential amplifier circuit 2 is driven by
3.25 are simultaneously driven and controlled by a drive control signal ICON of a drive current source control circuit 31 provided in common. Therefore, the phases of the output signals e31 and e3t sent out from the first and second phase shift circuits 22A and 22B vary according to the drive control signal, and thus the phase shift output signals e3 and e3
The relative phase difference between z can be set as necessary.

Accordingly, the first and second phase shift circuits 22A and 2
2B are controlled by the common drive control signal I CON, the ratio of the absolute values of the phase-shifted output signals e31 and ezz can always be maintained at a constant value, so that the phase-shifted output signals e3. and e3□, it is possible to eliminate the need to readjust the levels of the phase shift circuits 22A and 22B each time the phase settings of e3□ and e3□ are changed.

〔Example〕

An embodiment of the present invention will be described in detail below with reference to the drawings.

FIG. 1 shows the variable phase shift circuit as a whole, and in the same manner as described above with respect to FIG. 9, a first phase shift circuit 22A receives the detection signal SA of one magnetic head 2A as a voltage input.
and a second magnetic head receiving the detection signal SB of the other magnetic head 2B.
It has a phase shift circuit 22B.

The phase shift circuit 22A includes a pair of differential amplifier circuits 23 and 24, and is connected to the differential amplifier circuit 23 via an input transistor Q1.
The base of the transistor Q2 of the differential amplifier circuit 24 receives the detection signal SA, and the base of the transistor Q5 of the differential amplifier circuit 24 receives the reference voltages v, I of the reference power supply 25 via the reference voltage transistor Q6.

The base of the transistor Q3, which forms a differential pair together with the transistor Q2 of the differential amplifier circuit 23, is connected to the base of the transistor Q3 of the differential amplifier circuit 24.
It is connected to the base of a transistor Q4 which forms a differential pair with a transistor Q5.

In addition, a series circuit of resistors RZI and czi that operates as a phase shift element is connected between the emitter of the input transistor Ql and the emitter of the reference voltage transistor Q6, and the midpoint of the connection is connected between the emitter of the input transistor Ql and the emitter of the reference voltage transistor Q6. connected to the base.

As a result, transistor Q2 constituting one differential pair
and the output voltage e corresponding to the voltage drop VMI caused by the current i flowing through the resistor RZI on the collector side of Q3. III % eORI and the transistors Q4 and Q that constitute the other differential pair.
On the collector side of 5, an output voltage C6Cl, e., 1 corresponding to the voltage drop V(H) caused by the current il flowing through the capacitor CZl can be obtained.

In the case of the first phase shift circuit 22A, the collectors of the positive phase side transistors Q3 and Q5 of each differential pair are commonly connected to the output terminal R3, and the output terminal T is connected from the common connection point to the output terminal T.
An output voltage 031 is derived with respect to OI.

Here, as the emitter voltage of the human power transistor Q1 changes with respect to the emitter voltage of the reference voltage transistor Q6 in accordance with the detection signal SA, the current 11 flowing through the resistors RZI and CZt that constitute the phase shift element changes to the detection signal of the head 2A. It can be expressed by an equivalent circuit as shown in FIG. 2 using signal SA as a voltage source. In other words, the configuration is such that resistors R and C, each consisting of resistors RZI and CZI, are connected in series to the voltage source e formed in connection with the head 2A, and this series circuit has the following formula: When a current i expressed by % flows, the value of voltage e can be determined as the vector sum of the voltage drops across the resistor R and the capacitor C, as shown by the following formula.

In equations (5) and (6), I8 is the amplitude of current i,
ω is the carrier angular frequency of the detection signal obtained from the head,
θ is the phase angle.

Here, the phase angle θ is expressed by the following formula (7), and as shown in FIG. 3, the voltage drop e11 of the resistor R
It represents the phase difference between the voltage e obtained by vector synthesis of the voltage drop -ec of the capacitor C and the voltage e on the real axis, and the phase angle -ec is determined by the values of the resistor R and the capacitor C.
θ is determined. For example, ωCR=1
If the values of the resistor R and capacitor C are selected as shown in (8), the value of the phase angle -θ will be 45°.

In this way, the phase shift circuit 22A generates an output signal e31 obtained by rotating the phase of the detection signal SA of the head 2A by one θ.
can be obtained.

The second phase shift circuit 22B has the same configuration as the first phase shift circuit 22A, and includes resistors RZ2 and C20 as phase shift elements.
The detection signal SB of the head 2B is supplied to the series circuit of the head 2B via the input transistor Q20, and changes in the detection signal SB are performed based on the reference voltage □ of the reference power supply 25 supplied from the reference voltage transistor Q25. Transistors Q21 and C22, C23 and C24 that form a differential pair output voltage corresponding to the corresponding drop voltage
It is designed to be generated on the collector side.

In the second phase shift circuit 22B, the collector of the positive phase side transistor Q22 of one differential pair and the collector of the negative phase side transistor Q23 of the other differential pair are commonly connected to the load resistor R26. The point is connected to the output terminal TO2. Therefore, as shown in Figure 4, at the output terminal TO2, the phase of the output voltage e is only 1,000, since the curvature of the output voltage ec on the capacitor side is reversed compared to the case in Figure 3. There will be a phase shift. In this way, a phase output e3□ whose phase is rotated by +θ in the opposite direction to that of the phase shift circuit 22A is obtained from the second phase shift circuit 22B.

In addition to the above configuration, transistors Q4 and Q5 forming one differential pair of the phase shift circuit 22A are supplied with a constant current from a constant current source transistor Q9, and a constant current is supplied from a constant current source transistor Q9 to the transistors Q4 and Q5 forming one differential pair of the phase shift circuit 22A. Transistor Q forming a differential pair
A constant current is supplied to 23 and C24 from a constant current source transistor Q28.

On the other hand, transistors Q2 and Q3 constituting the other differential pair of the phase shift circuit 22A are supplied with drive current from the transistor Q8 as a controllable current source, and Transistor Q forming the pair
A drive current is supplied to 21 and Q22 from a transistor Q27 as a controllable current source. A drive current source control circuit 3 provided in common to the phase shift circuits 22A and 22B is provided at the bases of these transistors Q8 and Q27.
A control signal ICON is supplied from 1 to 1.

The drive current source control circuit 31 includes a constant current circuit 32 including a diode D31, and a controllable current source 33 connected in series to the constant current rgJ path 32, the connection point of which is connected to the transistors Q8 and Q27. The controllable current source 33 sends out a variable current Iv corresponding to a control signal supplied from a variable resistor 34 provided as a phase shift amount setting element. Among them, the amount of change that occurs based on the constant current flowing through the constant current circuit 32 is calculated by the transistors Q8 and Q.
By supplying the base of transistor Q
2, ■, 2 and IZI flowing to Q3, Q21, and Q22
It is designed to control the

Thus, transistor pair Q of phase shift circuits 22A and 22B
2, Q3 and Q21. Drive current supplied to Q22 1.
and 11. is controlled by the common drive current source control circuit 31, so that the real axis component e of the output signals effl and e3□. , lI and e. The value of NZ will be controlled by the same amount.

In addition, in FIG. 1, 41 indicates phase shift circuits 22A and 22B.
42 and 43 are bias power supplies for supplying detection signals SA and SB to the phase shift circuits 22A and 22B, respectively.

In the above configuration, as an initial condition, tz=I+z (9
), and similarly in the phase shift circuit 22B I z+
= Izz ・= ・= (1
0), drive current rz+ and 1. Select.

Moreover, the phase shift elements RZ1, c of the phase shift circuits 22A and 22B
zi and NZ2, CZ 2(7) relationship as ωCz+Rz+
= 1 ・=”(11)ω
C2□R2□=1...
...(12), the output signal e3. (Figure 3)
The signal components e0□ and -〇. The absolute values of R1 are equal to each other, Ieo□ 1±113oci...
- The relationship (13) can be obtained. Also, phase shift circuit 2
2B output signal'! 32 (FIG. 4) signal component e. R1
The absolute values of and e Oe2 are equal to each other, l eoy=t l = l 6ocz l
−・”(14) is obtained.

By the way, the gain G in each differential pair constituting the differential amplifier circuit is ra! It is expressed as Here, r8 is an emitter resistance of a transistor constituting each differential pair, RL is a load resistance connected to the collector side, RLX is a conversion resistance, and ■ is a drive current flowing through the collector-emitter.

In equation (15), it can be expressed as h 6 Therefore, from equation (17), it can be seen that the gain G of the differential amplifier circuit constituting each differential pair is proportional to the drive current ■.

However, in FIG. 1, since the drive currents 11□ and ■2□ of the differential pair on the capacitor side of the phase shift circuits 22A and 22B are constant, their gains are maintained at a constant value, so the output voltages -e oc+ and e. cz maintains a constant value.

On the other hand, the drive currents I11 and 1□ on the resistance side are the drive control currents I of the drive current source control circuit 31. 8, the gain of the differential pair is controlled by the drive control current ICON, and as a result, the output voltage e
. □ and e. .

The value of is variably controlled.

If we examine this relationship in Figures 3 and 4, we get
In the phase shift circuit 22A, the voltage -〇〇cl is constant, whereas the voltage e0□ is the drive control current ■. 8, the angle -0 formed between the composite voltage e31 and the real axis is variably controlled in the negative direction.

On the other hand, in the phase shift circuit 22B, the voltage e oc
While z is constant, the voltage e. By variably controlling R1 with the drive control current ICON, the phase θ of the composite voltage 632 is drive-controlled in the positive direction.

In this way, the combined voltage es+ and e32 are the common drive control current I. 9, the relative phases of the voltages e31 and e3□ output from the phase shift circuits 22A and 22B are equal to the drive control current ■.

By turning ON, the phase difference is variably controlled from an initial state with a phase difference of 90 degrees to a direction in which the phase difference increases or decreases.

(The relative phases of the detection signals SA and SB input to the phase shift circuits 22A and 22B, respectively, are adjusted by an amount as required by controlling the phase shift amount setting element 34 of the drive current source control circuit 31.) Can be adjusted.

When performing such adjustment, the absolute values of the output voltages eff+ and e3□ of the phase shift circuits 22A and 22B are respectively the voltage e. R1 and e. , I2 changes accordingly, but the ratio of their absolute values is always maintained at a constant value. Incidentally, the absolute values of output voltages e31 and e3□ are (17
) Based on the formula, the gain of the differential pair on the resistor side of the phase shift circuits 22A and 22B is a common drive control current (2). 8, the voltage e on the resistor side. R
1 and e. The absolute value of R2 will change in the same proportion, thus the voltage e. R1 and e. . The ratio of the absolute values of the output voltages eil and esz is always maintained at a constant value, and as a result, the relative ratio of the absolute values of the output voltages eil and esz is always maintained at a constant value.

According to experiments, the drive control current ■,. 9, the phase of the two-phase output signal can be changed by the amount of phase shift shown in Fig. 5, whereas the change in gain at this time is as shown in Fig. 6, in practical terms. We were able to obtain results that maintained a constant value.

The fact that the relative ratio of the absolute values of the two-phase output signals e11 and e3t can be maintained at a constant value even though the relative phases change in this way means that the output voltages e31 and e from the phase shift circuits 22A and 22B, respectively.

To variably control the phase of the two-phase signal obtained as
This means that there is no need to readjust the gain of each phase shift circuit 22A and 22B. Therefore, when converting the variable phase shift circuit configured as shown in FIG. As shown in FIG. 7, since it is no longer necessary to provide a gain adjustment element for the phase shift circuit as an external component, the number of pins of the IC 41 can be significantly reduced.

Moreover, if configured as shown in FIG. 1, the drive current source control circuit 3
Output voltage e31 whose phase changes in accordance with the control output of 1
and e3t, by configuring the resistor and capacitor as phase shift elements by a differential amplifier circuit,
Since the circuit has a well-balanced configuration, it is possible to realize a variable phase shift circuit that operates stably with respect to temperature characteristics.

In the case of the embodiment shown in FIG. 1, a phase-shifted output of 1θ is obtained from the phase-shifting circuit 22A side, and ++ is obtained from the phase-shifting circuit 22B side.
Although the phase shift output of θ is obtained, if the outputs of the differential pairs of the phase shift circuits 22A and 22B are combined as necessary, the phase shift of +θ can be obtained, as shown in FIG. Since the circuits 42A, 42B and -θ phase shift circuits 43A and 43B can be configured, by using this, +θ phase shift output can be obtained from the phase shift circuit 22A side and the phase shift circuit 22B can be configured. Even if the −θ phase-shifted output is obtained from the side, the same effect as in the above case can be obtained.

〔Effect of the invention〕

As described above, according to the present invention, when the relative phase difference of two-phase input signals is set as required, the variable phase shifter can always maintain the relative ratio of the absolute values of the two-phase signals constant. Therefore, when setting the amount of phase shift, it is possible to easily realize a variable phase shift circuit with a convenient configuration that does not require complicated operations such as re-adjusting the gain.

[Brief explanation of drawings]

FIG. 1 is a connection diagram showing an embodiment of the variable phase shift circuit according to the present invention, FIG. 2 is a connection diagram showing an equivalent circuit of the main part thereof, and FIG.
The figure and FIG. 4 show the output voltage e3. of the phase shift circuits 22A and 22B of FIG. and ezz signal components, Figures 5 and 6 are curve diagrams showing experimental results, and Figure 7 is a route map without showing the pin configuration when the configuration in Figure 1 is converted into an IC. , FIG. 8 is a block diagram showing another embodiment of the present invention, FIG. 9 is a block diagram showing a displacement measuring device to which the present invention can be applied, and FIG. 10 is a block diagram showing a conventional variable phase shift circuit. , FIG. 11 is a route diagram showing the pin configuration of a conventional IC. 22A, 22B... Phase shift circuit, 31...
- Drive current source control circuit, 41...constant current source.

Claims (1)

[Claims] In a variable phase shift circuit that outputs two-phase output signals having a predetermined phase difference by adjusting the relative phase difference between two-phase input signals, the first and second phase shift circuits; and a drive current source control circuit that generates a drive control signal for controlling drive currents of the first and second phase shift circuits; The phase circuits each include: a, a first differential amplifier circuit having a first transistor at its base supplied with an input signal and driven by a drive current of a controllable current source; and b, a reference at its base. a second differential amplifier circuit having a first transistor to which a voltage is supplied and driven by a drive current of a constant current source; and a resistor and a capacitor connected in series, and the series circuit one end connected to the base of the first transistor of the first differential amplifier circuit, the other end connected to the base of the first transistor of the second differential amplifier circuit, and the resistor and a phase shift element having a connection midpoint of the capacitor connected to the base of the second transistor of the first and second differential amplifier circuits, and the drive current source control circuit includes a phase shift amount setting element. and generates a drive control signal corresponding to the phase shift amount setting output of the phase shift amount setting element, and supplies this drive control signal to the controllable current sources of the first and second phase shift circuits. and obtaining the two-phase output signal by vector-synthesizing the output signals obtained corresponding to the voltage drops of the resistor and capacitor, respectively, at the output terminals of the first and second differential amplifier circuits. A variable phase shift circuit featuring: