JPS5942244B2 - Magnetic compass device and magnetic compass data transmission method that compensates for two-cycle errors - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to means for compensating for undesired errors or changes in the signal characteristics of a flux valve data repeater device, and more particularly to flux valve compass data repeat peaks. Fluctuations in device output (including errors due to variations in the horizontal component of the earth's magnetic field, indexing angle errors, and errors in cardinal heading and heading between cardinal headings).

) and a magnetic compass data transmission method.

Disturbances occur when maneuvering with flux valve magnetic compass devices at high latitudes due to the reduced strength of the horizontal component of the earth's magnetic field, especially at high latitudes.

Flux bulb magnetic compasses are typically arranged to sense only the horizontal component of the Earth's magnetic field.

As a result of high latitude, the strength of the sensed component is proportionately smaller, resulting in a compass device with reduced sensitivity, resulting in less accurate heading information.

Prior art devices, although in a relatively complex manner, generally incorporate amplifiers in separate channels of a data oscillator in an inverse relationship to the signal strength as measured by the flux valve itself. It was sought to solve this reinforcement problem by controlling the effective value of the gain or impedance to provide the input of the magnetic-compass data repeater independent of variations in the strength of the horizontal component of the earth's magnetic field.

These prior art techniques include U.S. Pat. - Automatic gain control scheme for mechanical transducers).

Although these prior art techniques have been useful in providing sufficient magnetic field compensation under most conditions, the compensation signal actually only compensates for the horizontal magnetic field component, and is typically associated with variations in circuit elements, such as temperature or power supply voltage. It does not completely compensate for gain changes due to drift or aging of elements.

Furthermore, the characteristics of the individual gain control elements of the separate channels of the data transmitter will change relative to each other without appropriate corrective adjustments, thus introducing a two-cycle transmission error into the automatic gain control stage. It ends up being a problem.

The improved scheme disclosed in U.S. Pat. No. 3,784,753 (Multiplexing Gain Control Scheme for Synchronized Data Transmission Apparatus) completely overcomes the drawbacks of the prior art by using a relatively complex and expensive correction circuit. are doing.

Although this scheme overcomes the shortcomings of the prior art, it introduces an undesirable two-cycle error in its relatively complex automatic gain control stage, and it does It was realized that there was a need for a simple method (while retaining the advantages of this patent) to uniformly vary the gain of .

Prior art devices compensate for cardinal heading errors in compass data transmission systems by using circuits that include precision differential synchronizers and dual potentiometers that do not introduce errors. was additionally requested.

According to the invention, the cost of obtaining such selected synchronizers and precision potentiometers is advantageously reduced.

Index angle errors were similarly corrected in prior art compass data transmission systems by employing high precision synchronizers and dual potentiometers of similar quality.

It would be highly desirable to be able to replace it with a simple, inexpensive circuit that would allow a single adjustment control for each of these correction purposes while maintaining a high degree of accuracy.

The present invention not only compensates for changes in the strength of the earth's magnetic field, but also uses a simple common automatic gain control in the circuit configuration that compensates for the effects of other error sources without introducing the errors of prior art schemes. The object of the present invention is, in part, to provide a means for correcting for undesirable changes in signal amplitude in multichannel flux valve data peaking devices.

The novel control circuit of the present invention monitors the data repeater control signal near the input of the device in use rather than simply monitoring the signal at the output of the flux valve.

By monitoring the input at the device used and by using the data repeater excitation voltage as the switching reference signal, the gain control circuit (which is part of a closed feedback loop) is sensitive not only to changes in operating latitude. It also compensates for gain changes caused by device parameter variations and other effects without introducing new errors.

In accordance with the present invention, electrically cross-coupled circuit means correct in the sine and cosine channels of the flux valve data transmission scheme to provide two-cycle cardinal error correction with only one adjustment setting. Do the following.

A similar adjustment (requiring only one adjustment) is used for correction of index angle errors.

In a variant of the compensator, the latter two compensations are achieved by a direct current signal applied linearly at a predetermined ratio to the inductive winding of the flux valve.

The present invention will be explained in detail below based on the drawings.

1A and 1B, the compensated compass device includes a magnetic orientation detector or flux valve 11. In FIGS.

The flux valve 11 may be as shown in commonly assigned US Pat. No. 2,852,859 (Flux Valve Compensator).

Other types of flux valves, such as U.S. Pat.
No. 73,610, No. 3,641,679 and US Pat.

Flux valve 11 is a 400 Hz AC signal source 2
is excited by

The AC signal source 2 is the excitation winding 12 of the flux valve 11
It is connected to the.

U.S. Patent Nos. 2,852,859 and 35,736
Flux valve 1 as described in No. 10
1 has three Y-connected dicurrent windings 13, 14 and 15 on one Y-shaped core, each winding being connected together to a common ground terminal F.

The respective terminals of windings 13, 14 and 15 opposite terminal F are labeled AB and C.

If necessary, terminals A, B, and C may be
A single cycle compensation signal from the No. 852,859 single cycle compensator (not shown) may be provided.

Terminal A of the flux valve 11 is connected to one winding 20 of a Scott-T transformer 21 via a blocking capacitor 16, while terminals B and C are connected to the Scott-T transformer via respective blocking capacitors 17 and 18. each end of the second input winding 22 of the device 21 .

Winding 22 has a center tap connected to the other end of winding 20.

As is well known, the signal outputs of windings 13, 14 and 15 have a frequency twice that applied to excitation winding 12.

The double frequency cosine output on winding 23 of transformer 21 and the double frequency sine output on winding 27 are provided to current servo loop 31 .

The current servo loop 31 is further supplied with the output of the frequency doubler 29 via a lead 29a.

Since the frequency doubler 29 is excited by the AC signal generator 2, the output on lead 29a is at a frequency of 800 Hz, which is used as a reference signal for the current servo loop 31.

Current servo loop 31 provides an output to leads 32 and 33, which outputs are respectively , the sine and cosine ( ) of the magnetic heading of the aircraft (m sinψ and HmC
It is a DC signal whose amplitude is proportional to O3ψ).

Thus, the horizontal component of the earth's magnetic field sensed by the flux valve windings 13, 14 and 15 is divided into sine and cosine component values and then converted into a DC signal provided on leads 32 and 33 by a current servo loop. Ru.

As shown in U.S. Pat. No. 3,678,593, these DC signals are fed back to windings 13 and 15 of flux valve 11 via beams 10 and 10a to cancel the earth's magnetic field. work like that.

Details of this feedback arrangement and its advantages are described in the above-mentioned US Pat. No. 3,678,593.

That is, this patent details closed loop operation that provides a high precision output in the form of an analog DC output proportional to the sine and cosine of the aircraft's magnetic heading.

Therefore, the 800 Hz three-wire magnetic heading information provided by the horizontal magnetic field detector or flux valve 11 is proportional to the sine and cosine of the aircraft's heading by the combined structure of the Scott T transformer 21 and current servo loop 31. converted to a DC signal.

Therefore, the magnitude of the output at leads 32 and 33 is a function of the magnetic heading or heading of the aircraft and the strength of the horizontal component of the earth's magnetic field.

Variations in the magnitude of the sine and cosine outputs caused by changes in the magnetic field strength Hm affect only the output gradient (voltage/azimuth angle), which is a trigonometric function of the input magnetic heading angle ψ and the output voltage of the current servo loop 31. without changing the relationship
Therefore, the following equation is given.

■3゜=1S1nψ・・・・・・ （1）■3
3- to 1CO3ψ・・・・・・・・・ (2) Here,
K1 provides the gain of the current servo loop 31 and has dimensions in Volts/Oersteds.

The signals on leads 32 and 33 are input to an automatic gain control circuit 34.

This circuit 34 also receives certain feedback signals on leads 56 and 57.

As will be discussed below, these feedback signals are generated at the outputs of buffer amplifiers 52 and 53 after the output of automatic gain control circuit 34 has been processed by at least dual channel modulator 45.

In order to understand the operation of the gain control circuit 34, for convenience, we will briefly explain the index error compensator 37 and the two-cycle compensator 48.
I decided to ignore the existence of .

The most compact output of the compass device, which is supplied by leads 61, 62, and 63 to the aircraft controls or other equipment 64, is the same as the scales and leads 61, 62, 62, and A proportional voltage between pairs of such leads, such as between 63, 63 and 61, is usually required.

These may be for example 11.8V for each eye and must be maintained at a constant slope from meeting the required compass accuracy over a wide range of horizontal magnetic field strengths Hm.

The output of the flux valve 11 and therefore the output of the current servo loop 31 is the horizontal magnetic field strength (which of course varies according to latitude).
Automatic gain control stage 34 maintains the output signals at leads 61, 62, and 63 at the desired lag-to-lag constant slope of 11.8 V, respectively. is required.

For this purpose, the DC output at automatic gain control circuit 34 is fed to a known dual channel modulator 45.

A 400 Hz reference signal is supplied from the alternating current signal source 2 by each of the two individual channels of the dual channel modulator 45 by the beam board 2a.

Leads 46 and 47 therefore produce 400 Hz signals proportional to the sine and cosine of the aircraft's magnetic heading, respectively.

These equally amplified signals 54 and 55 are fed separately to buffer amplifiers 52, 53.
occurs in

Each of these leads 54 and 55 has a feedback
Leads 56 and 57 are connected.

Automatic gain control circuit 34 shown in more detail in FIG.
are output leads 54 and 5 of buffer amplifiers 52 and 53.
monitoring the slope at 5 and comparing this to a reference voltage level;
The gain of the entire system is then varied in response to the comparison by controlling the gain of automatic gain control circuit 34.

If the slope at the output of buffer amplifiers 52, 53 of FIG. increase to the level.

The outputs and voltage slopes of buffer amplifiers 52,53 are similarly controlled.

The signal level at the output leads 35 and 36 of the automatic gain control circuit 34 is optimally passed through the output short tee transformer 60.

The output of the transformer 60 is therefore completely independent of variations in the earth's magnetic field strength.

That is, ■3. -2S1nψ (3) and (3) 2CO8ψ (4) where K2 is a new proportionality constant.

The automatic gain control circuit 34 controls the unbalance (u
nbalance), which would result in periodic errors in the aircraft heading output data.

The individual gains of the sine and cosine channels are now controlled identically and no offset voltage is introduced into the DC signal representing the sine and cosine components of the aircraft magnetic heading.

As shown in more detail in FIG. 2, the DC signals at leads 32 and 33 are produced by the cooperative action of flux valve 11, Scott T-transformer 21, and current servo loop 31, with sinψ and current servo loops, respectively. COSψ
The amplitude is proportional to.

Output lead 32 connects dual channel modulator 4 in series via resistor 75, circuit point 76, resistor 77 and input lead 35.
Connected to one channel of 5.

A capacitor 78 whose other end is grounded is connected to the input lead 35, and together with the resistor 77 forms a low-pass filter.

Similarly, the second output lead 33 has a resistor 79 and a circuit point 80.
, connected in series via resistor 81 and input lead 36 to the second channel of dual channel modulator 45 .

A capacitor 82 whose other end is grounded is connected to the input lead 36, and together with the resistor 81, this capacitor forms a low-pass filter.

Switching or chopper transistors 83 and 84
are connected (at their respective collectors and emitters) between ground and circuit points 76 and 80, respectively, to control the current through the emitters and collectors according to the base voltage of the respective transistors.

The DC signals on leads 32 and 33 are choppered by transistors 83 and 84, respectively, and after being smoothed by low-pass filters 7718 and 81-82,
They are individually modulated in a dual channel modulator 45 by a 400 Hz reference signal on lead 2a.

These dual channel output voltages are supplied by Scots L-T transformer 6
0 is provided as three-wire synchro data to the flight control or other user device 64.

For the purpose of controlling the automatic gain control circuit 34, the lead 5
The same 400 Hz modulated output currents on 4 and 55 are provided to the constant amplitude variable phase circuit by leads 56 and 57, respectively.

This circuit consists of a resistor 95 and a capacitor 96 connected in series between the leads 56 and 57 and a circuit point 97, respectively.

This time f%95-96 was filed by the present applicant (this U.S. Patent No. 3
548,284 (Synchronized Data Transmission Apparatus with Discrete Gain Changes to Compensate for Undesirable Signal Slope Variations) and US Pat. No. 3,617,863 (Constant Amplitude Variable Phase Circuit).

The constant amplitude variable phase signal generated at circuit point 97 is rectified by diode 94 and inputted to one input (inverting input) of integral operation amplifier 92 as a variable unipolar voltage.

Amplifier 92 has a capacitor 91 connected between the one input and the output.

A second input (non-inverting input) of amplifier 92 receives a stable positive unidirectional reference voltage through resistor 93 from a preferred voltage source (not shown) connected to terminal 98.

As shown, the amplifier 92 and its associated circuitry include:
Comparing the output slopes of leads 54 and 55 with a constant level voltage at terminal 98 acts as a conventional comparator means providing an integrated output as a function of the difference between the two voltage levels at terminals 97 and 98.

The positive signal at the output of amplifier 92 is applied via resistor 88 to one input (non-inverting) of amplifier 87.

A 400 Hz excitation signal (from AC signal source 2) is applied to the other input of amplifier 87 via terminal 90a and resistor 89.

Amplifier circuit 87 receiving a variable amplitude DC input current and a constant amplitude AC input current operates as a known variable pulse width signal generator to provide a 400 Hz variable pulse width signal at circuit point 74. It is configured as follows.

This variable pulse width signal is passed from circuit point 74 in parallel through respective resistors 85 and 86 to chopper transistor 8.
3 and 84 to control the conduction versus non-conduction time relationship of these chopper or switching transistors.

Transistors 83 and 84 are synchronously conductive and both nonconductive for a controlled period of time depending on the pulse width of the amplifier 87 output.

As the non-conducting portion of the cycle is increased in time, the total current per cycle flowing from lead 32 to lead 35, for example, is increased.

In other words, the current flowing to ground at lead 32 is proportionally smaller.

In this manner, the voltage across leads 54 and 55 depends on amplitude variations in the total flux valve data and other disturbance factors in the signal channel between current servo loop 31 and buffer amplifiers 52 and 53. be made irrelevant.

Thus, the three-wire output provided by transformer 60 to use device 64 of FIG. 1 is maintained constant for each purpose from lag to lag, for example 11.8V.

The overall system shown in FIG. 1 (where the outputs V35 and VB2 of automatic gain control circuit 34 on leads 35 and 36
can be processed by the novel index angle error compensator 37 first, i.e. before being subjected to the modulation of 40 (Hlz).

For this purpose, the compensation circuit of FIG. 3 is used.

The index angle error compensated by circuit 37 is caused by the usual difficulty in achieving perfect alignment between the aircraft longitudinal axis and the electrical longitudinal axis of flux valve 11.

Therefore, index angle error compensation circuit 37 provides manual correction after installation of the device by performing essentially the same function provided by the relatively expensive differential servos used by some prior art devices. It is provided to make the person do the following.

However, since the installation accuracy is typically within 10 degrees above, the compensation function can be performed accurately by the relatively inexpensive circuit of FIG. 3 in which only one potentiometer shaft is used for adjustment.

It will be clear that the correction is made by the compensation circuit described below for the value of angle ψ when in trigonometric form such as sin ψ and COS ψ data.

Therefore, the apparatus of FIG.
2β sin ψ is internally generated at and -.

The value of K2S1nψ and the value of -2βcosψ are 2S1
0(ψ+β) according to well-known trigonometric identities.

Here, f=ψ+β can be used to represent the corrected value of ψ.

The value of 2βSjnψ is also added to the value of K2CO3ψ and - to form 2CO3(ψ+β).

In accordance with the technique of the present invention, the β term must be the same in both sine and cosine output channels for accurate compensation, and the same source for the β term must be present in these two channels of the circuit. used.

In operation, the circuit of FIG. 3 receives as its inputs at leads 35 and 36, respectively, negative valued DC voltages representing 2 sin ψ and 2 COS ψ.

These voltage signals are fed directly to the inputs of conventional unity gain output amplifiers 145 and 155, respectively, shown on the right side of the figure.

A third
It is used in other or main parts of the circuit shown in the figure.

For the purpose of providing compensation voltages to amplifiers 145 and 146, one 81nf term of lead 35 is connected to a conventional inverting amplifier 1.
03 to switching transistor 107.

Amplifier 103 connects its input terminal 101 via resistor 102.
a second input terminal (non-inverting) which has an output terminal 104 connected to the (inverting side) and is further grounded via a resistor 105;
have.

The -COS ψ term on lead 36 is connected directly to switching transistor 109.

Transistors 107 and 109 are alternately made sufficiently conductive and sufficiently non-conductive to first produce the output of amplifier 103 on lead 108a and then produce the signal through switching transistor 109 on lead 108b. It has become.

Both leads 108a and 108b are the potentiometer 11
3, so that the signals passed alternately through switching transistors 107 and 109 are connected to contact 11 for time-sharing purposes in amplifier 120.
3a alternately.

Switching transistors 107 and 109 are alternately rendered conductive under the control of a sinusoidal signal appearing on lead 2b.

This signal is conveniently obtained from a 400Hz signal.

Other stable frequency signals may be used instead of the signal from the AC signal source 2 of FIG. 1A.

In reality, the 400Hz signal on lead 2b is connected to lead 10.
6 is used to control the conduction of transistor 107.

Since the time division technique is used (here, the signal on lead 2b is transferred to lead 111, 180° phase shifter 112 and
is applied to switching transistor 109 via.

In this manner, the signals on leads 35 and 36 are alternately applied to selected contact positions of potentiometer 113.

Potentiometer 113 has terminals 113b and 113c connected to both terminals of operational amplifier 120.

The output terminal 121 of the amplifier 120 is connected to its terminal 113b (inverting terminal) via a resistor 115, and the other input terminal 113c is grounded via a resistor 114.

Therefore, the input to amplifier 120 is time-shared and terminal 121
Its output above is supplied to a second pair of switching transistors 122,123.

These transistors 122, 123 each have a resistor 1
26 and 127 to control the signals given to amplifiers 128 and 129.

The effective gain of amplifier 120 is varied according to the settings of a single controller 37a, whose control is based on conventional ground swinging operation.
is manually set according to the known magnitude of the index angle error determined as a result of (rat 1on).

The conduction of switching transistor 122 and the conduction of switching transistor 109 occur simultaneously.

In the same manner, switching transistor 123 is rendered conductive at the same time as switching transistor 107 is rendered conductive.

This operation occurs when the signal on lead 2b is applied directly to switching transistor 123 via lead 125.
This is achieved by controlling the conduction of 3.

The desired synchronous operation of switching transistor 122 is achieved by connecting transistor 12 from circuit 112 via lead 124.
This is accomplished by providing a 180° phase-shifted signal to 2.

In this manner, both channels of the circuit have a common amplifier 120
by time-sharing the use of , so that the same correction is applied to the two channels, i.e. the amount of the sine ring added to the cosine ring is equal to the amount of the cosine ring subtracted in the sine term channel. It is considered possible.

It will be appreciated that adjustment of the single controller 37a allows adjusting the potentiometer 113 so that both channels are set identically according to the magnitude of the index angle error.

The time-shared current passing alternately through switching transistors 122 and 123 is connected to unity gain amplifiers 128 and 12.
9 is given alternately.

Output terminals 132 and 1 of amplifiers 128 and 129
The signal at 33 is passed through resistors 141 and 150 to amplifier 1.
45 and 155 (the same input terminals connected to leads 35 and 36).

The outputs of amplifiers 145 and 155 may be smoothed by the action of suitable low pass filters to remove the 400 Hz modulation component from the outputs appearing on respective output leads 38 and 39.

In the illustrated embodiment, these filters include amplifier 1
28 and 129, and consists of resistors 126 and 127 and capacitors 128a and 129a, respectively.

The mathematical relationship representing the index angle error as a function of the sine and cosine components of the magnetic heading signal is as follows.

here, ψ′ − Compensated output.

ψ - uncompensated output.

β = Tangent component of index angle error.

Therefore, by adjusting the gain of amplifier 120 according to the value of β, equation (5) can be satisfied as follows.

The output of amplifier 128 is as follows.

■13□=-2βcosψ・・・・・・・・・・・・
(6) This is when transistors 109 and 122 are conducting.

Further, when transistors 107 and 123 are conductive, the output of amplifier 129 is as follows.

■, 33-1 2βS1nψ・・・・・・・・・・・・
. . . C7) The summation at amplifier 145 yields the output 38 (this signal).

V38 = K2 (Sinψ−βcosψ)−・・−
..., -... (8) On the other hand, the sum at amplifier 155 is output 3
9 to generate the following signal.

■3. − to 2 (CO3ψ−βsinψ)・・・・・・
...(9) That is, equations (8) and (9) are as follows: 2S1nψ' for V38-...
・・・・・・・・・・・・ αO)■2C to 39-
O3ψ′・・・・・・・・・・・・・・・・・・・・・
...... 1 circle If these DC signals are in terms of ψ' that include the desired indexing angle error compensation as expressed in equation (5), they are applied to the dual channel modulator 45. It will be converted at

Therefore, dual channel modulator 45 has its output lead 5
0 and 51 are supplied with signals having the following amplitudes.

V46=to3Sinψ' ・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・ (12) V4□
= K3CO3ψ′ ・・・・・・・・・・・・・・・
(13) These signals are input to the two-cycle error compensation circuit 48.

The two-cycle error in magnetic field sensors, including the flux valve type sensors disclosed herein, is due to flux
Caused by the presence of soft iron near the valve.

The presence of such soft iron would destroy the surrounding earth's magnetic field at that location.

This resulting error in the middle is called a sine wave error or two cycle error and has two complete cycles within a 360 degree azimuth rotation of the aircraft.

Generally, the average position of the soft iron introduced into the flux valve determines the direction of its effective vector.

For convenience, two-cycle error compensation is achieved by effectively dividing the entire vector into orthogonal components.

One of these orthogonal components is named the cardinal two-cycle error component, and the other is named the inter-cardinal two-cycle error component.

The basic heading two-cycle error component has the same values as the nose heading angle O0°, 90°, 180°, and 270°.

On the other hand, the two-cycle error component between cardinal directions is 45°.

It has extreme values at azimuth values of 135°, 225°, and 315° (i.e., the above-mentioned nose direction angle values).

In FIG. 4, this latter error is easily corrected by placing a variable resistor 199 in the feedback path of AC amplifier 200 excited by the signal on lead 46.

By adjusting controller 48b in accordance with the data taken during the swing of the installed compass, the output at lead 50 is corrected in an appropriate manner.

The two-cycle heading error between cardinal headings is the sine and cosine error that provides outputs to the nodes 50 and 51, respectively.
It is compensated by changing the gain balance between channels.

Correction of cardinal heading errors is accomplished by a single adjustment means 48.
This is achieved by the circuit of FIG. 4, which uses simple common circuit stages in a manner that minimizes a and sources of error.

The amount of adjustment of controller 48a is also determined by the ground swing method of the installed compass.

The 2S1nψ' signal on lead 46 is applied by lead 175 through resistors 177 and 180 to respective inputs of differential operational amplifier 188.

The 2CO8ψ' signal on lead 47 is summed with the 2S1nψ' term at input 183 via resistor 178, and similarly
The 2Sinψ' term signal from the node 46 is also added to the 2CO8ψ' term signal via the resistor 180 at the terminal 185.

Amplifier 188 has a resistor 187 connected between its output 189 and input terminal 183 (inverting side).

A variable resistor 186 with an adjustable controller 48a is connected to terminal 185.
and ground.

Variable resistor 186 constitutes a single cardinal heading error adjustment means, the adjustment of which affects the effective gain γ of amplifier 188.

Depending on the setting of controller 48a, output 18 of amplifier 188
9, the following compensation voltage is generated.

V] 89-7'(Sinψ' ten cosψ') -----
(14) This signal value v189 is connected to a branch lead at circuit point 190 for supplying this signal to the input of each amplifier 200 and 201 via resistors 192 and 193. given to.

As mentioned above, amplifier 200 has a variable resistor 199 connected between its output 203 and its human lead 195.

The other input of amplifier 200 is connected to ground through resistor 196.

Another amplifier 201 is connected to lead 47 via resistor 194.
2CO8ψ' is supplied to the signal from.

This amplifier 201 also connects its output 204 to its input lead 1.
It has a resistor connected to 97.

Amplifier 201 also has a resistor 198 between the second input (non-inverted uID) and ground.

Amplifiers 200 and 201 are provided with respective leads 195 and 197 to function as adder and inverter circuits, so that the signal of 231nψ' terms on lead 46 is converted into a correction term produced at terminal 190. and a summation signal is produced on output lead 50.

In the same manner, the 2CO8ψ' term signal applied to lead 47 is summed by amplifier 201 with the compensation signal at node 190 to produce an inverted signal on output lead 51.

In this manner, the voltages V50 and ■5□ (respectively lead 50
and the signal obtained at step 51) are as follows.

V2O=to4(sinψ'10γ(S1nψ' +CO3
ψsu) - (15)■5, -4[C]O3ψ'-1-7
(sin ψ'-H: os ψ) - (16) In equations (15) and (L6), the value 4 may include the influence of the adjustment of the resistor 199.

Therefore, the voltage at the output lead 50 is expressed as sin ψ'', and the voltage at the output lead 51 is expressed as COS ψ〃.

In this case, ψ“ is the basic heading and the basic heading towards the nose. represents the imposed ψ′.

From equations (15) and (I6), it is clear that the value of the corrected angle ψ" is expressed by the following equation.

where ψ" is the final output heading value component corrected for cardinal heading and two-cycle error between cardinal headings.

Amplifier 188 (whose effective gain γ is controlled by the setting of variable potentiometer 186) has a function γ(sinψ
'+CO3ψ') and that this function is added in the sine and cosine channels by the respective actions of amplifiers 200 and 201 and their associated circuits. Ru.

It is clear that correction of the cardinal heading two cycle error is made by a single adjustment.

Furthermore, a single stage associated with amplifier 188 is effective in reducing potential error sources and simplifying this adjustment means.

The invention can be used in other forms and the compass device of FIGS. 1A and 1B can be modified as shown in FIG.

In the embodiment of FIGS. 1A and 1B, the index angle error compensation and two-cycle error compensation signals are generated from the sine and cosine outputs of current servo loop 31 and are later generated from the sine and cosine outputs of current servo loop 31.
is used to be summed with the original or uncompensated value in one channel.

In Embodiment 1 of FIG. 5, the sine and cosine outputs of current servo loop 31 are used in essentially the same manner to generate compensation signals as DC signals, or, however, these signals and the original The summation with the signal is performed directly in the flux valve 11 by direct current feeding of the compensation signal to the flux valve windings, and is generally performed in accordance with the teachings of the above-mentioned U.S. Pat. No. 2,852,859. Effectively compensates for output.

In FIG. 5, like reference numerals are used to represent elements corresponding to those in FIGS. 1A-4.

The embodiment of FIG. 5 is similar to the embodiment of FIGS. 1A and 1B;
In series configuration, reference signal generator 2, flux valve 11, blocking capacitors 16, 17 and 18
, input insulator short T transformer 21, current servo loop 21,
An automatic gain control circuit 34, a dual channel modulator 45, power amplifiers or buffer amplifiers 52 and 53, an output short tee transformer 60 and a device 64 are used.

In generally the same manner as shown in FIG. 1A in connection with current servo loop output leads 10 and 10a, respective correction currents are provided to summing points 320 and 321 in FIG. flow to ground via the respective winding of the valve 11, in which case it is prevented from flowing to the Scott T-transformer 21 by capacitors 16, 17 and 18;

In the lower part of FIG. 5, the index angle error compensation circuit is shown.

The similarity to the corresponding structure in FIG. 3 is obvious and the simplification of the illustration will therefore be easily understood.

The sin ψ and cos ψ DC signal outputs (respectively produced at output nodes 32 and 33 of current servo loop 31) are inverted by impark 340 and then connected to switches corresponding to transistor switches 107 and 109 of FIG. The signal is alternately applied via means 300 to the input of variable gain amplifier 120a.

The gain of amplifier 120a is determined by an adjustment knob 37a generally corresponding to gain control of amplifier 120 by controller 37a and potentiometer 1131 according to the value β in FIG.
(as shown).

The output of amplifier 120a is connected to switch 301, which generally corresponds to transistor switches 122 and 123 of FIG.
Thus, the two branch leads as in FIG. 3 are alternately switched in the same way.

The control of switches 300 and 301 in FIG. 5 is the same as that in FIG. 3, but in FIG.
switch control means 302 controlled by the signal source 2 of
For convenience, it is abbreviated as follows.

In FIG. 3, the outputs of switches 122 and 123 are divided into two branches including amplifiers 128 and 129 for summation with the original sinψ and COSψ DC signals of current servo loop 31 by amplifiers 145 and 146. given to the circuit.

On the other hand, the two output branches of switch 301 in FIG. 15 at an appropriate ratio.

In FIG. 5, the ratio of DC currents flowing in the branches is determined by selecting resistors 303, 304 and 305 to have the ratio shown.

Current from resistors 303 and 304 is provided to flux valve winding 15, while current from resistor 305 is provided to flux valve winding 13.

If desired, resistor-capacitor circuits 306 and 307 may be used to reduce the transient effects of switches 300 and 301.

Thus, as in FIG. 3, the device of FIG. 5 performs time division of a single amplifier 120a between sin ψ and COS ψ by alternating operation of switches 300 and 301 to eliminate index angle errors. It serves to provide a compensation in which case the gain of amplifier 120a can be controlled by a single control element 37a according to the magnitude of this error.

Due to this operation, the winding 13 of the flux valve
, 14 and 15 and contributes to the 00Sψ output channel of the current servo loop 31 (the amount of sinψ current that contributes to It will be the same as the amount of COS ψ current contributing to the channel.

A variation of the cardinal and intercardinal two-cycle error compensator of the device of FIG. 4 is shown in FIG.

An important feature of this embodiment is that the compensation signal is summed with the main signal, ie at the flux valve 11 rather than at the output of the current servo loop 31.

In FIG. 5, the current servo loop 31 in leads 32 and 33 is used to compensate for cardinal two-cycle errors.
The sin .phi. and COS .psi.

The adder circuit 310 in FIG. 5 is the adder circuit 177-1 in FIG.
80, while the gain of amplifier 188a is controlled by knob 48 according to the magnitude γ corresponding to the tangent value of the desired correction.
Adjustment of a (shown as being changed accordingly).

As in FIG. 4, the output of amplifier 188a is provided to summing circuits 320 and 321 via branch leads and respective resistors 311 and 312.

The compensation current is added to winding 1 of flux valve 11 instead of being summed back to the uncompensated sinψ and COSψ channels in amplifiers 200 and 201 (FIG. 4).
3 and 15 are used with the values shown in FIG.

These compensation currents are effectively summed with the flux valve winding outputs that contribute to the sin ψ and COS ψ signal outputs of the current servo loop 31.

The 2-cycle error compensation signal between basic directions is also fluxed.
given to the valve winding.

The DC sin ψ signal output of the current servo loop 31 is given to a variable gain amplifier 199a.

The gain of amplifier 199a is varied by knob 48b depending on the amount of correction required.

Amplifier 199a in FIG. 5 corresponds to variable impedance 199 and amplifier 200 in FIG.

The output of amplifier 199a is modulated by resistors 313 and 314 according to the proportions shown in FIG. , 2 cycle error correction signal between basic directions is current servo loop 3
1 is added to the flux valve winding output contributing to the output of 1.

As described above, in the embodiment of FIG. 5, the index angle error and two-cycle error compensation signals are generated from the sin ψ and cos ψ DC outputs of the current servo loop 31.
The flux valve output signal provided to the current servo loop 31 is fed back to the appropriate flux valve windings at the desired ratio to be compensated.

Note that in the embodiment of FIG. 5, these feedback compensation signals are generated from the current servo output before the latitude compensated automatic gain control stage 34.

This is preferred because the compensation DC signals provided to the flux valve windings are related to the direction of the magnetic field sensed by these windings and not to latitudinal gain compensation.

[Brief explanation of the drawing]

FIGS. 1A and 1B are block circuit diagrams showing an embodiment of the present invention. FIG. 2 is a diagram showing details of the novel gain control circuit in FIG. 1. FIG. 3 is a diagram showing details of the novel index angle error compensation circuit shown in FIG. FIG. 4 is a diagram showing details of the novel two-cycle error compensator shown in FIG. FIG. 5 is a block diagram showing another embodiment of the present invention. In the figure, 177, 178, 179 and 180 are resistors, 188 is a differential operational amplifier, 186 is a variable resistor, and 191
, 192, 193 and 194 are resistors, 200 and 201
indicates an operational amplifier.

Claims (1)

[Claims] 1 In response to the geomagnetic field, the direction ψ of the geomagnetic field is expressed for a maneuverable navigational vehicle, and is proportional to tSinψ and cosψ, but is substantially unrelated to the magnitude of the geomagnetic field. In a magnetic compass data transmission system for a maneuverable vehicle having generator means for generating first and second alternating current output signals, the system comprises: (a) first and second input means; (b) first summing means for applying said signal proportional to sinψ and COSψ from said generator means to said first input means; second summing means for applying said signal proportional to sin ψ and cos ψ to said second input means, said amplifier circuit being connected in parallel relationship with said second input signal stage; controlling the magnitude of the effective gain γ of the means in proportion to the two-cycle data transmission error, thereby cooperating with the first and second summing means at the output of the amplifier circuit means; variable impedance means for providing a signal representing γ (sin ψ + COS ψ);
(E) Depending on the output of the amplifier circuit means, sinψ+
γ (sin ψ + cos ψ) and cos ψ + γ (S1 n ψ
a magnetic compass comprising means for correcting said two-cycle data transmission error, said third and fourth summing means for producing corrected data transmission signals each proportional to・Data transmission method. 2. In a magnetic compass device for a maneuverable vehicle that includes a flux bulb type magnetic compass for sensing the direction of the geomagnetic field with respect to the maneuverable vehicle, (a) an angle on the said vehicle; magnetic field detector means comprising a plurality of inductive elements arranged in parallel with each other and for providing said plurality of alternating current signals to said navigation vehicle proportional to the direction and magnitude of the horizontal component of the earth's magnetic field; ) signal processing for providing first and second DC signals connected to the inductive element and responsive to the AC signal and whose direction and magnitude are proportional to a predetermined function of the direction of the earth's magnetic field; (c) an amplifying means having an input means and an output means; (c) means for changing the first and second DC signals according to the output of the amplifying means; (e) an input means; means for supplying the first and second DC signals to give the output means a signal corresponding to the sum of the first and second DC signals; adjustable means for controlling the first and second DC signals in accordance with a function of cycle error; - means for correcting for errors; A magnetic compass device comprising means for correcting a two-cycle error in the magnetic compass device. 3. The generator means (a) includes a plurality of inductive windings responsive to the direction and magnitude of the earth's magnetic field, a first winding representing the direction and magnitude of the earth's magnetic field;
magnetic field detector means for providing a plurality of alternating current signals; (b) first and second unidirectional signals representing the direction and magnitude of the earth's magnetic field in response to the first plurality of alternating signals; (c) current servo means for generating, in response to the first and second unidirectional signals, first and second unidirectional signals representing the direction ψ of the earth's magnetic field and substantially independent of the magnitude thereof; (a) first and second variable conduction circuits having respective first and second conduction states; and (b) said first and second separate unidirectional signals; variable conduction circuit,
(C) switching means for simultaneously changing from one of said first and second conduction states to the other with a controlled duty cycle; and (C) present in each of said first and second variable conduction circuits; filter means for smoothing the respective currents flowing through the variable conduction circuits to form the third and fourth unidirectional signals; and gain control means comprising; (e) circuit means for separately modulating said third and fourth unidirectional signals with a reference AC signal to form first and second AC output signals representing said first and second AC output signals; comparing means for controlling said gain control means in response to an alternating current output signal, and (a) comparing a signal representative of said first and second alternating current output signals with a reference unidirectional signal; and (b) variable pulse width generator means for providing a controlled pulse width modulated signal to control the controlled duty cycle in response to the comparison means; 2. A magnetic compass data transmission system as claimed in claim 1, further comprising means for additionally responding to the alternating current output signals of 2. 4. A first and a second, which are proportional to sin ψ and cos ψ and substantially represent the direction ψ of the earth's magnetic field for the maneuverable vehicle, but are substantially independent of the magnitude of said earth's magnetic field.
In a magnetic compass data transmission system for a vehicle comprising means responsive to the earth's magnetic field having generator means for generating a unidirectional output signal of: (a) proportional to 31nψ and cosψ; cyclically and alternately responsive to said signal and having an adjustable effective gain β representing the indexing angle error of said vehicle and with a single output βcosψ
and (b) first time-sharing amplifier circuit means for generating periodic alternating output signals proportional to β cos ψ and β sin ψ; (c) second amplifier circuit means for passing the respective outputs of the first and second modes; a) first summing means for forming a signal representative of said first unidirectional output signal proportional to the difference in said vehicle index angle error and corrected for the effects of index angle errors of said vehicle; a second unidirectional output signal proportional to the sum of said signals proportional to cos ψ and β sin ψ and representative of said second unidirectional output signal corrected for the effects of said index angle error; 2. A magnetic compass data transmission system comprising means for correcting the indexing angle error, comprising: (e) utilization means corresponding to the first and second addition means; 5. To sense the direction of the geomagnetic field with respect to a maneuverable navigation vehicle (in a magnetic compass device for the navigation vehicle, which includes a flux bulb-type magnetic field detector mounted on the navigation vehicle, Means for compensating for errors in the road surface of the detector with respect to the directional axis of the body comprises: (a) a plurality of inductive elements disposed at an angle to the vehicle; magnetic field detector means for sensing the direction and magnitude of the horizontal component of the earth's magnetic field and generating said plurality of alternating current signals proportional thereto; (0) connected to said inductive element and responsive to said alternating current signals; , signal processing means for generating first and second DC signals whose direction and magnitude are proportional to a predetermined function of the direction of the earth's magnetic field; a) means for changing the first and second DC signals according to the output of the amplifying means; and (e) respective inputs of the amplifying means. and an input switching means connected to receive said first and second signals and an output switching means connected to supply the output of said amplifying means to said means. input and output switching means comprising: (i) an output thereof providing said second signal to said means when said amplifying means is switched to receive said first signal; alternately and simultaneously controlling said switching means such that said switching means are switched and vice versa, whereby said amount of said first signal that changes said second signal changes said second signal; (g) means for causing the amount of signal to be equal to the amount of the signal;
A magnetic compass device comprising: means for controlling said amplifier gain control means in accordance with said orientation error. 6. The generator means (a) includes a plurality of inductive windings responsive to the direction and magnitude of the earth's magnetic field, a first winding representing the direction and magnitude of the earth's magnetic field;
magnetic field detector means for generating a plurality of alternating current signals; (b) first and second unidirectional signals representing the direction and magnitude of the earth's magnetic field in response to the first plurality of alternating signals; (c) current servo means for generating the first and second unidirectional signals representing the direction ψ of the earth's magnetic field and having substantially no relation to the magnitude thereof; first and second variable conduction circuits generating a second unidirectional signal and having (a) respective first and second conduction states; and (b) separate first and second conduction circuits; The variable conduction circuit of
←) switching means for simultaneously changing from one of said first and second conduction states to the other with a controlled duty cycle; filter means for smoothing the respective currents flowing through the variable conduction circuits to form third and fourth unidirectional signals; and gain control means comprising: (b) representing the direction ψ of the earth's magnetic field; circuit means for separately modulating said third and fourth unidirectional signals with a reference alternating current signal to form first and second alternating current output signals; (e) said first and second alternating current output signals; a comparison means for controlling said gain control means in response to an output signal, and (a) for comparing a signal representative of said first and second AC output signals with a reference unidirectional signal; and (b) said variable pulse width generator means for providing a controlled pulse width modulated signal to control said controlled duty cycle in response to said comparison means; 5. A magnetic compass data transmission system according to claim 4, further comprising means for additionally responding to an AC output signal.