CA1065599A - Flux valve heading repeater compensation systems - Google Patents
Flux valve heading repeater compensation systemsInfo
- Publication number
- CA1065599A CA1065599A CA311,387A CA311387A CA1065599A CA 1065599 A CA1065599 A CA 1065599A CA 311387 A CA311387 A CA 311387A CA 1065599 A CA1065599 A CA 1065599A
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Abstract
ABSTRACT OF THE DISCLOSURE
A flux valve compass heading repeater system is provided with a compensating system which, when connected to a three-legged flux valve, provides fully compensated, three-wire output signals of the synchro data transmitter type for direct use in apparatus requiring precision three-wire heading data. The compensating system includes control circuits for generating sine and cosine components of magnetic heading and for compensat-ing them for typical compass errors such as those induced by changes in operating latitude and two cycle and index errors.
Latitude compensation is accomplished by a novel proportional automatic gain control; two cycle cardinal heading error compen-sation is accomplished by a compensation circuit having only a single manual control, while index error compensation is similarly accomplished by a compensation circuit requiring only a single manual control.
A flux valve compass heading repeater system is provided with a compensating system which, when connected to a three-legged flux valve, provides fully compensated, three-wire output signals of the synchro data transmitter type for direct use in apparatus requiring precision three-wire heading data. The compensating system includes control circuits for generating sine and cosine components of magnetic heading and for compensat-ing them for typical compass errors such as those induced by changes in operating latitude and two cycle and index errors.
Latitude compensation is accomplished by a novel proportional automatic gain control; two cycle cardinal heading error compen-sation is accomplished by a compensation circuit having only a single manual control, while index error compensation is similarly accomplished by a compensation circuit requiring only a single manual control.
Description
655i~3t~
BACKG~OUND OF THE I~VENTIO~
1. ~L~ ~
The invent.ion pertains to mean~ for the compen~ation of undesirable errors or changes in th~ signa:L characteri~tics o flux valve data repeater systems and more particularly relates to apparatus for the correction of variations in the outputs of flux valve compass data repeater sys~ems, :including errors due to variation in th~ horizontal co~ponent of the earth's magnetio ield, index angle errors~ and cardinal and inter-cardinal heading errors.
BACKG~OUND OF THE I~VENTIO~
1. ~L~ ~
The invent.ion pertains to mean~ for the compen~ation of undesirable errors or changes in th~ signa:L characteri~tics o flux valve data repeater systems and more particularly relates to apparatus for the correction of variations in the outputs of flux valve compass data repeater sys~ems, :including errors due to variation in th~ horizontal co~ponent of the earth's magnetio ield, index angle errors~ and cardinal and inter-cardinal heading errors.
2~ Descri~tion of the Prior Art When navigating at high latitudes with flux valve magnetic compass systems, difficulty is experienced because o:E the decreas- -ing strength o the horizontal component of the earth's magnetic ~ield, especiall~ at high latitudes~ A flux valve ~agnetic com-pass is normally arranged to ~ense only the horiæontal component . ~ .
~ of the earth's field. As a con~equence at high latitudes, the . ....................................................................... .
;~ strength of the sen~ed component i5 proportionally le~sened, and -~ the compass system experiences decreasing sensitivity, resulting :~t~ 0 in headin~ informati~n of diminished accuracy.
Prior art sy~tems have sought to solve this compensation ~ problem of providing an input of the magnetic compass data ;i repeater 3ubstantiaIly independent o variation~ in the strength : :
, o~ the horizontal component of the earth's field, as by control--! ling the gains o~ ampli~iers or the effective values o impedances t ~1 in the separate cnannel~ of the data transmitter system in a J 1' :
relatively complex manner, but generally in inverse relation to the si~nal strength as measured at the flux valve it~elf~ Such priox art arxangements are described in D. A~ E~pen in the United 30 States patent 3,548~284 for "Synchro Data Tran~mission Appara~us aving D~crete Gain Changing to Compensate for Undesi:rable 6~
1 Signal Gradient Variations", issued December 15, 1970, and in J, R. Erspamer and G~ W. Snyder in the United States patent
~ of the earth's field. As a con~equence at high latitudes, the . ....................................................................... .
;~ strength of the sen~ed component i5 proportionally le~sened, and -~ the compass system experiences decreasing sensitivity, resulting :~t~ 0 in headin~ informati~n of diminished accuracy.
Prior art sy~tems have sought to solve this compensation ~ problem of providing an input of the magnetic compass data ;i repeater 3ubstantiaIly independent o variation~ in the strength : :
, o~ the horizontal component of the earth's field, as by control--! ling the gains o~ ampli~iers or the effective values o impedances t ~1 in the separate cnannel~ of the data transmitter system in a J 1' :
relatively complex manner, but generally in inverse relation to the si~nal strength as measured at the flux valve it~elf~ Such priox art arxangements are described in D. A~ E~pen in the United 30 States patent 3,548~284 for "Synchro Data Tran~mission Appara~us aving D~crete Gain Changing to Compensate for Undesi:rable 6~
1 Signal Gradient Variations", issued December 15, 1970, and in J, R. Erspamer and G~ W. Snyder in the United States patent
3,646,537 for an "Automatic Gain Control for an Electromechanical Transducer", issued February 29, 1972, both patents being assigned to the Sperry Rand Corporation. While these concepts ~j:
have been useful in providing adequate magnet:ic field comperlsation in most circumstances, the compensating signa:Ls ~ctually compen-sat~ only for variation in the horizontal magnetic field componen~
and generally do not addition~lly correct ~ully for gain changes caused by co~ponent variations or due to temperature or power supply voltage drifts or to component aging. Furthermore~ the characteristics of the individual gain control elements of the individual channels of the data system may vary without proper corrective relative adjustments whereby two-cycle transmission errors are induced within automatic ~ain control stages.
The improved system di~closed by J. R. Erspamer and G. W. .
Snyder in the United States pate~t 3,784,753, issu~d January 8, ~-1974 ~or a "Multiplexed Gain Control for a Synchro Data Trans-mis~ion System" sought more ully to overcome these prior æ t deects by a relati~ely complex and expensive correction circuit.
Though it generally overcame such defectsJ it was found tha~ some undesirable two cycle error could be generated in its relatively complex automatic gain control stage, and that a simple way was needed ~or identically changing the gains of both of the sine ~ i a~d oslne channels of the data transmission system, but retain~
ing th~ advantages o~ the concept of patent 3,784,753 Prior art ~ystens have additionally sou~ht ~o provide correc ion ~:or the cardinal heading error in compa~s data ::
transmission systems by u~e of networks including precision - :~
. :, -3~ ~ di~erential synchros or ganged dual potentiometers which ~ust :
traa~ each other with high preci~ion i they are not themselves i to in~roduce errors. According to the present invention, the expense of obtaining such selected synchros or precision potentiometers is desirably eliminated. Index angle error was similarly corrected in prior compass data ~ransmission systems by using precision synchros or ganged dual potentiometers of similar quality. It is found increasingly desirable to substitute simple and less expensive networks permitting single adjustment control for each of these correction purposes and, at the same time, retaining a high degree of precision.
The present invention provides means for correction of undesirable changes in the signal amplitudes in multiple channel Elux va1ve data repeater systems partly by the employment of a simple common automatic gain control in a circuit configuration which not only compensates for earth's magnetic field strength changes, but also corrects for the effects of other error sources without introducing the errors of prior art systemsO The novel control circuit of the present invention monitors the data repeater control signals near the inputs to the utilization device3 rather than merely at the outputs of the flux valve. By monitoring the inputs at the utilization device, and by using the data repeaker excitation voltage as a switching reference, the gain control, being part of a closed feed back loop, compensates not only for 2Q changes in the operating latitude bu~ also for gain changes caused by varia-tions of component parameters and by other effects without itsel introducing naw errorsn According to a broad aspect of the present invention, there is provlded in a magnetic compass system for navigable craft including a magnetic field detector of the flux valve type mounted in the craft for sensing the ; direction o the earth~s magnetic field relative to the crat~ the combination for correcting for any errors in the orientation of said detector relative to the di~ection axis o~ said aircraft comprising, magnetic field detec~or means ~ including a plurality oE inductor elements angularly disposed on said cxaft for sensing the direction and magnitude of the correspondingly disposed horizontal components of the earth's magnetic field relative to the craft and -1~655~
for provid.ing a corresponding plurality of alternating signals proportional thereto, signal processing means coupled with said inductor elements and responsive to said plurality of alternating signals for pro~i.ding first and second direct current signals proportional in sense and magnitude to predeter-mined functions of said earth's magnetic field directionl amplifier means having an input an~an output and means for effectively controlling the gain thereof, means for modi~ying said first and second direct current signals in accordance with the output of said amplifier means, input and output switching means at the respective input and output of said amplifier means, said input switching means being connected to receive said first and second signals and said output switching means being connected to supply ~he output of said amplifier means to said modifying means, means al~ernately and simultaneously controlling said switching means such that when said amplifier mecms is ;
switched to receive said first signal its output is switched to supply the means for modifying said second signal and vice versa whereby to insure that :- . :. :
the amount of said first signal modifying the second signal is identical to the amount of said second signal modifying said first signal, and means for controlling said amplifier gain control means in accordance with said :
orientation errorO ~
The invention will now be described in greater detail with ~ :
reference to the accompanying drawings, in which:
Figures lA and lB illustrate, partly in block diagram form, the principal elements of the invention and their electrlcal interconnections; -- :
Figure 2 is a portion of Figures lA and lB showing details o~ the :-novel automatic gain control circuit;
Figure 3 is a detailed circuit diagram of the novel index error angle compensa$or of Figure lB;
.. . .
F~ure 4 is a detailed circuit diagram of the novel two cycle compensator also of Figure lB; and : Figure 5 lS a bloclc diagram of an embodiment of the invention alternative to that of Figures lA and lBo : ~ ~ 5 . :
:
s~
In Figures lA and lB, the novel compensated compass system in-cludes a magnetic azimuth detector or flux valve 11 wh.ich may be of the general type disclosed in the M. C. Depp United States patent 2,852,859 for a "Flux Valve Compensating System", issued September 23, 1953 and assigned to Sperry Rand Corporation. Other details of such flux valve devices are disclosed in the D. J. Kesselring United States patent 3,573,610, issued April 6, 1971, in the D. JO Kesselring et al United States patent 3,641,679, issued February 15, 1974, and in the United Sta~es patent application SO No 380,523 for "A Flux Valve Apparatus for Sensing Both Horizontal and Vertical Components of an Ambient Magnetic Field", filed July 18, 1973, issued as patent 3,873,914 March 25, 1975 and assigned to Sperry Rand CorporationO
Flux valve 11 is excited by alternating current source 2, which may be a : :
conventiQnal 400 Hz. oscillator or signal generator and which is coupled to : ..
exc.itation winding 12 of the flux valve 11. .
.
: ~ ' .
: : :
- 5a -::
: ' ' ~6~
As disclosed in the aforementioned Depp and ~Cesseling patents, f lux valve 11 ha~; three wye~collnected inductive windings 13, 14, and 15 on a corresponding wye-~hap~3dl core, the winding legs meeting at a common grounded terminal E'. The terminals of windings 13, 14, and 15 opposite terminal F are respectively labelled A, B, and C. Terminals A, B, and C may, i desired, be supplied with single cycle compensation slignals from a single cycle compensator (not shown) of the general type shown in the af orementioned U. S. patent ~, 85 2, 859 ..
10Terminal A of flux valve 11 is connected via a blocking ~:.
capacitor 16 to one winding 20 of a Scott tee transformex 21, while $erminals B and C are connected via re~pective blocking capacitors 17 and 18 to the respective ends of a second input ~: :
winding 22 of Scott tee transfoxmer 21~ Winding 22 has a center tap connected to the other end of winding 20.
As is well known, the signal outputs of windings 13, 14, ::
and 15 ha~e a requency double that applied to exci ation winding 12. The ~requency doubled cosine output o winding 23 of trans- :
former 21 and its frequency doubled sine output in winding 27 are conne~ted to current servo loop 31. Additionally supplied to current ~ervo loop 31 via lead 2~a is the output of ~requency doubler 29. Since frequency doubler 29 i~ excited ~y generator 2, its output on lead 29a will have an 800 Mz frequency and serveq as a re~çrence sigDal source for servo 31~
As describçd in detail in the D. H~ Baker~ F. H~ Kallio U~S. patent-3,67~,593 for a "Compass System and Components :Thexe~or Having Automatic ~ield Cancellation", issued July 25, ..
19~2 to Sperry Rand Corporation, current servo 31 supplies out puts on leads 32 and 33 which are direct current signals re~pec-~30 tlvely proportional ~n amplitude to the sine and cosine of craft 6. . ~ ~.
5~ 3 1 magnetic heading ~Hm sin yJand Hm cos y~). Accordingly~ the horizontal components of the earth's magnatic field sensed by the flux value windings 13, 14, and 15 are resolved into sine and cosine component.values that are then convert~d by current servo 31 into proportional direct currents on leadls 32 and 33. As taught in the aforementioned Baker et al patent, these direct current components are fed back via leads 10 and lOa into wind-ings 13 and 15 o~ flux valve 11, which currents tend to cancel the earth's magnetic field therein. The feed back arrangement and its many advantages are d~scussed in detail in th~ afore-mentioned patent 3,678,593,includlng closed loop operation afford-ing high accuracy outputs in the ~orm of analog direct current outputs proportional to the sine and cosine of craft magnetic headlng .
Accordingly, the 800 Hz., three-wire magnetic azimuth inormation derived by the horizontal magnetic field detector or flux valve 11 is converted to direct current signals proportional to the sine and cosine of craft heading by the cooperation of Scott tee transformer 21 and current servo 31. The magnitudes of ~he outputs on leads 32 and 33 are thus a function of craft magnetic azimuth or heading and the intensity of the horizontal co~ponent of the earth'~ magnetic field. The variation in the magnitude of the sine and cosine outputs caused by any change in magnetic ield strength Hm affects only the output gradient (volts per a~imuth degree~ and does not change the trigonometric rel~tionship of khe input magneti¢ heading angle ~ and the output vol~age~ of current servo 31, which may therefore be expressed as follows:
V32 ~ Xl sin and . . .
V 3 - K c~ ~ (2) ' .':
7.
;55~
1 where ~1 allows for the gain of ~urrent servo 31 and has dimensions of volts per oersted.
The signals V32 and V33 on leads 32 and 33 serve as two inputs to the automatic gain control circuit 34, which circui.t also receives certain fed back signal~ on leads 56 and 57. As will be further discus~ed, the fed back signals arise at the outputs o buf~er amplifiers 52 and 53 after the outputs of automatic gain control 34 are processed at least by dual channel modulator 45. To understand the operation o~ the gain control circuit 34, the presence of the index error compensator 37 and khe two-cycle compensator 48 may be ignored for the moment as a matter of convenience.
The final output of the compas~system supplied by leads 61, 62, and 63 to an aircraft navigation system or other utilization clevice 64 is usually re~uired to be useful in a three wire synchro data transmitter s~stem and to consist of proportional voltages between pairs of such leads, as betwee~
leads 61 and 62, 62 and 63, 63 and 61. These may nominally be 11.8 volts, for example, and mu~t be maintained at a constant gradient in the interes~ of me~ting required compass accuracy over a wide range of horizontal magnetic field strengths Hm.
Becau3e ~he output o~ flux valve 11, and there~ore the output of current servo 31, has a gradient which i~ directl~ proportional in magnitude to the horizontal magnetic field ~trength which, of course, varie~ with latitude~ the automa~ic gain controlstage 34 is required to hold the system output signals at leads 61~
62, and 63 at the de~ired nominal 1108 volt leg-to le~ constant ~radient.
For this purpose, the direct current outputs on leads 3S
; 30 and 36 of gain control 34 are suppliad to the con~entional dual channel modulator 45, each o~ the two individual channels of 8. - :
1 which are supplied by lead 2a with the 400 Elz. reference signal output of generator 2. The direct current signals on lead~ 35 and 36 are modulated by the 400 ~Iz. alternat:ing current signal in khe conventional manner so that 400 Hz. signals appear on leads 46 and 47, proportional respectively to the sine and cosine of the magnetic heading of the craft. After individually separate supply to buffer ampli~iers 52, 53, equally amplified versions of these signals appear on leads 54, 55 to which the feed back leads 56 and 57 are respectively connected.
The automatic gain control 34, shown in greater detail in Figure 2, monitors the gradient at the output lea~ls 5~, 55 of buffer amplifiers 52, 53, respectively, compares the result to a reference voltage level, then varies the system gain accordingly by control o~ the gain of automatic gain control circuit 34. If the gradient at the outputs of buffer amplifiers 52, 53 of Figure lA is less than a predetermined level, the voltage gain of circuit 34 i9 increased to bxing the output of the buffer amplifi~rs 52, 53 up to the proper level~ The output of the buffer amplifiers 52, 53 and the voltage ~radient is similarly controlled. The signal levels at output leads 35, 36 of the automatic qain control : 34 are ultimately passed through the outpu~ Scott tee transformer 60~ The outputs of transformer 60 are therefore fully independent of any earth~s magnetic field strength variation. ~ :
Thu8:
V3~ - K2 ~in ~ t3 . .
~ and.
:~ V36 = ~2 ~os ~ (4) : wher~ K2 is a new propor~ionality con~tant.
Autom ~ gain control circui1: 34 is desi~ned to prevent thc inkroduction of any stand-off or unbalance between trans~ -mls~1on channels, resulting in cyclic errors, into the c:raft g~
~55g~
1 heading output data. The individual gains of the sine and cosine channels are now identically controlled and there are no of~-set voltages induced into the direct current s.ignal~ representing sine and cosine of craft magnetic heading. As is seen in more detail in ~igure 2, the direct current .signals at leads 32 and 33 are, as before, provided by the cooperative action o~ ~lux valve 11, Scott tee transformer 21, and current servo 31, an~ are respec- :
tively proportional in amplitude to sin ~ and cos ~ . Output lead 32 is coupled in series through resistor 75, ~unction 76, . .
re~istor 77 and input lead 35 to one channel of the dual modulator 45, At input lead 35 is a capacitor 78 coupled tc ground and forming a low pass filter with resistor 77. Likewise, the second output lead 33 i5 coupled in series through resistor 79, junction 80, resistor 81, and input lead 36 to a second channel of dual modulator 45. At input lead 36 is a capacitox 82 forming a low pass filter with resistor 81. Switching or c~opper transistors B3 and 84 are respectively coupled to ground from junctions 76 and 80 and control current flow through their emitter and collector electrodes in accordance with their respective base voltages.
The direct current signals on leads 32 and 33 are chopped by transistors 83 and 84, respectively, and after ~moothing by low pass filters 77-78 and 81-82~ form direct currents tha-t axe ~.
individually modulated in dual channel modulator 45 ~y the 400 Hz. re~erence signal on lead 2a. These dual channel output vol~ages are directed by Scott tee transformer 60 as three-wire : ~ynchro data to a navigation system or oth~r utilization device 64.
For purposes of controlling the automatic gain contxol ~ ~
30 ~ircui~ 34, the same 400 Hz. modulated output currents on leads : :
54 and 55 axe respectivel~ coupled by leads 56 and 57 to a ~ ~
~.:
~ ~ :
.: . .
10.
.. ..
js~
1 constant amplitude, variable phase circuit comprising resistor 95 and capacitor 96 coupled in ~eries with l~eads 56, 57 at junction 97. Circuit 95 96 is of the general kind discussed in ~he D, A~ Espen UOS. patent 3,548,28~, entitled "Synchro Data Transmission Apparatus ~aving Discrete Gain Changing to Compen-sate for Undesixable Signal Gradien~ Variation", issued December 15, 1970 and in the D~ A~ Espen U.S. patent 3,617,863, entitled "Constant Amplitude Variable Phase Circuit", issued November 2, 1971, both patents being assigned to Sperry Rand. The constant amplitude, variable phase signal found at junction 97 is recti-fied by diode 94 and appears as a variable unipolar voltage at one input of a conven~ional integrating operational amplifier 92 having its output coupled by capacitor 91 to its same input. To the second input of amplifier 92 is coupled through resi~tor 93 a stable positive unidirectional reerence voltage from a suit-able ~ource (not shown~ connected to terminal 98. As ~hown in ~:
the drawing, amplifier 9~ and its a~sociated circuit act as -~
conventional comparator means or in ~ffect comparing the ou~put .
gradie~t on leads 54, 55 with the fixed level voltage at terminal ~8 yielding an integra~ed output as a function of the diffexence of the two voltage levels at leads 97 and 98.
The posi~ive signal at the output of device 92 i~ coupled through re~istor as to one input o~ ampliier 87, to the other input of which is supplied at terminal 90a and through re~i~tor 89 the 400 Hzo excitation signal from generator 2. Under control of i*s varying amplitude direct current and constant ampli~ude alternating input current~, circuit ~7 acts as a conventional variable pulse-width generator for supplying a 400 H~ vaxiable pulse-width signal at junction 74.
The variable pul~e~width signal is coupled in parallel from junction 74 through t~e respective resistors 85, 86 to the base 11, . ,: : : . . . - ... . . ..
~v~
1 electrodes of chopper transistors 83 and 8~ to control the rela tion of the conduction to non-conduction times of these switchiny transistors~ The transistors 83, 84 are synchronously corlducting at the same time and then are both non~conducting for a con~
trolled period of time depending upon the pulse width o~ the out-put of amplifier 87~ As the non-conducting part o the cycle is increased in time duration, the ~otal current per cycle passing from lead 32 to lead 35, for example, is increasedO In other words, proportionately less of the current a~ailable on lead 32 is dumped to ground. In this manner, the voltage between leads 54, SS is made independent of any ~mplitude variations in the total flux valve data as well as amplitude variation~ resultin~
from other distuxbing factors in the signal channels between current servo 31 and buffer ampli~iers 52 and 53~ Accordingly, the three-wire output supplied to utilization device 64 of Figure 1 by transformer 60 is maintained nominally constant from leg to ;:~ -leg) such as at 11.8 volts.
In the comple~e system as illustrated in Figure 1, the outputs V35 and V36 of the automatic gain control 34 on leads 35 and 35 may be first processed by the novel index error angle compensator 37 prior to 400 Hzo modulation. For this purpo~e, the compensation circuit of Figure 3 is employedD The index ~:
error angle comp~nsated by circuit 37 is pre~ent because of the noxmal difficulty o~ achieving perfect alignment between the air- ~ :
craft fore-aft axt~ and the effective electrical ore-a~t axis of the 1ux valve 11. Accordingly, index angle error compen- :~
sator 37 is provided to permit a manual correction to be made : after ~ystem installation by performing; in essence, the ~ame unction as might be provided by a relatively expensive servo differential which so~e prior art sy~stems have employe~O How-ever~ since installation accuracies are usually within ~10, the ::
:
.
12.
;~ ii5~
1 compensation ~unction may beaccurately performed by the relatively inexpensive circuit of Figure 3 wher~in only a single potenti~
ometer shaft need be adjusted. It will be apparent that the correction is made by the novel compensa~or herein di~closed to the value of angle ~ when it i9 still in the trigonometric form of sin ~ and cos ~ data.
Accordingly, the apparatus of Figure 3 accepts two inputs K2 sin ~ and K2 cos ~ and internally generates two values -K
cos ~ and -K2 ~ sin ~ . The K2 sin ~value and the -~2 ~ cos ~
value are added according to the well known trigonometric .identity to form K2 sin ( ~ ~ ~ ) where ~ + ~ may be used to repre-sent a corrected value of ~ . The K2 cos y value and the -K2 sin ~ values are similarly added to form K2 cos ( ~ + ~ ). In accord with the teachings of the present in~ention, the ~ terms must be identical in both the sine ~nd cosine output channels to effect preci~e compensation; the same source for the ~ term is used in the two channels of the circuit.
In greater detail, the circuit of Figure 3 has, in operation D
negative valued direct voltages respre~enting K2 sin ~ and K2 cos 20 ~ as re~pactive inputs on leads 35, 36, and these are respec~ ~ -tiv~ly supplied directly to inputs o conventional unity gain output amplifiers 145 and 155 at the right side o the figure~
The ~ame two direct neg~tive voltage~ are used in the remaining or major part of the circuit to produc~ compensating voltages also for insertion into amplifiers 145, 15S. For t~e latter purposes, the -sin ~ t~rm on lead 35 is coupled through a con-ventionaI inverting amplifier 103 to the switching tr~nsi~tor 107. Ampliier 103 has its output terminal 104 coupled through a re~i~tor 102 to its input terminal 101 and additionally ha~ a second input terminal coupled to ground through resistor 105.
The -cos ~ term vn lead 36 is coupled dixectly to ~witchin 13.
~3~5~
1 transistor 109. Transistors 107 and 109 are made alternately ~ully conducting and fully non-conducting so that, first, the output of amplifier 103 appears on lead 108a and then, the signal passed by switching transistor 109 appears on lead 108b.
Since both of the leads 10~a and 108b are coupled to the adjus-t-able contact 113a of potentiometer 113, it i.s seen tha-t the signals alternately passed hy switching transistors 107 and 109 are alternately applied to contact 113a for time-sharing purposes .
in the shared amplifier 1~0, The switching transistors 107 and 109 are made alternately conducting under control of a sine wave signal ap~earing on -lead 2b; thi~ signal is conveniently obtained from ~he 400 Hz.
generator 2 of Figure lA, though other regul~x stable-fxe~uency signals may alternatively be employed. In practice, the 400 Hz.
cycle signal on lead 2b is applied by lead 106 to control the con~uction of transistor 107. So that time sharing may be ;:
employed, the signal on lead 2b is coupled via lead ~11, the 180 phase shifter 112, and lead 110 to control the operation vf switching transistor 109. -:~
In this manner, the signals on leads 35 and 36 are .
alternately supplied at the selected contact point o potenti- :
ometer 113, the latter having its opposed terminals 113b and 113c coupled to inputs of operational amplifier 120~ The output terminal 121 of amplifier 120 is coupled to its input at termlnal 113b via resistor 115, and terminal 113c is connected through rssl~tor 114 to ground in conventional a~hionO The input of ampli~ier 120 is thus time 3hared and its output on terminal 121 iB ~pp1ied to a secc~nd pair of switching t~ansistors 122, 123, these transistors being arranged for rontr~lling the series 30 signal flow through the respective resistors 126, 127 to ampll~iers 128, 1~9~ The ef~ective ga.in of amplifier 120 is ~ .
. ' ' 14.
. .
~55i~
1 changed according to the setting of the single control 37a, which control is manually set in accordance with the known magnitud~ of the index error de~ermined as a result of conven-tional ground swinging operations~
Conductivity of switching transistor 122 occurs simul-taneously with the conductivity of switching transistor 109.
Xn like manner, conductivity of switching transistor 123 is made simultaneous with the periods of conductivity of switching transistor 107. This operation is accomplish~d by controlling the conductivity of switching transistor 1~3 according to the si~nal on lead 2b when supplied directly to switching transistor 123 via lead 125. The desired s~nchronous operation of switch ing transistox 122 is accomplished by providing the 180 phase :~
shifted signal from circuit 112 via lead 124 to transistor 122.
In this manner, both channels o the circuit time share the use of the common amplifier 120, ensuring that identical corrections .
are applied to the two channels; l.e" that the amount of the ~:
sine term added to the cosine term is identical to the amo~mt : .
o the coslne term subtracted in the sine channel. It i9 :Eurther observed that adjustment of the single control 37a allows adjust-ment of potentiometer 113 so that both channels are identically set in accord wlth the magnitude of the index error.
The time shared cuxrents alternately flowirlg through switching transistors 122,123 are alternately supplied to the conventional unity gain amplifiers 128/ 129, and the respec~ive outputs on the terminals 132, 133 flow ~hrou~h resistors 141, ~ .
~ 150 to the sam~ respectlve input terminals of ampli~iers 145, ~ -: 155, as a~e connected to the respective leads 35, 36. ThP out- : -puts of amplifiers 145 and 155 may be smoothed by the action of appropriate low:pass ilters so as to remove any 400 Hz, modula~
tion ~xom the outputs appearing in the xespective output leads 383 39. In the embodiment illustrated, the filters are placed '' '::
5g~
1 at the input~ of amplifiers 128 and 12~ and comprise resistors 126, 127 and capacitors 128a, 129a, respectively.
The Mathematical relation expressing the index error as a function o~ the sine and cosine of magnetic heading is:
~' = tan 1 sin ~ - ~ cos ~ (S
where: cos i ~ ~ sin ~
~'~ the compensated output, _ the uncorrected input, and ~ = the tangent of the index errox.
~hus, by adjusting the gain of the time ~hared amplifier 120 in acc~rdance with the value of ~, expxe~sion (5) is satisfied as follows. The output of amplifier 128 is:
V132 - ~K ~ cos ~ (6 when transistors 109 and 122 are conductin~, and the output of amplifier 129 is~
V133 = -K2 ~ ~in ~ (7) when tran~istors 107 and 123 are conducting. The addition at amplifier 145 produces on output:
38 K2 ~sin ~ - ~ cos ~ ) (8) while the addition at ampli*i~r 155 produces an output:
V39 = K2 (cos ~ in ~ ) or: 38 K~ sin y (10) and:
V3~ ~ ~2 cos ~
These direc~ current signals are ready ~or conversion iTI dual channel modulator 45, providing as they do direct current ~ignals in terms of ~ ' containing the desired index angle error compen, sation~as set ~orth in equation (S)~ Thu~, the dual chan~el ~ modulator ~5 supplies on it~ output leads 50, 51 400 Hz. ~ignals ~ -whose amplltude~ are:
.'~
.'" ' , ' " ~ .
:' 16.
,, . ~ , ., . , ,. . :
:
1V46 = K3 sin ~ (12) 47 3 c s ~ ~13) and these signals serve as inputs to the two cycle error compensator 48 Two cycle error in a magnetic field sensor, including the flux valve type of sensor disclosed herain, i~ induced by the presence o~ a so~t iron mass or masses in the vicinity of the flux valve which tend~ to distort the earth's ambient magnetic field thereat. As the name implies, the error is a sinusoidal error and has two complete cycles within 360 of azimuth :rota-tion of the craft. In general, the average location of the oft iron mass relative to the flux valve determines the direction of its effective vector. For convenience, the two cycle co~pensa-tion is accomplished by effectively breaking down the total .:
vector into orthogonal components, one termed the cardinal two cycle error component and the other the intercardinal error : component~ The cardinal two cycle error has extremllm values at heading angle values 0~ 90~, 180t and 270. Intercardinal two cycle error, on the other hand/ has extremum values at 45, 1~5, : ~20: ~ 225, and 315 azimu~h values. Xn the invention o~ ~igure 4, the ; ~ latter error is readily corrected ky placing an adjustable series resistor l9g in the ~eed baclc path of arC. amplifi~r 200 that is excited by lead 461 Adjustment of control 48b in accord with data ;liakeal during installation compa~3s swings then corrects , .
the output at lead S~ in the appropriate mannerr The inter- .~
card1nal two cycle headin~ error .is co~pensated by chan~ing the .~.
ga~n balance hstween the sine and cosine channels supplying outp~ts on the respective leads 50, 51~ : :
Correction of the cardin~l heading error lS accomplished ~y ~ ~ . . ..
~ 30~ the ~:1mpl~ circuit o~Figure 4, using a sin~le adjustment 48a ::
,. ..
17- :
1 and a common circuit stage in a manner mlnimizing error sources and characterized by simplicity. The amount o adjustment of control 48a i~ also determined by the installation ground 3wing-ing process. The X2 sin ~ signal on lead 46 is supplied via lead 175 through resistors 177 and 180 to the re~pective inputs of diferential operational amplifier 188. Thevalue K2 co~
from lead 47 is added through resistor 178 al; terminal 183 to the K2 sin ~ ' term ~imilarly, the term K2 sin ~ ' from lead 46 is added through resistor 180 at terminal 185 to the K2 cos 10 term. Amplifier 188 has a resistor 187 coupled between its output 189 and the input terminal 183 in conventional f a~hion .
A variable resis~or 186 with an adjustable control 48a is coupled : -between terminal 185 and ground. The variable resistor 186 con- :
stitutes the single cardinal heading error adju~tment, its varia-tion affecting the effective gainYof amplifier 188~ According to the setting of control 48a, a compensating voltage appear~
at the output 189 o~ amplifier 188:
: V18g = - ~lsin ~ ' + cos ~') t14) The single value V189 is coupled at junction lgO ~o branching : 20 leads for ~upplying this signal throu~h r~sis~ors 192, 193 t.o the input of respective amplifiers 200 and 201~ A~ n~ted previously, amplifier 200 has a varia~le resistor 199 coupled between its output 203 and it~ inpu~ lead l~S. The other input to ampli~ier 200 i~ connected ~hrough resistor 196 to ground. A
furth~x amplifier 201 is supplied with the signal K2 CQS ~ ~ from :
the lead 47 through resistor 194 and is similarly provided wi~h a re~istor 202 connecting its output 204 to its input lead 197. :
: : It similarly employ~ a r~si~tor 198 coupled between a second input and ground. Amplifiers 200 and 201, through th2 respective : :
connectors l9S and 197, serve a~ adding and inverting circuit~
.
18.
- . ~ . . . .
1 so that the K2 sin ~' term on lead 46 has added to it the correction term appearing at terminal 190 and the summation is found on output lead 50. In a similar manner, the K2 cos ~' signal supplied on lead 47 is added to the compensating signal on junction 190 by amplifier 201 and its associated circuit, an inverted ~ignal being generated on output lead Sl. In this manner, the voltage VsQ is:
V50 , K4 Esin ~ (sin l~l + cos ~ '~ (lS) and that on output lead 51 is:
V51 = K~ rcos ~ -t ~(sin ~' ~ cos ~'~ (16) In equations ( 15 ~ and (16), the oew value K4 may include the effect of the adjustment of resistor 199~ Thus, the volta~e on .:
output lead SO is sin ~ ' and the voltage on output lead 51 i~
cos ~'', where ~ represents ~ corrected both for cardinal and intercardinal heading errors. From equations tl5) and ~16), it is ev.ident that the value of the corrected angle ~'~ i5 expressed by the following equation:
= tan_ ~1 + ~ cos~ ' + ~sin ~
~ '~ being the final output headlng ~alue components corrected 20 for cardinal and intercardinal ~o cycle errors. It is seen ~ :
that amplifier 188, the effective gain ~ of which is controlled ~:
by the setting o~ the variable potentiometer 186 t cooperates in .
the circuit in generating the unction ~ ~sin ~ ' + cos~
and this function is added in the sine an~ cosine channels by : the respective action of amplifiers 200 and 201 and their associated circuitsO It is observed that correction of the ...
cardinal heading two cycle error is accomplished by manual opera-ion of a single adjustment. Fuxthermore, the single stage . .
a~ociated with amplifier 188 mini.mizes potential error sources :;
. .
and aids in simpli~ying the adjustment procedure.
.
:., :.
19~
1 It will be understood that the invention may be employed in alternative oxms and that the compass system of Figures lA
and lB may be modified within the scope of the claims appended hereto, for example, as illustrated in Figure 5. In the embod-iment o Figures lA and 1~, the index compe:nsation and two cycle error compensation signals are generat~ed from the sine and oosine outputs of the current servo 31 and are re-applied downstream in the two channels to be summed with their origi~al or uncompensated values. In the modification illustrated in Figure 5, the sine and cosine outputs of the current servo 31 are used in essentially the same manner to generate the compen- `
sating si~nal~ as direct current signals however, the summing ~:
of these signals wi*h the original data is accomplished directly at the ~lux valve 11 by feeding back the compenæating signals as direct currents into the flux valve legs themselves so as to, in eect, compensate the output of the flux valve itself broadly in accordance with the concept of the above referenced Depp patent 2,852,859.
Referrin~ now to Figure 5, similar reference numeral~ are used ~o designat~ elements corresponding to those found in Figures lA through 4; ele~ents not found in the latter figures are identified by reference numerals in the three hundreds. It will be ~een that the embodiment o Figure 5, like ~hat of Figures lA and 1~, employ~ in serial arr~y ~ reference signal generator 2, a ~lux valve 11, blocking capacitors 169 17, and :~ 18, an input Scott tee trans~ormer 21, a current servo 31, an automatic gain control 34, a dual channel modula~or 45, power ampll~iers 52 and 53, an output Scott tee transformer 60, and a utllization device 64. In a mannar genexally slmilax to that ~ :
employed in Figure lA with respect to current ~ervo leads 10 20~
;
S~3~
1 and 10a, the respective correction ourrents are fed to summation points 320 ~ 321 of Figure 5 so that they flow through their respective legs of flux valve 11 to ground/ being ~locX fxom flowing into the Scott tee transformer 21 by capacitors 16, 17 and 18.
In the lower portion of Figure 5 ~ the inde~ compensation circuit 37 is schematically illustrated. The similarity with the corresponding structure of Figure 3 will be immediately apparent and the simplificatioD of the illustration correspond-ingly apparent. Thu~, the sin ~ and cos ~ direct current signaloutputs of the current servo 31 on leads 32J 33, re~pectively, are alternately appli~d aftar one is inverted b~ inverter 340 to ~-the input of a variable gain amplifier 120a through swikch means 300 corresponding to transistor ~witches 107, 109 of Figure 3~
The gain of amplifier 120a is illustrated schematicaily as being ::
controlled by an adjustment knob 37a corresponding generally to the gain control of amplifier 120 of Figure 3 by Xnob 37a and potentiometer 113 in accordance with the value ~ The ou~put of amplifier 120a i5 similarly al~ernately swi~ched to two branch :
20 lead~ as in ~igure 3 by mean~ o~ switch 301 corresponding , . -.: :
generall~ to tran~istor switches 122, 123 o Figure 3. The control of switches 300 and 301 of Figure 5 is the same as that ~ ~
o~ Figuxe 3, but is illustrated for convenience schematically in ::
Figure~5 by switch control means 302 controlled, for example~ by ~he 400 Hz. source 2~ In Figure 3, the output~ o~ ~witches 122 ~-and 123 are applied ~o two branching circuit~ inclu~ing ~mpli- .
fier~ 128 and 129 for modifying or summing with the orlginal ~ :
~in ~ and cos ~ dir~ct cuxrent outputs o~ the current servo 31 . :
thr:ough~amplifiers~l45 and ~55~ On the other hand, two output ~30 bra~ches of ~witch 301 of Figure S ar~ applied correspondingly ';,'' . "
' '~, .. .
2:1:`, 55~
1 to control direct cuxrent ~low for supplying comp~nsating currents in the pxoper ratio to the 120 spaced inductor coils 13, 14, and 15 of flux valve 11 for effecti~e summing with the original sources of the sin ~ and co~ ~ ~ignals oE the current servo 31. In Figuxe 5, these direct current: ratios are deter-mined by the selected re~istor~ 303, 304, and 305, in the ratios indicated~ The c-urrents from resistors 304 and 305 are applied to flux valve winding 15, while that from re~.istor 303 is supplied to flux valve winding 13. If desired, the resistor capacitor circuits 306 and 307 may be used to reduce transient effects of switches 300, 301. ~ :~
Thus~ as in Figure 3, the apparatus of Figure 5 ~erves to provide index error compensation thxough the time sharing of a single amplifier 120a between the sin ~ and cos ~ channels by alternate operation of the ~witches 300, 301, the gain of the amplifier 120a being controlled by a single control element 37a ~ :
in ac~ord~nce with the magnitude of the error. Such operation in~ures that the ~mount of sln ~ current supplied to the flux valve legs 13, 14, 15 and contribut:ing to the cos S~output channel o current servo 31 is identical to the amount of cos current subtracted ro~ the flux valve legs 13, 14, 15 and con-tributing to the sin ~ output channel of current servo 31.
A modifica~ion o~ the cardinal and intercardinal two cycle error compensator of the heading repeater system of Figure 4 is ~hown in ~igure 5. Again, the ~ignificant feature of the embod-iment re~ides in the manner in which the compensating si~nals : are su~med with the primary signals, i r e ., at the flux valve 11 rather than at the output o~ the current servo 31. In Figure 5, ~or the cardinal two-cycle error compensatisn, the sin ~ and ao~ ~ direct eurrent outputs o~ current servo 31 o~ leads 32 and --".
22.
~5~
1 33 are summed together in a summing circuit, ~chematicallyillustrated at 310, prior to supply to the input of variable g~in ampli~ier 188a~ The summin~ circuit 310 of Figure 5 cor-responds to the summing network 177-180 of Figure 4, while the gain of the amplifier lB8a is illustrated schematically as being varied by the adjustmen~ of knob 48a in accordance with the magnitude ~ corresponding to the tangent of t:he desired correc-tion. As in Figure 4, the output of amplifier 188a is coupled ~ :
through branching leads and the respective coopera-ting resistors 311 and 312 to summation circuits 320 and 321. Instead of being added back into the uncompensated ~in ~Jand cos ~ channels at amplifiers 200 and 201 in Figure 4, the compensation curr~nts .
are employed in the strengths indicated in the drawing of Figure 5 so as to be fed directly into the winding legs 13 and 15 of .. .
flux valve 11. The currents are effectively summed with the flux valve winding outputs which contribute to the sin ~ and ~ :
cos ~ signal outputs of the current servo 31.
~ he intercardinal two c~cle error compensating signal is similarly applied to the flux valve windings. The direct current sin ~ signal outpu~ of the current servo 31 i5 app:Lied to variabl~ gain amplifier l9ga, th~ gain of which is varied by knob 48b in accordance with the magnitude of the required cor- ~;
rections, amplifi~r l99a of Figure 5 corresponding to the variable impedance 199 and amplifier 200 of Figure 4. The output of amplifiex 199a is modified by resistoxs 313 and 314 in accord with the ratios indicated in Figure 5 for application to the respective summation elements 320 and 321 and thu~ to ~he wind- :
in~s 13 and 15 o ~lux valve 11, so that the intercardinal two cycle correction signal is effectively summed with the flux valve winding outputs contributing to th~ outpu~s of the current servo 31.
, , ' ' , .
23- .
9 ~5~
1 Thus, in the modificat.ion of Figure 5, the index and two cycle error compensation signals are g~nerated from the sin ~
and cos ~ direct current outputs of the curr,ent servo 31 and are then Eed back into the appropriate flux valve indicato:r w.indings in the required ratios so that the flux valv~s output supplied to the current servo 31 is compensatedO It will be noted in the Figure S embodiment that the feedback compensation signals are generated from the current servo outputs prior to the latitude compensation automatic gain stage 34. This is desir-10 able because the compensating direct current signals supplied :~
to the valve windings a.re essentially associated with the direc-tion of the magnetic field sensed by the lnductors and in this sense are not related to the latitude gain compensation.
; ~ -.
24.
.. :
:.
.- , . . ~, , . , :, . , , i .. .. .
have been useful in providing adequate magnet:ic field comperlsation in most circumstances, the compensating signa:Ls ~ctually compen-sat~ only for variation in the horizontal magnetic field componen~
and generally do not addition~lly correct ~ully for gain changes caused by co~ponent variations or due to temperature or power supply voltage drifts or to component aging. Furthermore~ the characteristics of the individual gain control elements of the individual channels of the data system may vary without proper corrective relative adjustments whereby two-cycle transmission errors are induced within automatic ~ain control stages.
The improved system di~closed by J. R. Erspamer and G. W. .
Snyder in the United States pate~t 3,784,753, issu~d January 8, ~-1974 ~or a "Multiplexed Gain Control for a Synchro Data Trans-mis~ion System" sought more ully to overcome these prior æ t deects by a relati~ely complex and expensive correction circuit.
Though it generally overcame such defectsJ it was found tha~ some undesirable two cycle error could be generated in its relatively complex automatic gain control stage, and that a simple way was needed ~or identically changing the gains of both of the sine ~ i a~d oslne channels of the data transmission system, but retain~
ing th~ advantages o~ the concept of patent 3,784,753 Prior art ~ystens have additionally sou~ht ~o provide correc ion ~:or the cardinal heading error in compa~s data ::
transmission systems by u~e of networks including precision - :~
. :, -3~ ~ di~erential synchros or ganged dual potentiometers which ~ust :
traa~ each other with high preci~ion i they are not themselves i to in~roduce errors. According to the present invention, the expense of obtaining such selected synchros or precision potentiometers is desirably eliminated. Index angle error was similarly corrected in prior compass data ~ransmission systems by using precision synchros or ganged dual potentiometers of similar quality. It is found increasingly desirable to substitute simple and less expensive networks permitting single adjustment control for each of these correction purposes and, at the same time, retaining a high degree of precision.
The present invention provides means for correction of undesirable changes in the signal amplitudes in multiple channel Elux va1ve data repeater systems partly by the employment of a simple common automatic gain control in a circuit configuration which not only compensates for earth's magnetic field strength changes, but also corrects for the effects of other error sources without introducing the errors of prior art systemsO The novel control circuit of the present invention monitors the data repeater control signals near the inputs to the utilization device3 rather than merely at the outputs of the flux valve. By monitoring the inputs at the utilization device, and by using the data repeaker excitation voltage as a switching reference, the gain control, being part of a closed feed back loop, compensates not only for 2Q changes in the operating latitude bu~ also for gain changes caused by varia-tions of component parameters and by other effects without itsel introducing naw errorsn According to a broad aspect of the present invention, there is provlded in a magnetic compass system for navigable craft including a magnetic field detector of the flux valve type mounted in the craft for sensing the ; direction o the earth~s magnetic field relative to the crat~ the combination for correcting for any errors in the orientation of said detector relative to the di~ection axis o~ said aircraft comprising, magnetic field detec~or means ~ including a plurality oE inductor elements angularly disposed on said cxaft for sensing the direction and magnitude of the correspondingly disposed horizontal components of the earth's magnetic field relative to the craft and -1~655~
for provid.ing a corresponding plurality of alternating signals proportional thereto, signal processing means coupled with said inductor elements and responsive to said plurality of alternating signals for pro~i.ding first and second direct current signals proportional in sense and magnitude to predeter-mined functions of said earth's magnetic field directionl amplifier means having an input an~an output and means for effectively controlling the gain thereof, means for modi~ying said first and second direct current signals in accordance with the output of said amplifier means, input and output switching means at the respective input and output of said amplifier means, said input switching means being connected to receive said first and second signals and said output switching means being connected to supply ~he output of said amplifier means to said modifying means, means al~ernately and simultaneously controlling said switching means such that when said amplifier mecms is ;
switched to receive said first signal its output is switched to supply the means for modifying said second signal and vice versa whereby to insure that :- . :. :
the amount of said first signal modifying the second signal is identical to the amount of said second signal modifying said first signal, and means for controlling said amplifier gain control means in accordance with said :
orientation errorO ~
The invention will now be described in greater detail with ~ :
reference to the accompanying drawings, in which:
Figures lA and lB illustrate, partly in block diagram form, the principal elements of the invention and their electrlcal interconnections; -- :
Figure 2 is a portion of Figures lA and lB showing details o~ the :-novel automatic gain control circuit;
Figure 3 is a detailed circuit diagram of the novel index error angle compensa$or of Figure lB;
.. . .
F~ure 4 is a detailed circuit diagram of the novel two cycle compensator also of Figure lB; and : Figure 5 lS a bloclc diagram of an embodiment of the invention alternative to that of Figures lA and lBo : ~ ~ 5 . :
:
s~
In Figures lA and lB, the novel compensated compass system in-cludes a magnetic azimuth detector or flux valve 11 wh.ich may be of the general type disclosed in the M. C. Depp United States patent 2,852,859 for a "Flux Valve Compensating System", issued September 23, 1953 and assigned to Sperry Rand Corporation. Other details of such flux valve devices are disclosed in the D. J. Kesselring United States patent 3,573,610, issued April 6, 1971, in the D. JO Kesselring et al United States patent 3,641,679, issued February 15, 1974, and in the United Sta~es patent application SO No 380,523 for "A Flux Valve Apparatus for Sensing Both Horizontal and Vertical Components of an Ambient Magnetic Field", filed July 18, 1973, issued as patent 3,873,914 March 25, 1975 and assigned to Sperry Rand CorporationO
Flux valve 11 is excited by alternating current source 2, which may be a : :
conventiQnal 400 Hz. oscillator or signal generator and which is coupled to : ..
exc.itation winding 12 of the flux valve 11. .
.
: ~ ' .
: : :
- 5a -::
: ' ' ~6~
As disclosed in the aforementioned Depp and ~Cesseling patents, f lux valve 11 ha~; three wye~collnected inductive windings 13, 14, and 15 on a corresponding wye-~hap~3dl core, the winding legs meeting at a common grounded terminal E'. The terminals of windings 13, 14, and 15 opposite terminal F are respectively labelled A, B, and C. Terminals A, B, and C may, i desired, be supplied with single cycle compensation slignals from a single cycle compensator (not shown) of the general type shown in the af orementioned U. S. patent ~, 85 2, 859 ..
10Terminal A of flux valve 11 is connected via a blocking ~:.
capacitor 16 to one winding 20 of a Scott tee transformex 21, while $erminals B and C are connected via re~pective blocking capacitors 17 and 18 to the respective ends of a second input ~: :
winding 22 of Scott tee transfoxmer 21~ Winding 22 has a center tap connected to the other end of winding 20.
As is well known, the signal outputs of windings 13, 14, ::
and 15 ha~e a requency double that applied to exci ation winding 12. The ~requency doubled cosine output o winding 23 of trans- :
former 21 and its frequency doubled sine output in winding 27 are conne~ted to current servo loop 31. Additionally supplied to current ~ervo loop 31 via lead 2~a is the output of ~requency doubler 29. Since frequency doubler 29 i~ excited ~y generator 2, its output on lead 29a will have an 800 Mz frequency and serveq as a re~çrence sigDal source for servo 31~
As describçd in detail in the D. H~ Baker~ F. H~ Kallio U~S. patent-3,67~,593 for a "Compass System and Components :Thexe~or Having Automatic ~ield Cancellation", issued July 25, ..
19~2 to Sperry Rand Corporation, current servo 31 supplies out puts on leads 32 and 33 which are direct current signals re~pec-~30 tlvely proportional ~n amplitude to the sine and cosine of craft 6. . ~ ~.
5~ 3 1 magnetic heading ~Hm sin yJand Hm cos y~). Accordingly~ the horizontal components of the earth's magnatic field sensed by the flux value windings 13, 14, and 15 are resolved into sine and cosine component.values that are then convert~d by current servo 31 into proportional direct currents on leadls 32 and 33. As taught in the aforementioned Baker et al patent, these direct current components are fed back via leads 10 and lOa into wind-ings 13 and 15 o~ flux valve 11, which currents tend to cancel the earth's magnetic field therein. The feed back arrangement and its many advantages are d~scussed in detail in th~ afore-mentioned patent 3,678,593,includlng closed loop operation afford-ing high accuracy outputs in the ~orm of analog direct current outputs proportional to the sine and cosine of craft magnetic headlng .
Accordingly, the 800 Hz., three-wire magnetic azimuth inormation derived by the horizontal magnetic field detector or flux valve 11 is converted to direct current signals proportional to the sine and cosine of craft heading by the cooperation of Scott tee transformer 21 and current servo 31. The magnitudes of ~he outputs on leads 32 and 33 are thus a function of craft magnetic azimuth or heading and the intensity of the horizontal co~ponent of the earth'~ magnetic field. The variation in the magnitude of the sine and cosine outputs caused by any change in magnetic ield strength Hm affects only the output gradient (volts per a~imuth degree~ and does not change the trigonometric rel~tionship of khe input magneti¢ heading angle ~ and the output vol~age~ of current servo 31, which may therefore be expressed as follows:
V32 ~ Xl sin and . . .
V 3 - K c~ ~ (2) ' .':
7.
;55~
1 where ~1 allows for the gain of ~urrent servo 31 and has dimensions of volts per oersted.
The signals V32 and V33 on leads 32 and 33 serve as two inputs to the automatic gain control circuit 34, which circui.t also receives certain fed back signal~ on leads 56 and 57. As will be further discus~ed, the fed back signals arise at the outputs o buf~er amplifiers 52 and 53 after the outputs of automatic gain control 34 are processed at least by dual channel modulator 45. To understand the operation o~ the gain control circuit 34, the presence of the index error compensator 37 and khe two-cycle compensator 48 may be ignored for the moment as a matter of convenience.
The final output of the compas~system supplied by leads 61, 62, and 63 to an aircraft navigation system or other utilization clevice 64 is usually re~uired to be useful in a three wire synchro data transmitter s~stem and to consist of proportional voltages between pairs of such leads, as betwee~
leads 61 and 62, 62 and 63, 63 and 61. These may nominally be 11.8 volts, for example, and mu~t be maintained at a constant gradient in the interes~ of me~ting required compass accuracy over a wide range of horizontal magnetic field strengths Hm.
Becau3e ~he output o~ flux valve 11, and there~ore the output of current servo 31, has a gradient which i~ directl~ proportional in magnitude to the horizontal magnetic field ~trength which, of course, varie~ with latitude~ the automa~ic gain controlstage 34 is required to hold the system output signals at leads 61~
62, and 63 at the de~ired nominal 1108 volt leg-to le~ constant ~radient.
For this purpose, the direct current outputs on leads 3S
; 30 and 36 of gain control 34 are suppliad to the con~entional dual channel modulator 45, each o~ the two individual channels of 8. - :
1 which are supplied by lead 2a with the 400 Elz. reference signal output of generator 2. The direct current signals on lead~ 35 and 36 are modulated by the 400 ~Iz. alternat:ing current signal in khe conventional manner so that 400 Hz. signals appear on leads 46 and 47, proportional respectively to the sine and cosine of the magnetic heading of the craft. After individually separate supply to buffer ampli~iers 52, 53, equally amplified versions of these signals appear on leads 54, 55 to which the feed back leads 56 and 57 are respectively connected.
The automatic gain control 34, shown in greater detail in Figure 2, monitors the gradient at the output lea~ls 5~, 55 of buffer amplifiers 52, 53, respectively, compares the result to a reference voltage level, then varies the system gain accordingly by control o~ the gain of automatic gain control circuit 34. If the gradient at the outputs of buffer amplifiers 52, 53 of Figure lA is less than a predetermined level, the voltage gain of circuit 34 i9 increased to bxing the output of the buffer amplifi~rs 52, 53 up to the proper level~ The output of the buffer amplifiers 52, 53 and the voltage ~radient is similarly controlled. The signal levels at output leads 35, 36 of the automatic qain control : 34 are ultimately passed through the outpu~ Scott tee transformer 60~ The outputs of transformer 60 are therefore fully independent of any earth~s magnetic field strength variation. ~ :
Thu8:
V3~ - K2 ~in ~ t3 . .
~ and.
:~ V36 = ~2 ~os ~ (4) : wher~ K2 is a new propor~ionality con~tant.
Autom ~ gain control circui1: 34 is desi~ned to prevent thc inkroduction of any stand-off or unbalance between trans~ -mls~1on channels, resulting in cyclic errors, into the c:raft g~
~55g~
1 heading output data. The individual gains of the sine and cosine channels are now identically controlled and there are no of~-set voltages induced into the direct current s.ignal~ representing sine and cosine of craft magnetic heading. As is seen in more detail in ~igure 2, the direct current .signals at leads 32 and 33 are, as before, provided by the cooperative action o~ ~lux valve 11, Scott tee transformer 21, and current servo 31, an~ are respec- :
tively proportional in amplitude to sin ~ and cos ~ . Output lead 32 is coupled in series through resistor 75, ~unction 76, . .
re~istor 77 and input lead 35 to one channel of the dual modulator 45, At input lead 35 is a capacitor 78 coupled tc ground and forming a low pass filter with resistor 77. Likewise, the second output lead 33 i5 coupled in series through resistor 79, junction 80, resistor 81, and input lead 36 to a second channel of dual modulator 45. At input lead 36 is a capacitox 82 forming a low pass filter with resistor 81. Switching or c~opper transistors B3 and 84 are respectively coupled to ground from junctions 76 and 80 and control current flow through their emitter and collector electrodes in accordance with their respective base voltages.
The direct current signals on leads 32 and 33 are chopped by transistors 83 and 84, respectively, and after ~moothing by low pass filters 77-78 and 81-82~ form direct currents tha-t axe ~.
individually modulated in dual channel modulator 45 ~y the 400 Hz. re~erence signal on lead 2a. These dual channel output vol~ages are directed by Scott tee transformer 60 as three-wire : ~ynchro data to a navigation system or oth~r utilization device 64.
For purposes of controlling the automatic gain contxol ~ ~
30 ~ircui~ 34, the same 400 Hz. modulated output currents on leads : :
54 and 55 axe respectivel~ coupled by leads 56 and 57 to a ~ ~
~.:
~ ~ :
.: . .
10.
.. ..
js~
1 constant amplitude, variable phase circuit comprising resistor 95 and capacitor 96 coupled in ~eries with l~eads 56, 57 at junction 97. Circuit 95 96 is of the general kind discussed in ~he D, A~ Espen UOS. patent 3,548,28~, entitled "Synchro Data Transmission Apparatus ~aving Discrete Gain Changing to Compen-sate for Undesixable Signal Gradien~ Variation", issued December 15, 1970 and in the D~ A~ Espen U.S. patent 3,617,863, entitled "Constant Amplitude Variable Phase Circuit", issued November 2, 1971, both patents being assigned to Sperry Rand. The constant amplitude, variable phase signal found at junction 97 is recti-fied by diode 94 and appears as a variable unipolar voltage at one input of a conven~ional integrating operational amplifier 92 having its output coupled by capacitor 91 to its same input. To the second input of amplifier 92 is coupled through resi~tor 93 a stable positive unidirectional reerence voltage from a suit-able ~ource (not shown~ connected to terminal 98. As ~hown in ~:
the drawing, amplifier 9~ and its a~sociated circuit act as -~
conventional comparator means or in ~ffect comparing the ou~put .
gradie~t on leads 54, 55 with the fixed level voltage at terminal ~8 yielding an integra~ed output as a function of the diffexence of the two voltage levels at leads 97 and 98.
The posi~ive signal at the output of device 92 i~ coupled through re~istor as to one input o~ ampliier 87, to the other input of which is supplied at terminal 90a and through re~i~tor 89 the 400 Hzo excitation signal from generator 2. Under control of i*s varying amplitude direct current and constant ampli~ude alternating input current~, circuit ~7 acts as a conventional variable pulse-width generator for supplying a 400 H~ vaxiable pulse-width signal at junction 74.
The variable pul~e~width signal is coupled in parallel from junction 74 through t~e respective resistors 85, 86 to the base 11, . ,: : : . . . - ... . . ..
~v~
1 electrodes of chopper transistors 83 and 8~ to control the rela tion of the conduction to non-conduction times of these switchiny transistors~ The transistors 83, 84 are synchronously corlducting at the same time and then are both non~conducting for a con~
trolled period of time depending upon the pulse width o~ the out-put of amplifier 87~ As the non-conducting part o the cycle is increased in time duration, the ~otal current per cycle passing from lead 32 to lead 35, for example, is increasedO In other words, proportionately less of the current a~ailable on lead 32 is dumped to ground. In this manner, the voltage between leads 54, SS is made independent of any ~mplitude variations in the total flux valve data as well as amplitude variation~ resultin~
from other distuxbing factors in the signal channels between current servo 31 and buffer ampli~iers 52 and 53~ Accordingly, the three-wire output supplied to utilization device 64 of Figure 1 by transformer 60 is maintained nominally constant from leg to ;:~ -leg) such as at 11.8 volts.
In the comple~e system as illustrated in Figure 1, the outputs V35 and V36 of the automatic gain control 34 on leads 35 and 35 may be first processed by the novel index error angle compensator 37 prior to 400 Hzo modulation. For this purpo~e, the compensation circuit of Figure 3 is employedD The index ~:
error angle comp~nsated by circuit 37 is pre~ent because of the noxmal difficulty o~ achieving perfect alignment between the air- ~ :
craft fore-aft axt~ and the effective electrical ore-a~t axis of the 1ux valve 11. Accordingly, index angle error compen- :~
sator 37 is provided to permit a manual correction to be made : after ~ystem installation by performing; in essence, the ~ame unction as might be provided by a relatively expensive servo differential which so~e prior art sy~stems have employe~O How-ever~ since installation accuracies are usually within ~10, the ::
:
.
12.
;~ ii5~
1 compensation ~unction may beaccurately performed by the relatively inexpensive circuit of Figure 3 wher~in only a single potenti~
ometer shaft need be adjusted. It will be apparent that the correction is made by the novel compensa~or herein di~closed to the value of angle ~ when it i9 still in the trigonometric form of sin ~ and cos ~ data.
Accordingly, the apparatus of Figure 3 accepts two inputs K2 sin ~ and K2 cos ~ and internally generates two values -K
cos ~ and -K2 ~ sin ~ . The K2 sin ~value and the -~2 ~ cos ~
value are added according to the well known trigonometric .identity to form K2 sin ( ~ ~ ~ ) where ~ + ~ may be used to repre-sent a corrected value of ~ . The K2 cos y value and the -K2 sin ~ values are similarly added to form K2 cos ( ~ + ~ ). In accord with the teachings of the present in~ention, the ~ terms must be identical in both the sine ~nd cosine output channels to effect preci~e compensation; the same source for the ~ term is used in the two channels of the circuit.
In greater detail, the circuit of Figure 3 has, in operation D
negative valued direct voltages respre~enting K2 sin ~ and K2 cos 20 ~ as re~pactive inputs on leads 35, 36, and these are respec~ ~ -tiv~ly supplied directly to inputs o conventional unity gain output amplifiers 145 and 155 at the right side o the figure~
The ~ame two direct neg~tive voltage~ are used in the remaining or major part of the circuit to produc~ compensating voltages also for insertion into amplifiers 145, 15S. For t~e latter purposes, the -sin ~ t~rm on lead 35 is coupled through a con-ventionaI inverting amplifier 103 to the switching tr~nsi~tor 107. Ampliier 103 has its output terminal 104 coupled through a re~i~tor 102 to its input terminal 101 and additionally ha~ a second input terminal coupled to ground through resistor 105.
The -cos ~ term vn lead 36 is coupled dixectly to ~witchin 13.
~3~5~
1 transistor 109. Transistors 107 and 109 are made alternately ~ully conducting and fully non-conducting so that, first, the output of amplifier 103 appears on lead 108a and then, the signal passed by switching transistor 109 appears on lead 108b.
Since both of the leads 10~a and 108b are coupled to the adjus-t-able contact 113a of potentiometer 113, it i.s seen tha-t the signals alternately passed hy switching transistors 107 and 109 are alternately applied to contact 113a for time-sharing purposes .
in the shared amplifier 1~0, The switching transistors 107 and 109 are made alternately conducting under control of a sine wave signal ap~earing on -lead 2b; thi~ signal is conveniently obtained from ~he 400 Hz.
generator 2 of Figure lA, though other regul~x stable-fxe~uency signals may alternatively be employed. In practice, the 400 Hz.
cycle signal on lead 2b is applied by lead 106 to control the con~uction of transistor 107. So that time sharing may be ;:
employed, the signal on lead 2b is coupled via lead ~11, the 180 phase shifter 112, and lead 110 to control the operation vf switching transistor 109. -:~
In this manner, the signals on leads 35 and 36 are .
alternately supplied at the selected contact point o potenti- :
ometer 113, the latter having its opposed terminals 113b and 113c coupled to inputs of operational amplifier 120~ The output terminal 121 of amplifier 120 is coupled to its input at termlnal 113b via resistor 115, and terminal 113c is connected through rssl~tor 114 to ground in conventional a~hionO The input of ampli~ier 120 is thus time 3hared and its output on terminal 121 iB ~pp1ied to a secc~nd pair of switching t~ansistors 122, 123, these transistors being arranged for rontr~lling the series 30 signal flow through the respective resistors 126, 127 to ampll~iers 128, 1~9~ The ef~ective ga.in of amplifier 120 is ~ .
. ' ' 14.
. .
~55i~
1 changed according to the setting of the single control 37a, which control is manually set in accordance with the known magnitud~ of the index error de~ermined as a result of conven-tional ground swinging operations~
Conductivity of switching transistor 122 occurs simul-taneously with the conductivity of switching transistor 109.
Xn like manner, conductivity of switching transistor 123 is made simultaneous with the periods of conductivity of switching transistor 107. This operation is accomplish~d by controlling the conductivity of switching transistor 1~3 according to the si~nal on lead 2b when supplied directly to switching transistor 123 via lead 125. The desired s~nchronous operation of switch ing transistox 122 is accomplished by providing the 180 phase :~
shifted signal from circuit 112 via lead 124 to transistor 122.
In this manner, both channels o the circuit time share the use of the common amplifier 120, ensuring that identical corrections .
are applied to the two channels; l.e" that the amount of the ~:
sine term added to the cosine term is identical to the amo~mt : .
o the coslne term subtracted in the sine channel. It i9 :Eurther observed that adjustment of the single control 37a allows adjust-ment of potentiometer 113 so that both channels are identically set in accord wlth the magnitude of the index error.
The time shared cuxrents alternately flowirlg through switching transistors 122,123 are alternately supplied to the conventional unity gain amplifiers 128/ 129, and the respec~ive outputs on the terminals 132, 133 flow ~hrou~h resistors 141, ~ .
~ 150 to the sam~ respectlve input terminals of ampli~iers 145, ~ -: 155, as a~e connected to the respective leads 35, 36. ThP out- : -puts of amplifiers 145 and 155 may be smoothed by the action of appropriate low:pass ilters so as to remove any 400 Hz, modula~
tion ~xom the outputs appearing in the xespective output leads 383 39. In the embodiment illustrated, the filters are placed '' '::
5g~
1 at the input~ of amplifiers 128 and 12~ and comprise resistors 126, 127 and capacitors 128a, 129a, respectively.
The Mathematical relation expressing the index error as a function o~ the sine and cosine of magnetic heading is:
~' = tan 1 sin ~ - ~ cos ~ (S
where: cos i ~ ~ sin ~
~'~ the compensated output, _ the uncorrected input, and ~ = the tangent of the index errox.
~hus, by adjusting the gain of the time ~hared amplifier 120 in acc~rdance with the value of ~, expxe~sion (5) is satisfied as follows. The output of amplifier 128 is:
V132 - ~K ~ cos ~ (6 when transistors 109 and 122 are conductin~, and the output of amplifier 129 is~
V133 = -K2 ~ ~in ~ (7) when tran~istors 107 and 123 are conducting. The addition at amplifier 145 produces on output:
38 K2 ~sin ~ - ~ cos ~ ) (8) while the addition at ampli*i~r 155 produces an output:
V39 = K2 (cos ~ in ~ ) or: 38 K~ sin y (10) and:
V3~ ~ ~2 cos ~
These direc~ current signals are ready ~or conversion iTI dual channel modulator 45, providing as they do direct current ~ignals in terms of ~ ' containing the desired index angle error compen, sation~as set ~orth in equation (S)~ Thu~, the dual chan~el ~ modulator ~5 supplies on it~ output leads 50, 51 400 Hz. ~ignals ~ -whose amplltude~ are:
.'~
.'" ' , ' " ~ .
:' 16.
,, . ~ , ., . , ,. . :
:
1V46 = K3 sin ~ (12) 47 3 c s ~ ~13) and these signals serve as inputs to the two cycle error compensator 48 Two cycle error in a magnetic field sensor, including the flux valve type of sensor disclosed herain, i~ induced by the presence o~ a so~t iron mass or masses in the vicinity of the flux valve which tend~ to distort the earth's ambient magnetic field thereat. As the name implies, the error is a sinusoidal error and has two complete cycles within 360 of azimuth :rota-tion of the craft. In general, the average location of the oft iron mass relative to the flux valve determines the direction of its effective vector. For convenience, the two cycle co~pensa-tion is accomplished by effectively breaking down the total .:
vector into orthogonal components, one termed the cardinal two cycle error component and the other the intercardinal error : component~ The cardinal two cycle error has extremllm values at heading angle values 0~ 90~, 180t and 270. Intercardinal two cycle error, on the other hand/ has extremum values at 45, 1~5, : ~20: ~ 225, and 315 azimu~h values. Xn the invention o~ ~igure 4, the ; ~ latter error is readily corrected ky placing an adjustable series resistor l9g in the ~eed baclc path of arC. amplifi~r 200 that is excited by lead 461 Adjustment of control 48b in accord with data ;liakeal during installation compa~3s swings then corrects , .
the output at lead S~ in the appropriate mannerr The inter- .~
card1nal two cycle headin~ error .is co~pensated by chan~ing the .~.
ga~n balance hstween the sine and cosine channels supplying outp~ts on the respective leads 50, 51~ : :
Correction of the cardin~l heading error lS accomplished ~y ~ ~ . . ..
~ 30~ the ~:1mpl~ circuit o~Figure 4, using a sin~le adjustment 48a ::
,. ..
17- :
1 and a common circuit stage in a manner mlnimizing error sources and characterized by simplicity. The amount o adjustment of control 48a i~ also determined by the installation ground 3wing-ing process. The X2 sin ~ signal on lead 46 is supplied via lead 175 through resistors 177 and 180 to the re~pective inputs of diferential operational amplifier 188. Thevalue K2 co~
from lead 47 is added through resistor 178 al; terminal 183 to the K2 sin ~ ' term ~imilarly, the term K2 sin ~ ' from lead 46 is added through resistor 180 at terminal 185 to the K2 cos 10 term. Amplifier 188 has a resistor 187 coupled between its output 189 and the input terminal 183 in conventional f a~hion .
A variable resis~or 186 with an adjustable control 48a is coupled : -between terminal 185 and ground. The variable resistor 186 con- :
stitutes the single cardinal heading error adju~tment, its varia-tion affecting the effective gainYof amplifier 188~ According to the setting of control 48a, a compensating voltage appear~
at the output 189 o~ amplifier 188:
: V18g = - ~lsin ~ ' + cos ~') t14) The single value V189 is coupled at junction lgO ~o branching : 20 leads for ~upplying this signal throu~h r~sis~ors 192, 193 t.o the input of respective amplifiers 200 and 201~ A~ n~ted previously, amplifier 200 has a varia~le resistor 199 coupled between its output 203 and it~ inpu~ lead l~S. The other input to ampli~ier 200 i~ connected ~hrough resistor 196 to ground. A
furth~x amplifier 201 is supplied with the signal K2 CQS ~ ~ from :
the lead 47 through resistor 194 and is similarly provided wi~h a re~istor 202 connecting its output 204 to its input lead 197. :
: : It similarly employ~ a r~si~tor 198 coupled between a second input and ground. Amplifiers 200 and 201, through th2 respective : :
connectors l9S and 197, serve a~ adding and inverting circuit~
.
18.
- . ~ . . . .
1 so that the K2 sin ~' term on lead 46 has added to it the correction term appearing at terminal 190 and the summation is found on output lead 50. In a similar manner, the K2 cos ~' signal supplied on lead 47 is added to the compensating signal on junction 190 by amplifier 201 and its associated circuit, an inverted ~ignal being generated on output lead Sl. In this manner, the voltage VsQ is:
V50 , K4 Esin ~ (sin l~l + cos ~ '~ (lS) and that on output lead 51 is:
V51 = K~ rcos ~ -t ~(sin ~' ~ cos ~'~ (16) In equations ( 15 ~ and (16), the oew value K4 may include the effect of the adjustment of resistor 199~ Thus, the volta~e on .:
output lead SO is sin ~ ' and the voltage on output lead 51 i~
cos ~'', where ~ represents ~ corrected both for cardinal and intercardinal heading errors. From equations tl5) and ~16), it is ev.ident that the value of the corrected angle ~'~ i5 expressed by the following equation:
= tan_ ~1 + ~ cos~ ' + ~sin ~
~ '~ being the final output headlng ~alue components corrected 20 for cardinal and intercardinal ~o cycle errors. It is seen ~ :
that amplifier 188, the effective gain ~ of which is controlled ~:
by the setting o~ the variable potentiometer 186 t cooperates in .
the circuit in generating the unction ~ ~sin ~ ' + cos~
and this function is added in the sine an~ cosine channels by : the respective action of amplifiers 200 and 201 and their associated circuitsO It is observed that correction of the ...
cardinal heading two cycle error is accomplished by manual opera-ion of a single adjustment. Fuxthermore, the single stage . .
a~ociated with amplifier 188 mini.mizes potential error sources :;
. .
and aids in simpli~ying the adjustment procedure.
.
:., :.
19~
1 It will be understood that the invention may be employed in alternative oxms and that the compass system of Figures lA
and lB may be modified within the scope of the claims appended hereto, for example, as illustrated in Figure 5. In the embod-iment o Figures lA and 1~, the index compe:nsation and two cycle error compensation signals are generat~ed from the sine and oosine outputs of the current servo 31 and are re-applied downstream in the two channels to be summed with their origi~al or uncompensated values. In the modification illustrated in Figure 5, the sine and cosine outputs of the current servo 31 are used in essentially the same manner to generate the compen- `
sating si~nal~ as direct current signals however, the summing ~:
of these signals wi*h the original data is accomplished directly at the ~lux valve 11 by feeding back the compenæating signals as direct currents into the flux valve legs themselves so as to, in eect, compensate the output of the flux valve itself broadly in accordance with the concept of the above referenced Depp patent 2,852,859.
Referrin~ now to Figure 5, similar reference numeral~ are used ~o designat~ elements corresponding to those found in Figures lA through 4; ele~ents not found in the latter figures are identified by reference numerals in the three hundreds. It will be ~een that the embodiment o Figure 5, like ~hat of Figures lA and 1~, employ~ in serial arr~y ~ reference signal generator 2, a ~lux valve 11, blocking capacitors 169 17, and :~ 18, an input Scott tee trans~ormer 21, a current servo 31, an automatic gain control 34, a dual channel modula~or 45, power ampll~iers 52 and 53, an output Scott tee transformer 60, and a utllization device 64. In a mannar genexally slmilax to that ~ :
employed in Figure lA with respect to current ~ervo leads 10 20~
;
S~3~
1 and 10a, the respective correction ourrents are fed to summation points 320 ~ 321 of Figure 5 so that they flow through their respective legs of flux valve 11 to ground/ being ~locX fxom flowing into the Scott tee transformer 21 by capacitors 16, 17 and 18.
In the lower portion of Figure 5 ~ the inde~ compensation circuit 37 is schematically illustrated. The similarity with the corresponding structure of Figure 3 will be immediately apparent and the simplificatioD of the illustration correspond-ingly apparent. Thu~, the sin ~ and cos ~ direct current signaloutputs of the current servo 31 on leads 32J 33, re~pectively, are alternately appli~d aftar one is inverted b~ inverter 340 to ~-the input of a variable gain amplifier 120a through swikch means 300 corresponding to transistor ~witches 107, 109 of Figure 3~
The gain of amplifier 120a is illustrated schematicaily as being ::
controlled by an adjustment knob 37a corresponding generally to the gain control of amplifier 120 of Figure 3 by Xnob 37a and potentiometer 113 in accordance with the value ~ The ou~put of amplifier 120a i5 similarly al~ernately swi~ched to two branch :
20 lead~ as in ~igure 3 by mean~ o~ switch 301 corresponding , . -.: :
generall~ to tran~istor switches 122, 123 o Figure 3. The control of switches 300 and 301 of Figure 5 is the same as that ~ ~
o~ Figuxe 3, but is illustrated for convenience schematically in ::
Figure~5 by switch control means 302 controlled, for example~ by ~he 400 Hz. source 2~ In Figure 3, the output~ o~ ~witches 122 ~-and 123 are applied ~o two branching circuit~ inclu~ing ~mpli- .
fier~ 128 and 129 for modifying or summing with the orlginal ~ :
~in ~ and cos ~ dir~ct cuxrent outputs o~ the current servo 31 . :
thr:ough~amplifiers~l45 and ~55~ On the other hand, two output ~30 bra~ches of ~witch 301 of Figure S ar~ applied correspondingly ';,'' . "
' '~, .. .
2:1:`, 55~
1 to control direct cuxrent ~low for supplying comp~nsating currents in the pxoper ratio to the 120 spaced inductor coils 13, 14, and 15 of flux valve 11 for effecti~e summing with the original sources of the sin ~ and co~ ~ ~ignals oE the current servo 31. In Figuxe 5, these direct current: ratios are deter-mined by the selected re~istor~ 303, 304, and 305, in the ratios indicated~ The c-urrents from resistors 304 and 305 are applied to flux valve winding 15, while that from re~.istor 303 is supplied to flux valve winding 13. If desired, the resistor capacitor circuits 306 and 307 may be used to reduce transient effects of switches 300, 301. ~ :~
Thus~ as in Figure 3, the apparatus of Figure 5 ~erves to provide index error compensation thxough the time sharing of a single amplifier 120a between the sin ~ and cos ~ channels by alternate operation of the ~witches 300, 301, the gain of the amplifier 120a being controlled by a single control element 37a ~ :
in ac~ord~nce with the magnitude of the error. Such operation in~ures that the ~mount of sln ~ current supplied to the flux valve legs 13, 14, 15 and contribut:ing to the cos S~output channel o current servo 31 is identical to the amount of cos current subtracted ro~ the flux valve legs 13, 14, 15 and con-tributing to the sin ~ output channel of current servo 31.
A modifica~ion o~ the cardinal and intercardinal two cycle error compensator of the heading repeater system of Figure 4 is ~hown in ~igure 5. Again, the ~ignificant feature of the embod-iment re~ides in the manner in which the compensating si~nals : are su~med with the primary signals, i r e ., at the flux valve 11 rather than at the output o~ the current servo 31. In Figure 5, ~or the cardinal two-cycle error compensatisn, the sin ~ and ao~ ~ direct eurrent outputs o~ current servo 31 o~ leads 32 and --".
22.
~5~
1 33 are summed together in a summing circuit, ~chematicallyillustrated at 310, prior to supply to the input of variable g~in ampli~ier 188a~ The summin~ circuit 310 of Figure 5 cor-responds to the summing network 177-180 of Figure 4, while the gain of the amplifier lB8a is illustrated schematically as being varied by the adjustmen~ of knob 48a in accordance with the magnitude ~ corresponding to the tangent of t:he desired correc-tion. As in Figure 4, the output of amplifier 188a is coupled ~ :
through branching leads and the respective coopera-ting resistors 311 and 312 to summation circuits 320 and 321. Instead of being added back into the uncompensated ~in ~Jand cos ~ channels at amplifiers 200 and 201 in Figure 4, the compensation curr~nts .
are employed in the strengths indicated in the drawing of Figure 5 so as to be fed directly into the winding legs 13 and 15 of .. .
flux valve 11. The currents are effectively summed with the flux valve winding outputs which contribute to the sin ~ and ~ :
cos ~ signal outputs of the current servo 31.
~ he intercardinal two c~cle error compensating signal is similarly applied to the flux valve windings. The direct current sin ~ signal outpu~ of the current servo 31 i5 app:Lied to variabl~ gain amplifier l9ga, th~ gain of which is varied by knob 48b in accordance with the magnitude of the required cor- ~;
rections, amplifi~r l99a of Figure 5 corresponding to the variable impedance 199 and amplifier 200 of Figure 4. The output of amplifiex 199a is modified by resistoxs 313 and 314 in accord with the ratios indicated in Figure 5 for application to the respective summation elements 320 and 321 and thu~ to ~he wind- :
in~s 13 and 15 o ~lux valve 11, so that the intercardinal two cycle correction signal is effectively summed with the flux valve winding outputs contributing to th~ outpu~s of the current servo 31.
, , ' ' , .
23- .
9 ~5~
1 Thus, in the modificat.ion of Figure 5, the index and two cycle error compensation signals are g~nerated from the sin ~
and cos ~ direct current outputs of the curr,ent servo 31 and are then Eed back into the appropriate flux valve indicato:r w.indings in the required ratios so that the flux valv~s output supplied to the current servo 31 is compensatedO It will be noted in the Figure S embodiment that the feedback compensation signals are generated from the current servo outputs prior to the latitude compensation automatic gain stage 34. This is desir-10 able because the compensating direct current signals supplied :~
to the valve windings a.re essentially associated with the direc-tion of the magnetic field sensed by the lnductors and in this sense are not related to the latitude gain compensation.
; ~ -.
24.
.. :
:.
.- , . . ~, , . , :, . , , i .. .. .
Claims (7)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. In a magnetic compass system for navigable craft including a magnetic field detector of the flux valve type mounted in the craft for sensing the direction of the earth's magnetic field relative to the craft, the combination for correcting for any errors in the orientation of said detector relative to the direction axis of said aircraft comprising, mag-netic field detector means including a plurality of inductor elements angular-ly disposed on said craft for sensing the direction and magnitude of the correspondingly disposed horizontal components of the earth's magnetic field relative to the craft and for providing a corresponding plurality of alter-nating signals proportional thereto, signal processing means coupled with said inductor elements and responsive to said plurality of alternating sig-nals for providing first and second direct current signals proportional in sense and magnitude to predetermined functions of said earth's magnetic field direction, amplifier means having an input and an output and means for effectively controlling the gain thereof, means for modifying said first and second direct current signals in accordance with the output of said amplifier means, input and output switching means at the respective input and output of said amplifier means, said input switching means being connected to receive said first and second signals and said output switching means being connected to supply the output of said amplifier means to said modifying means, means alternately and simultaneously controlling said switching means such that when said amplifier means is switched to receive said first signal its out-put is switched to supply the means for modifying said second signal and vice versa whereby to insure that the amount of said first signal modifying the second signal is identical to the amount of said second signal modifying said first signal, and means for controlling said amplifier gain control means in accordance with said orientation error.
2. The combination set forth in Claim 1 wherein said first and second signals are proportional to the sine and cosine functions of said earth's magnetic field direction.
3. The combination set forth in Claim 2 wherein said first and second signals as supplied to said input switching means are of opposite polarities.
4. The combination set forth in Claim 2 wherein said switching means comprises a first pair of electronic switches, one for connecting said first signal to said amplifier input and the other for connecting the amplifier output to the means for modifying said second signal and a second pair of electronic switches, one for connecting said second signal to said amplifier input and the other for connecting said amplifier output to the means for modifying said first signal.
5. The combination set forth in Claim 4 wherein said switch control-ling means comprises a single phase alternating current source and said one pair of electronic switches is responsive to one phase thereof and said other pair of electronic switches is responsive to the opposite phase there-of.
6. The combination set forth in Claim 1 wherein said means for modi-fying said first and second signals in accordance with the said amplifier output switching means comprises: a first summing circuit responsive to said first signal and modified second signal, and a second summing circuit responsive to said second signal and said modified first signal.
7. The combination set forth in Claim 1 wherein said means for modifying said first and second signals in accordance with said amplifier output switching means comprises: a plurality of direct current signals of predetermined ratios having values dependent upon the angular orienta-tion of said magnetic field detector inductor elements, means for varying the relative magnitudes of said direct current signals in accordance with said amplifier output, and means for applying said varied direct current signals to predetermined ones of said detector inductor element whereby to vary said plurality of detector signals supplied to said signal processing means.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA311,387A CA1065599A (en) | 1974-12-02 | 1978-09-15 | Flux valve heading repeater compensation systems |
Applications Claiming Priority (5)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US05/528,760 US3939572A (en) | 1974-12-02 | 1974-12-02 | Latitude compensator for flux valve heading repeater system |
| US05/528,759 US3942257A (en) | 1974-12-02 | 1974-12-02 | Index error correction for flux valve heading repeater system |
| US05/528,758 US3938257A (en) | 1974-12-02 | 1974-12-02 | Two-cycle compensator for flux valve heading repeater system |
| CA238,149A CA1059311A (en) | 1974-12-02 | 1975-10-22 | Flux valve heading repeater compensation systems |
| CA311,387A CA1065599A (en) | 1974-12-02 | 1978-09-15 | Flux valve heading repeater compensation systems |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1065599A true CA1065599A (en) | 1979-11-06 |
Family
ID=27508047
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA311,387A Expired CA1065599A (en) | 1974-12-02 | 1978-09-15 | Flux valve heading repeater compensation systems |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA1065599A (en) |
-
1978
- 1978-09-15 CA CA311,387A patent/CA1065599A/en not_active Expired
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