JPS6063479A - Pulse compression device of radar - Google Patents

Abstract

PURPOSE:To select optionally a frequency difference between subpulses independently of intervals of sampling by correcting a phase difference between channels after sampling and performing the pulse compression processing thereafter in a pulse compressor of a radar in the spread spectrum system. CONSTITUTION:Operations of a transmitter 10 and a receiver 30 up to a mixer 62 of a spectrum condensing circuit 60 are similar to conventional those. The output of the mixer 62 passes an intermediate frequency amplifier 33 and a phase detector 34 without adding. A reception video signal outputted from the phase detector 34 is sampled and digitized by a sampling circuit 35. The phase difference between channels is corrected in the output of a phase correcting circuit 70, and this output is subjected to pulse compression by a pulse compressing circuit 36.

Description

DETAILED DESCRIPTION OF THE INVENTION This invention relates to a pulse compression device for a lamp radar.

A conventional pulse compression device is shown in FIG. In FIG. 1, 121 is an antenna, and fl+ is a transmitter/receiver switch. (+11 is a transmission 1d object signal generation circuit that generates a transmission expansion pulse signal 7 as an intermediate frequency band transmission signal;
021 is a local oscillator for the station waveband, (13) a mixer that creates a signal of the sum of the output frequencies of the transmission signal generation circuit (river and local oscillator (121), and (14111'i mixer (13)
It is a power amplifier that fully amplifies the output of 1, and is a transmitting device consisting of (10) lights and the above-mentioned Wr dispersion elements (14).

011-digit high-frequency amplifier, @ mixer that generates a signal of the difference in the output frequency of the high-frequency amplifier on and the local oscillator group, (G) is an intermediate frequency amplifier, ■ is a phase detector, 01
Gezanfling circuit, one-lamp pulse Fhi contraction circuit,
- is a receiving device consisting of the above circuit element Cn-(2). Further, (3) is an output terminal of this device.

In addition, there is a group of local oscillators with different frequencies L1 to
Lwi output multiple (N1 local oscillators (41-1
) to (41-IJ), and a dividing circuit 6 that divides the stretched wave pulse emitted from the transmission signal generating circuit (output) into a plurality of (N) sub-pulses. , a mixer which mixes this sub-pulse line h divided into a plurality of parts with the output of the local oscillator group @(1 line) to obtain sub-pulse groups of seven different frequencies f1 to fN. (52-1) to (52-N), and an adder Q that adds the outputs of these.

In addition, (field is a spectrum condensation circuit, distribution circuit +6
11, mixers (a2-1) to (62-IJ), and an adder circuit.

Next, the operation will be explained.

Here, for convenience of theory 9J, the above local oscillator (g).

The mixer on the spectrum spreading surface and the mixer on the spectrum condensing circuit are 6, and the number of subpulses obtained by each of them is 5 (N-5).

The transmission 1g signal generation circuit (Ill shown in FIGS. 2(a) and 2(b)) has a pulse width T, a center wave ridge f, (f, and an intermediate frequency).

A frequency modulated expanded pulse signal in an intermediate frequency band with a frequency sweep width Δf is generated. As is well known, the frequency spectrum of this signal has a spread of approximately Δf, as shown in FIG. 2(d).

Such a transmission signal generation circuit (output signal of the river (Fig. 3)
a)) is changed to N as shown in FIG.
(-5) Divide into equal parts and output separately. These outputs σ are added to the mixers (52-1) to (52-5), respectively, and the local oscillation signals (frequency LX to L
5), and the sum frequency signals fl, fQ, ......fδ are mixed, for example, as shown in FIG.
Obtain the Zabu pulse output of 1. By adding these in the adder circuit, a spread spectrum expanded pulse signal as shown in FIG. 3(b) is obtained. here,
Frequency interval between adjacent local oscillation signals (e.g. La - Lx
)Thfs, its frequency spectrum will be as shown in FIG. 4, and the spectral distribution range will be approximately 4fq.

For example, if Zl is selected as fB -10 MHz, the spectral distribution range will be approximately 40 MHz, and the conventional IMH2 (
It becomes possible to widen the band by 40 times (assuming that the pulse width after compression is one voice S).

The output of the spread spectrum circuit thus obtained and the local oscillation signal from the local oscillator α2 are mixed in a mixer Oj and converted into a high frequency signal of the sum frequency. This output is amplified by the power amplifier θ, becomes a high frequency transmission signal, and is sent to the antenna (2) via the transmitter/receiver switcher i1i k.
is radiated into the air. On the other hand, a reflected signal from a target object (not shown) is received by the antenna (2) and guided to the receiving device (7) through the transmitter/receiver switch (1). This high frequency received signal is amplified with low noise by a high frequency amplifier 0), mixed with the local oscillation signal from the local oscillator 021 mentioned above in a mixer, and converted into an intermediate frequency (difference frequency) received signal.

The spread spectrum expanded pulse signal inputted to the spectrum condensing port cttn is spectrum condensed by the reverse process as in the case of the spread spectrum circuit, and a reception expanded pulse signal appears at the output of the adder circuit. In other words, the distribution circuit 161 ) output signal (Fig. 3(g)) is sent to the mixer (62
−1) to (62-5), and the frequency L
It is converted into a difference frequency signal with the local oscillation signal of 1 to L5,
After that, they are added together at addition port g&- as shown in Fig. 3(a).
The received expanded pulse signal is shown as K.

The received expanded pulse signal as an intermediate frequency signal obtained in this way is amplified by an intermediate frequency amplifier, and is then amplified by an example of a phase detector at the transmission signal generation point w! The phase is detected by the reference signal sent from I (Ill), and the sampling circuit ■ performs sampling and digital conversion at a timing that is an integer multiple of the reciprocal of the carrier frequency difference between each sub-pulse, and the pulse compression port V & (to) performs pulse compression. The pulse compression circuit (to) is configured as a digital pulse compression circuit using, for example, FFT (Fast Fourier Transform), a delay circuit, and an addition B42, and outputs the signals shown in FIGS. 2(a) and (b). It has all the compression functions as shown in (c) of the same figure.At this time, the pulse width τ after compression is approximately the reciprocal of the frequency sweep width Δf 1/Af
will be equal to . The pulse signal output after compression from the pulse compression port @ tile is taken out from the terminal (3) and sent to a radar signal processing device (not shown) IC.

When performing one transmission using the above method, the transmitted signal is as follows.

That is, the output of the transmission 1g signal generation circuit 1111 is EC=cos
(ω0+Δωt)t ('o≦t<T)Δf (However, ωow2π(fo −-), Δω−2πbt.

2 T Local oscillator group θa ratio output Fir, 1-cos(ωL-2°s)tFiLa-co
g(ωL-"s) is a hole L!l"e08ω1t KL4-cos(ωL+ω5)t KL5-cos(ωL+2ω5)t (However, ωL-2πL3.ω9-2πfs) Spread spectrum circuit ■Output is 331-008(ω0+ △ωt)t-cos(ωL-2
ω5)t(0≦tku−) T 2T (i≦(i) FiB5−cos(ω0+Δωt)t−cogωLtl
1 −−coo(ω0+Δωt+ω1)t+−cos ・−
-22 2T 3T (-≦tku-) 5 5 EB <-CO8(ω0+Δωt)t-cos(ωL+
ω5it3T 4T (-≦t<-) 5 5 g55"Coθ(ωO+Δωt)t-cos(ωL+2
ωSatT (-≦t<T) The local oscillator α21 output is KL-C08ωLxt Therefore, the mixer (1 turbidity output (transmission output signal) is E71 = (
! 01i1ω1llcos (ωO+Δω is 1ωL−2ω
s)t(O≦t〈-) EccosωLlt”(!Oθ(ω0+Δωt+″L
-ω5)tBT3-CosωLlt-C06(ω0+Δ
ωt+ω1)tBT4-C08ωLl'c'C05(ω
0+Δωt+ωL+ωB)tET51+lIC0SωL
lt”COθ(ωθ+Δωt+ωL+2ωs)tAlso,
If the transmission-reception time is itl, the received video signal will be as follows.

That is, the local oscillator Qz output (IcL-C08ωLx ft-tR) mixer) output is EIFI"Co dayωLl (t-"R)・C08(ω1
．． l+ω0+△ωt+ωL-2ωs)tEIF2-c
osωLILt-t4)-cos(ωL1+ω0+Δ
ωt+ωL-ωB)tEIFB-C08ωLx(t-t
R)-coo(ωL1+ωO+△ωt+ωL)tEIF
4-Cot; ωLl (t-tR) fist cos (ωL1+
ω0+Δωt+ωL+ωq)tEIF5−CoθωLl
(t-1R)・008(ωLl+ω0+Δω is 10ω work 5
+2ωB)tHere, -ωr, 1t, and H'd are ignored because they have the same phase in each channel.

The local oscillator group output is EL, -C8(ωL-2ω5)(t-t)L)F! L2
-cos(ωl-ω3)(t-tR)EL3"QOB6
)L(t-tH) EC4=CO8(ωL+ω5)(t-tl+)EL5-
coe(ωL+2ω5)(t-tR) spectrum condensation circuit...Output is EC1-cos(ωL-2ω5)(t″″tR)・C0
8(ω0+Δωt+ωL-2ω5)tE(1-cos(
ωL−ω5)(t−tR) groan cos(ω0+Δωt+ω
L-ω5ltT 2T (-≦tku-) 5 5 IC! l”C0IIIωL(t-tR) ・C08(ω
O+△rut+ω1. )tl 1 --Coil(ωL t R+ (ω0+Δω1)1)
+−・−・・−・2 2 2T 3T (i≦t EC4=CO8(ωL+ω8)(t−tR)・cos(
ω0+△ωt+ωL+ω5)t3T 4T (-≦tku-) 5 5 Ec 5-cos (ωL+2ωs Ht-tR)・c
os(ω0+△ω is 1ω1+3ωBltT (-≦t<T1) Here, if ωLtRu is ignored because each channel has the same phase, the spectrum condensation circuit (- and intermediate frequency amplifier ■
The output of is Ecx=cos(-2ωStR+(ω0+△ωt)t)
(0≦tku-)T 2T EC2=COB(-ωst, +(ω0+Δω1)1)
(-≦tku-)5 5 The reference signal sent to phase detection b (to) is ER-008 (
ω0(t-tR)) Phase detector - output (1 Kv 1-coB(-2ωstR+(ω0+Δωt)t
) ・cos(Co(t-tu))(0≦t<-) Fiv2=cos(-ωstR+(ω0+△ω1)1)
・cos(Co(t-ti))Ev3-cos(ω0+
△ωt)t-cos(ω.(t-tR))Kv<”c
os(ωB t R+ (ωO+Δωt )t) ・c
og(ω, (t-tH)) Snow-C08(ωBtB+
Δωt2+ω0tR)+−・・−・・・・2 3T 4T (−≦tku−) 5 b Bv5=cos(2ωB t R+ (ω0 Plant ωt) t
)-coe(Co(t, tH))=-coe(2ωs
tR+△ωt''+ωOtB ) + ==-・2 (-≦t<T) ωotRUEach channel has the same phase, so if ignored, the received video signal will be Ev 1=cO8(-2ω5tlli+Δωt2) (
0≦t<-)T 2T Ev2-coe(-ωStR+Δωt2) (-≦tku-)5 5 2T"3T Evs-coe(Δωt2) (->t<-) 5 T E'V5-Coθ(2ωStR+Δωt2) (−≦t
<TI, and therefore, as shown in FIG. 5, the phase of each channel changes as the transmission-reception time tR changes.

Here, by setting the time tsk (m: integer, m-th sampling from the transmission time) at which the received video signal is sampled in the sampling circuit (2), the sampling output is -cos (-
2m-n・2π+Δωt2)-cos(Δωt2)
(○≦tku−) T 2T (−≦tku−) 5 ED4=008(ω5−m−n6.−, +Δωt2)=
cos (△ωt 2 ) (-≦tku-) 5 5 ED5-coθ(2ωs 'm'n--+△ωt2)=
cos (Δωt 2 ) 3 (-≦t<T) That is, as shown in FIG. 5, the transmission 15 to reception time tR
When is an integer multiple of the inverse number 17f9 of the frequency difference f8 between sub-pulses, the phase of each channel becomes 0, and the phase difference between each channel becomes 0. Therefore, by sampling the received signal at this time, the pulse pressure Nk can be done.

At this time, it is necessary that the following relationship VC exists between the sampling interval l\ts and the frequency difference fs between each subpulse.

ΔtB-n・-n=integer 3 As an example, sampling interval ΔtB p 1. IIs
Then, the frequency difference between each sub-pulse fS U IMH
It is necessary to choose an integer multiple of 2.

Since the conventional spread spectrum pulse compression device has the VC configuration as described above, it is necessary that there is a relationship between the sampling interval Δts and the carrier frequency difference between each sub-pulse, tsTry, and the carrier frequency difference between each sub-pulse. −
+9 had the disadvantage of being limited by the sampling interval ΔtsiC.

What are the drawbacks of this invention compared to the conventional ones mentioned above? This was done in order to eliminate this difference.In a spread spectrum radar pulse compression device, the phase difference of each channel 1 after spectrum condensation processing is determined by the carrier frequency difference between each subpulse and the transmission to reception time. , after sampling, the phase between channels 51. After the correction, pulse compression processing is performed, since the frequency difference between each sub-pulse and the sampling interval are not subject to control and can be freely selected.

Pulse compression device for spread spectrum radar? The purpose is to provide.

Hereinafter, one embodiment of the present invention will be explained with reference to the drawings.
In the figure, the river is a spectrum condensation circuit, which is composed of a distribution circuit (6B, mixers (62-1) to ((+2-N)). (33-1) to (33-N) ri intermediate frequency Amplifier, (3-1) to (34-N) are phase detectors, f
/l phase correction circuit, ('7l-1) to (71-
N) is a multiplier, and ('72-11 to (72-N) are correction value generating circuits.Other than this, the structure is the same as the conventional structure shown in FIG. 1).

Next, the operation will be explained.

49 works of the transmitting device (lO) in the present invention I:t i;
I: Same as US, and the mixer (62-1) of the spectrum condensing circuit IBIl in the receiver +* m.
The operation up to (62-N)' is the same as the conventional one.

In the present invention, the outputs of the mixers (62-1) to (62-Nl) are not added as in the conventional case, but instead are added to N intermediate frequency amplifiers (33-1) to (33-N), phase detectors ( 34-1
)~(34-N) 2 passes, but they are represented in the formula in the same way as before. That is, if the sub-pulse length N=5 as in the conventional case, the received video signal output from the phase detectors (34-1) to (34-5) has T2T Ev2-cos (-ωBJ+△GJt2) <-S
tku-)5 2T, 3T F2V5-cos(△ωt”) (-St< )5
5 3T 4T Eve-cos(ωstR+Δωt2) (-St<-
)5 5 T Ev5-cos(2ωStR+Δωt2) (-St(
T) This signal is sent to the Sangrinob circuit (35-1) to (35
-5), and if the time from transmission to sun 19 is kts, then Rnx-cos(-2ωst, 3+Δωt2) (o≦
t<-) shame”-ooS (-(us tB+am1;2
1 T 2T (-≦0ku-) 5 5 2T 3'l' EDsmcos (Δωt"(Go<-)5 3T, 4T KD4=cos (ωst6+△ωt2) (-≦oku-)
5T! D5=coa(2ωsts+△ωt2)(-≦0AT
) The correction values generated by the correction value generation circuits (72-1) to (72-5) are expressed as: 0x-cos (j2ω5ts) 02-cos (ω5ts) 3-1 C4-cos (-ω5ts) C5-co8(- 2ω5t8), the output of the phase correction circuit (to) is Eol-coo
(2ωsts) ・cos (-2ωsts+Δωt
2) Fog-cos(ω6tB)-cos(-ωsts
+Δω1. II) Eo E C08(-ω5ts)-co
s(ωsts+muωtQ)Eos−cos(−2ω5t
s)·cos(2ωsts+Δωt2), the phase difference between each channel is corrected, and pulse compression can be performed by the pulse compression circuit (to).

Since the carrier frequency difference between each sub-pulse is known, the correction value f ROM (ReadOnl
A correction value can be generated by storing the correction value in a file such as y Meory) and reading it out according to the transmission-to-sun 19-fufu time tS. Since the correction value is repeated every reciprocal of the carrier frequency difference between the transmission sub-pulses, it is not necessary to store the correction value for the entire mouvc + "i range (radar repetition period). In the above embodiment, the chirve pulse Although the example of compression has been described, it is also applicable to other methods, such as frequency coded pulse compression.Also, subpulse division does not need to be equal division.The frequency difference f9 between subpulses needs to be constant. Although the spectrum spreading and condensing processes have been described in the case of performing them at intermediate frequencies, they can be similarly overused at the high frequency stage.

Correction value generation port vIrC:ra fl Although explanation 7 was given on the method of reading out the correction value stored in the ROM according to the transmission to sample time, the difference between the carrier frequency fS and the transmission to sample time from 3 to c-cos(2π
It is also possible to generate a correction value sound by calculating fsts). In the Nuvector condensing circuit, mixers (62-1) to (
62-N) without adding the outputs of N intermediate frequency amplifiers (
33-1) to (33-N), phase detector (34-1)
~(34-N), sampling times @ (35-1) ~(
35-N)? As explained using an example, the outputs of the mixers (62-x) to (62-N) are added by full adders with the same configuration as the conventional one, and one system of times I#! After passing through the filter in the pulse compression circuit (7), the pulse can be re-divided into sub-pulses and subjected to phase correction (7).

As described above, according to the present invention, in a spread spectrum radar pulse compression device, the carrier frequency difference between each sub-pulse can be arbitrarily selected regardless of the sampling interval.

[Brief explanation of drawings]

Figure 1 is a configuration diagram showing an example of a pulse compression device for a conventional spread spectrum radar, and Figure 2 (al and b) are waveform diagrams of expanded pulse signals.
/< Signal waveform diagram after pulse compression, Figure 2 (a) is a diagram showing the spectrum of the expanded pulse signal, Figure 3 (a) is a waveform diagram of the Vi transmission and reception expanded pulse signal, Figure 3 (bl~( f) is an explanatory diagram of spectrum spreading, Figure 3 (ml is a waveform diagram of a pulse signal after spectrum spreading, Figure 4 is a diagram showing the entire spectrum of a spread spectrum expanded pulse signal, and Figure 5 is a diagram showing the entire spectrum of a spread spectrum expanded pulse signal). A diagram showing the phase of the mixer Q output, and FIG. 6 is a diagram showing an embodiment of the radar pulse compression device according to the present invention. In the figure, (10) is a transmitting device, (Ill is a transmitting signal generation circuit, 031 is a mixer, (+4)
is a power amplifier, Ill is a transmitter/receiver switch, (2) is an antenna,
) is the receiving device, C11) is the high frequency field @ mixer, @ge mixer, the trowel is the spectrum condensing circuit, the name is the intermediate frequency amplifier, C3
4 is a phase detector, (to) a sampling circuit, □□□ a pulse compression circuit, and (to) a phase correction circuit. The same reference numerals in each figure indicate all the same parts. Agent Masuo Oiwa Figure 2 Figure 3 Figure 4 Figure 5 (α) Procedural amendment (method) Commissioner of the Japan Patent Office 1, Indication of case Patent Application No. 171580/1983 3, Person making the amendment-t' :・-7・ ＼＼日に 6、Date of amendment order January 81, 1982 6、Object of amendment (1) Brief explanation of the drawings in the specification 7、Contents of amendment (1) Specification From the 19th line on page 22 to the 5th line on page 28, the text ``Figures 2 (a) and (b) are pulse signal waveform diagrams.'' The figure is an explanatory diagram of the waveforms of the transmitting device Ql in FIG. 1.''that's all

Claims (1)

[Claims] f+i The internally modulated transmission expanded pulse signal is divided into multiple subpulses (C, the carrier frequency of each subpulse is converted to a different frequency, and then the 7 layers are added together again to create a spread spectrum expanded pulse that has multiple channels) Spread spectrum circuit for transmitting signal sound and spread spectrum expansion of the received signal) A spectrum condensing circuit that generates a received expanded pulse signal, a phase correction circuit that detects and samples the received expanded pulse signal, and corrects a carrier frequency difference between channels during transmission.
The above phase correction signal? Using pulse compression? What is claimed is: 1. A radar pulse compression device characterized by comprising a pulse compression circuit for performing