JPS60221998A - Power generator - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

[Detailed description of the invention]

Relationship to Related Applications This invention is disclosed in pending U.S. patent application Ser. No. 564,622, filed on the same date as this application.
No. 03, No. 564,582, No. 564,550, and No. 564,621. TECHNICAL FIELD This invention relates generally to x-ray generators and more particularly to voltage feedback circuits for controlling the output voltage across an x-ray tube. When generating and utilizing x-rays, it is common to select specific voltage and current levels to suit the particular application or procedure at hand. For example, in the field of medical X-ray imaging, typical voltage levels used in common copper ray photography may be in the range of 50 to 150 kV; It is more often in the +20 kV range, and for X-rays used in mammography, 24 to 50 kV.
It is more likely to be in the kV range. Similarly, the level of applied current varies by 1250 mΔ for fluorography, consisting of 0.1 mA for fluorography. Traditionally, these voltage and current levels have been controlled by a circuit design that allows the A operator to set the desired kVp and A. Due to equipment variations that may occur during exposure, such as changes in load, line voltage, or filament temperature, kVrl and I
It was not possible to maintain the value of IIA precisely within the desired range. Traditionally, X-ray generator manufacturers have anticipated possible changes and designed circuitry to compensate for these variations in a manner that takes less than 1 minute to keep kVp and IIIA within planned tolerances. Recent developments have been progressing along the lines of closed-loop feedback schemes that overcome the drawbacks of open-loop schemes mentioned above. One method is to use closed-loop return;
This is to control mΔ of the raw equipment. Such a device is described in pending US Patent Application Serial No. 375,088. In the field of kVp control, there are satisfactory methods of sensing the output voltage and using the feedback signal to directly modulate the output voltage in a fast and effective response that maintains a predetermined voltage level. A closed loop method has not been developed. The conventional method of maintaining a nearly constant voltage level over 1q fluctuations occurring in the line is to use a so-called pol pack, a variable input/output transformer driven by a motor to obtain a variable output. . The main disadvantage of bolt packs is that they are relatively delicate in operation. That is, the bolt pack has a response time of about 1 second. For this reason, the Pol 1 Pack Control+ device is only used to set the correct voltage at the start of the exposure and for long exposures (fluoroscopic exposures).
The removal of the floor space will not be adjusted afterwards. In comparison, the desired response time for an x-ray generating device is in the millisecond range, allowing short, well-defined power pulses to be provided for a variety of procedures and applications. For example, a high voltage pulse with a very fast rise time, i.e., short, such as 1 ms, a short flat peak for exposure, such as 1 ms, and a fast fall time, is produced. or desirable. Therefore, correction needs to be made within less than 1 millisecond. It is known to use an inverter in an X-ray generation circuit to supply alternating current to the primary side of a high voltage transformer. However, transistors have generally not been used for this purpose, primarily because they are relatively difficult to control. Rather, it was the iristor that was used as a switching element for this purpose. Although Zyristors are generally considered robust and relatively easy to control, they have an inherent drawback in that they require the use of forced commutation circuits. Because of this, not only are extra components required, but each additional electrostatic set tends to substantially slow down the response time of the circuit. For example, when using a relay inverter, it is difficult to obtain short high voltage pulses in the 1 millisecond range while maintaining a reasonable level of repeatability. When controlling the AC output from an inverter, there are many possible ways to control the supply of DC voltage to the inverter. Some of these include phase-controlled rectifiers, transistor series or parallel regulators, and semiconductor switching DC voltage controllers. Among these, semiconductor switching elements are usually called "choppa", and can control DC voltage more efficiently and with faster response than other methods. However, in a DC circuit, a considerable amount of single-wave action is required.
response time is much slower when operating from a fully closed-loop, voltage-regulated inverter power supply. Such indirect methods introduce more circuit losses due to forced commutation circuits that must be used to handle the large voltage and current fluctuations required during X-ray generator operation. It happens. Furthermore, in such a configuration, the power sent out from the inverter is 1
It will be appreciated that the process is carried out twice, once by the DC voltage controller and once by the inverter. In addition to the inherent variations in sources and loads, accidental and unplanned conditions, such as arcing in an x-ray tube, can occur on the high pressure side and, if not controlled, can lead to component damage. Furthermore, as with any control circuit, there is a risk of malfunction or failure in the low voltage control circuit, which, if not detected and handled, may cause undesirable consequences on the output of the control circuit or within the control circuit itself.
Sometimes. For this reason, no matter what control or performance features are added to a conventional device, associated monitoring and adjustment functions must be provided to effectuate such improvements. In the field of X-ray generators, there has been a reluctance to make any significant changes to the conventional system. As desired, conditions typical of X-ray applications (i.e., variable load in the range 0.1 to 1250 nlA, variable box-to-box voltage from 24 to 150 kVp, and mΔ as low as 0.25) Therefore, it was difficult to create a suitable device like this.The ripple control is good, the reproducibility is high, the linearity is good, the rise time is fast, and the shape of the power waveform is good. The task is made even more difficult due to various performance requirements such as low temperature control, steady state conditions, low exposure times, and short fall times. When excessive voltage occurs in the device, excessive current may also occur in the device.
This may cause damage to the xm tube itself, or damage to high or low pressure components within the device. When using an inverter in an XS generator, it is desirable to reverse the current through the switching device as much as possible. However, in order to maintain a reasonable lifespan for such switching devices, it is imperative that the current 1 Novel does not exceed the predetermined level. The ability to determine when a current level is approaching its limit and take necessary measures to prevent this from happening is difficult to achieve. This is especially true when the inverter operates at higher frequencies. OBJECTS OF THE INVENTION Therefore, the main object of this invention is to provide x! An object of the present invention is to provide an improved X-ray generator having means for controlling the output power applied to the m tube. Another object of the invention is to provide an x-ray generator with a closed-loop voltage feedback loop that responds quickly and accurately to input voltage and load variations to maintain a desired voltage peak. be. Another object of the invention is to generate voltage pulses with fast rise times followed by relatively short exposure times of substantially constant voltage, followed by relatively fast fall times over a wide range of operating conditions. The goal is to provide an X-ray generator with a control circuit that is sufficiently fast and responsive. Another object of the invention is to reduce the current level by causing damage to components of the device.
The aim is to operate the x-ray generator at relatively high current levels while at the same time ensuring that B does not exceed certain safety limits of <1tl. The above and other objects, features and advantages will be better understood from the following drawings. Summary of the Invention Briefly stated, one aspect of the invention provides an X-ray generator that
It has a high voltage feedback from the x-ray tube and has a control circuit responsive to this feedback to control the operation of the system's inverter in a manner that maintains a predetermined voltage level to the x-ray tube. A feedback signal is thus applied to directly control the output of the inverter, allowing the device to respond quickly and accurately to variations in line voltage and load. Cooperative components such as the high voltage transformer, high voltage output U1 waveform, and high voltage shunt circuit are designed to match and contribute to the fast response characteristics of the device. The resulting control circuit has a short rise time of 1 ms, a short steady-state high voltage period of 1 ms with negligible ripple,
In particular, the device becomes capable of supplying a high voltage output to an X-ray tube with a fast fall time for very low IIIA such as 0.25 mA. Another aspect of the invention is that relatively high frequencies (i.e.
A transistor inverter operating within a few kHz) is adapted to generate a square wave pulse width modulated output. The rectified output voltage level is controlled by selectively varying the frequency as well as the mark/space ratio of the output waveform. The inverter is controlled in response to signals generated to represent the setting of the A verator, the feedback of the output voltage, and the operating conditions of the device. Another aspect of the invention is to sense the saturation condition of the transformer core and, in response, initiate corrections 1ii[r+fl to alleviate the problem. By sensing the current of the transformer,
Means is provided for integrating the resulting signal to provide an indication that the core is approaching saturation. That connection,
This signal is applied to a sawtooth generator to generate a control signal. This control signal is applied in such a way as to eliminate the saturation state.
It acts to selectively unbalance the two diagonal current flows. In the voltage feedback loop, a phase advance circuit is installed to dynamically change the gain of the C1 device so that a high gain is obtained during the first stage to reduce the rise time, and after that, the gain is A-Bashu 1 of kV at the end of the rise time by decreasing the
- Clamp. In order to reduce this effect by 1q, the voltage feedback signal is applied to the phase advancing circuit before being applied to the voltage request signal at the input of the amplifier. The random noise introduced by the phase advance circuit is installed in the feedback loop of the amplifier.

[This is reduced by the connected phase-to-phase circuit. Set up a high voltage voltage divider circuit: take out the low voltage control signal representing the output voltage for use in the control circuit. Instead of using a separate capacitor in the high voltage section of the 1° voltage divider, use a -wave capacitor for that purpose, This serves a dual purpose. This results in a substantial reduction in the number of components and results in a closed-loop voltage feedback signal with good transient response when used with the high frequency pulse width modulated output of the present invention. Possible malfunctions, voltage spikes, and flashovers In order to protect the equipment from undesirable conditions that may occur due to Certain conditions for which the protective device will reduce, prevent or stop operation of the device are excessive voltage at the output, unbalance of the anode and cathode with respect to earth, excessive current flow, Uncontrolled demand for excessive kilovolts.Start timing the X-ray exposure only after the output voltage level reaches 75% of the required value or device set point, thus ensuring improved performance. At the same time, the necessary adjustment conditions are fulfilled.
Preferred embodiments of this invention are shown in the drawings and will be described below, but it should be understood that various modifications may be made within the scope of this invention. DESCRIPTION OF THE PREFERRED EMBODIMENTS A typical prior art XS* generator is shown in FIG.
Connected to 3. The taps of the autotransformer 12 can be selectively changed relative to the primary coil 16 to change the connection of the primary to the input line, thus compensating for changing line conditions. . The electric transformer 13 is typically Yi! primary side 17 of im and secondary winding 1 of Δ-Y connection
8 is in type a and generates 12 or 6 output waveforms. After this, the output is the double wave rectifier bridge 19
．． 21, from which a high voltage is supplied to the X-ray tube 22. The power level to the x-ray tube 22 is varied by a variable input/output transformer 13. Its primary winding 17 is selectively closed by a static contactor 23, usually an SCR. Such conventional devices suffer from the various drawbacks mentioned above. The xm generator system of the present invention is shown in FIG.
It consists of a C filter 26 and a DC to AC pulse width modulated inverter 27 operating at variable high frequency conditions, ie in the range of several kilohertz. The output of inverter 27 is controlled by pulse width modulation by means of varying both the mark-space ratio and the frequency by a kVIIl'l remote controller, as will be explained in more detail below. The output of the PWM inverter 27 is sent to the high voltage transformer N28, and finally x
It is applied to the wA tube 31. The X-ray tube 31 operates at voltage levels up to 150 mA, but depending on the particular application and procedure, can have any load from 0.1 mA to 1250 mA to accommodate a variety of line photography applications. For this purpose, it must be possible to have a wide range of exposure times from 1 millisecond to several seconds. As will be explained in more detail below, the present invention allows such a wide range of operating conditions and performance parameters to be measured by rapidly increasing the X-ray output in a positive 111'' manner. Looking at Figure 2, it can be seen that the main feature of this fast-response device is the closed-loop feedback control device. A signal representing WM is sent to the high voltage feedback control drawing 33, and this feedback control drawing is one
- (Generates a control signal for the inverter 27. Although the power supply has been described as being 3-phase human power, it should be noted that single-phase human power may also be used.
Because it is designed to operate at a higher frequency than a conventional X-ray generator, the problem of waveform ripple is greatly reduced.For this reason, even when conventional X-ray generators are not capable of single-phase operation. , when the features of the present invention are incorporated and used, it can be practically achieved. A central control microprocessor 30, a mixing amplifier and feedback controller 34, an I/I tooth wave generator and comparator 36, a logic control unit 37, It has a dedicated microcomputer 41 that controls the power 1 transistor controller 38, the square wave pulse width modulation 1 to the transistor inverter 27, and the interlock of the safety bond from the power 1 transistor inverter 27 to the logic control unit 37. Inverter monitor 40, high voltage transformer 28, high voltage rectifier 29, high voltage shunt 32, high voltage divider feedback circuit 46, error signal and sawtooth generator and comparator 36 acting together to prevent saturation circuit 47, current limit circuit 48 , display console and A operator control device 49 (microphone [1 processor 1)]
) and an image forming device 51, which may be of a conventional type. The entire apparatus will now be described generally with reference to FIG. 3, and the individual parts will be explained in more detail thereafter. The operation of the entire control device is managed by a microcomputer 41 and a control microprocessor 30. The inverter microcomputer 41 continuously monitors the high power 1-transistor inverter 27 and inspects it 1-
A centrally controlled marquee, dedicated to
A processor 30 acts to control the required value before and during exposure. The control microprocessor 1) 30 also reads the kiloformyl number coming from the return device and precisely controls what is happening on the high pressure side during exposure. Any of the numerous microbroths and/or converters available on the market can be used in this invention. For example, an Intel 2+ 8085 type microcomputer 01 can be used for central control, and an Intel 8749 type microcomputer can be used to monitor the operation of the inverter. In response to signals from display console 49, central microprocessor 30 generates a kilovolt demand value and sends this signal via A/A converter 52 to mixing amplifier and feedback controller 34. The kilovolt number confirmation signal is Δ/1. )
It is sent via converter 53 to central microprocessor 30 . In addition to being monitored by the central microprocessor 30 for that purpose, the kilovolt request signal and the kilovolt confirmation signal must be kept in close proximity to each other. It is also used as an input for protective action in the event of an occurrence or damage to the component. In this case, the kilovolt request signal is followed by the key [
' volts confirmation signal does not follow and therefore,
The central microprocessor tree 30 stops the operation of the device. A central microprocessor 30 is connected via data link 54 to display console 49, via line 56. Ru. The operator enters the exposure time and other parameters at the console 49, from which data processing and communication begins. The microprocessor of a device such as Intel's 8088 type microprocessor, and the arithmetic processing such as Intel's 8087 type that handles all C operations that weigh on x#lA (! protection and exposure parameters). These parameters are analyzed and controlled by the device 1, console 49
and between 1 and 33 or type 8088 and 808
The communication between type 5 microblossom and no is
Via link 54, this is done by two data link protocol controllers 59,61. In one embodiment,
These controllers are Intel 8273 chips. These controllers ensure very high reliability in data transmission in both directions due to the N RZ I protection scheme with cyclic redundancy check words. In addition, the console 49 can also communicate with a further data link 62 overlapping the imaging device 5) 1, so that the communication is completely digital and provides high reliability during operation. Communication between the central microprocessor 30 and the impark microcombicoater 41 on the side 1~
This is done in both directions via lines 56,57. For this purpose, the operator's control status, kilovoltage and exposure time are supplied to the central control microprocessor 30 and from there to the inverter microcombicoater 41.
This micro combicoater controls the output voltage, inverter operation and exposure time during the exposure, so that the
Two microprocessors control the exposure time (i.e., a central control microphone [1 controller 8088 and an inverter microcombicoater 41 on the cabinet side, and a microprocessor 8088 in the display console on the console side as supporting equipment). type). This combination injury
Provides redundant protection against excessive radiation doses during exposure. One advantage of using closed-loop feedback and kilovoltage request and kilovoltage confirmation signals as shown is that in long-term exposures, such as in fluoroscopy operations, high-voltage feedback devices or any associated electronic If a component deviation occurs, closed-loop feedback automatically compensates for it. Furthermore,
Communication via line 5 (i, 57) between the inverter microcomputer 41 and the cabinet side microprocessor 30 is such that the central control microprocessor 30 sends the kilovoltage, exposure start command and exposure time command to the microcomputer 41. The microprocessor 41 continuously monitors the output status and sends a status signal and an acknowledgment signal back to the central microprocessor 30. A simple communication link, with several redundancy levels, can detect any potential problem in the power circuits and shut down the high power inverter within a few microseconds, or safely if necessary. It becomes possible to circuit the contactor 63. A mixing amplifier and feedback controller 34 generates a narrow signal that is the difference between the kilopol number request signal and the kilopol number confirmation signal or feedback signal. .The resulting k
The V error signal is amplified and processed in lead and lag circuits, which will be explained in more detail below, to stabilize the device. The kV error signal is sent to the copper wave generator and comparator 36 along with the signal from the anti-saturation circuit 41. The signal from the saturation prevention circuit 47 changes the slope of the sawtooth wave optical q-mirror aI! As will be explained in detail later, the high voltage transformer 28 is prevented from reaching saturation. The kV error signal is provided to a sawtooth curved wave generator and comparator 36 to generate a PWM pulse train with a variable mark/space ratio that controls the output voltage and through a dead loop kilovolt feedback operation. Applied to automatically adjust the output voltage. Copper wave generator and comparator 36 and logic control IT device 37
is controlled by the reset I signal or synchronization signal that is input every half cycle from the dedicated microprocessor +141 via line 64, and the kV error signal from the mixer 34 generates a copper peak wave once every half cycle. intersects the waveform of the vessel,
Avoids the habit of having multiple cross-overs which can lead to problems in the power stage circuits. Logic controller 37, which handles all protection and timing for the device, provides an output on optical #&H line 66, which output is routed through power transistor controller 38 (controls power transistor inverter 27). 3. Logic control device 31
also processes the output of current limit circuit 48. Current limit circuit 48 is responsive to the inverter current level sensed by current transformer 67 in series with the primary of high voltage transformer 28 . The sensed current level is connected to the current limit circuit 48.
If an overload condition occurs in the circuit, the output of the current limit W circuit 48 is compared with the planned safety level.
is applied to dynamically block and override the mark-space ratio. The output from current transformer 67 is also fed back to anti-saturation circuit 47, the output of which is applied via line 74 to a C sawtooth generator to electronically compensate for transformer saturation. Dynamically change the slope. In addition to generating signals that directly control power transistor inverter 27, power transistor controller 38 also generates signals that directly control power transistor inverter 27.
Indicates the power supply status and transistor status of the controller.1
The signal is fed back via line 68 to the inverter microcombicoater 41 which uses this information to control the logic controller 37 so that if any transistor or power supply fails, The information coming from the controller 38 is fed back to the logic controller 37 in real time, and the logic controller 31 first stops the inverter and then
Then, connect the appropriate safety contactor 63 to the circuit. The pulse width modulation type inverter 27 is composed of a plurality of transistors arranged in the form of a double-wave bridge and shown as -1-+-1'a in FIG. A current is passed alternately to the primary side 28 of the transformer via J. The transistors may be used as shown, or they may be used in parallel if the power conditions require it. One type of transistor that has been found useful with this invention is called the Wl'-5752, which was manufactured by Westinghouse Brake (Westcord) in England.
Commercially available from the company. Pulse width modulation is performed by selectively turning ON and OFF only the upper transistors F1 and V2. High voltage transformer 28 is described in pending US Patent Application Serial No. Box 564,612, filed on the same date as this application. In this case, the transformer 28 has very low leakage inbetance so that the rectangular waveform generated by the PWM inverter has very good waveform reproducibility and is sent in pulse form to the secondary side of the transformer. Suffice it to say that it is designed to be as small as possible. This minimizes pulse dropout, minimizes post-rectification ripple, and limits the size of the output waveformer. This easily increases the reproducibility of operation at low mA settings. Rectifier @29 is a normal single-phase type. The high pressure valve sluice or flow divider 32 is connected to the element 69.71.107
Contains unique resistive and capacitive electronic circuitry shown at 108 to improve the response of transformer 28 to dynamic load or source fluctuations or other transient conditions, minimizing rise and fall times. It is as it is. The output of high voltage shunt 32 is a kilovolt output and is provided via line 12 in step-down form to high voltage divider feedback circuit 46 . That is, because the output of the shunt 32 is higher than the voltage that can be applied directly to the control circuitry, the transmission of high voltage transients from the high voltage area that could damage the control circuitry is avoided. In order to reduce the voltage, it is necessary to step down the voltage in several steps using different surge arresters and overvoltage protection methods. This circuit will be explained in detail at the end. The operation of the closed loop kV feedback system relies primarily on the mixing amplifier and feedback controller 34 which generates an error signal equal to the difference between the kilovolt demand signal and the kilovolt output. The controller 34 is configured such that: (1) the error signal is conditioned to the level of the electronic circuit through a high voltage divider feedback circuit so as to generate a mark/space shaped pulse train having a specific ratio depending on the kilovoltage requirement; , (2) Three main variables that may disturb the operation of the equipment during exposure: (a) the fixed free-flow rail, which changes with the track and the adjusting action of the track; and (b) the invitance of the X-ray tube. It is necessary to compensate for the fluctuations that occur in the electronic circuitry, especially those that occur due to long-term exposure such as electronic cooling phenomena, and (c) the A-fret fluctuations that the electronic circuit itself has with respect to the overall device. be. Saturation Prevention Circuit and Sawtooth Generator and Comparator Referring to FIG. 4, a circuit diagram of the combination of saturation prevention circuit 47 (FIG. 3) and sawtooth generator and comparator 36 is shown. As shown in FIG. 3, the sawtooth generator and comparator 36 includes (1) a synchronization signal coming from line 64 to reset the range of the sawtooth every half cycle; (2) saturation prevention. Line 73 from circuit 41
and (3) a kilovolt error signal E from the mixing amplifier 34. Anti-saturation circuit 47 is responsive to the inverter current output received on line 14. Referring to FIG. 4, the inverter is shown to have one transistor T+-T,l and an associated flywheel diode D+-D4. The current in the inverter or the primary side of the transformer 28 is sensed by a current transformer 67 and sent via line 74 to an integrator 76 . The output of this integrator is fed via line 77 to an amplifier 78 whose output is applied to two comparators 79.81. These comparators each have positive and negative reference levels, which are usually very low values, ie, close to 1, and determine what is called the W saturation level of the transformer. The outputs of the comparators 79.810 are connected to the Punto gates G1. G2
It can be seen that the output is applied to Nando Goo 1-03. The output of NAND gate G3 acts to close FET switch E1 for one diagonal T+, l-4 or the other diagonal -1-2, T3 of the power inverter. FET switch F1 becomes 111L;
The error amplifier output signal, which is proportional to the sensed current, is now provided to a precision rectifier 82. This rectifier operates on first and fourth slopes and provides a linear output to a sawtooth generator or compensator via FET switch F1. The sawtooth generator is an integrator 83 with line 64
It is preset by synchronization No. 11 from , and generates a sawtooth wave. The slope of the sawtooth wave is constant at zero saturation level, as shown in FIGS. 5A and 5B, and is limited by the range of a portion of the overall closed loop feedback system. After this, the sawtooth waveform signal is sent to the comparator 8 together with the kV error signal E.
5, which comparator responds by generating a PWM pulse train that controls the inverter. For example, diagonal line T1. When saturation begins to occur in the direction of T'4, error amplifier 78 provides a DC level input to precision rectifier 82, the output of which is linearly proportional to this input. Precision rectifier 82 is responsive to either positive or negative DC voltage input from error amplifier 78. Its sign depends on the direction of the relevant current in the power transistor inverter. As a result, the magnitude of a particular DC saturation level determines the output of precision rectifier 82, and if this saturation level is higher than the preset base f4j level to correct the condition, this output will - Pass through switch F1. For example, if saturation occurs on the diagonal T+, -r4, the sawtooth generator will increase the slope of the portion of the waveform where the diagonal 1-+, T'a is conducting, as seen in Figure 5B. As such, this increase in slope leads to a decrease in the mark/space ratio for a given error signal of the feedback, so that in alternating cycles the mark/space ratio of this diagonal decreases. C1 dynamically compensates for unbalances caused by, for example, differences in hysteresis time of transistors or local saturation of the iron core. If such a saturation condition were to occur in FIG. , the A2 time of the T4 diagonal is equal to +1J, so when the transistor of the T+ , T4 diagonal turns on, the waveform of the sawtooth generator automatically becomes steeper and has a smaller mark/space ratio. Become. The saturation level is thus dynamically and electronically compensated by closed-loop proportional control. That is, the mark period X becomes shorter and the period Y becomes longer. The advantage of this electronic FET compensation is that - the negative current device 67
When the current level of the inverter exceeds a preset reference saturation level, it operates continuously and compensates in a proportionally controlled manner by appropriately reducing the mark/space ratio as the inverter diagonal approaches saturation. It is to be. Logic Control Unit Logic control unit 37, which operates in conjunction with microcomputer 41, will now be described for both its analog and digital functions. The analog +i function is shown in FIG. 6 and the logic signal flow is shown in FIG. The outputs of the signal processing circuits, whether analog or digital, are supplied to a microcomputer 41, which controls the overall inverter operation, protection and performance. In FIG. 6, on the left side, a series of input signals 111 to 16 are shown.
And on the right side, a series of output signals 0+ to 06 are shown. These input and output signals are applied to or derived from protection circuitry within the device and control overall device operation on the high and low sides. First the ground anode and cathode signals II+, +1
2 comes from a voltage divider or flow divider 32, as seen in FIGS. 313 and 8. These signals are sent to operational amplifier 8
4 and the output represents the kVV pressure output. This output is fed back to the main or central microphone processor () 30 in the KV network 1, and as shown in FIG. The output of the operational amplifier 84 is fed to the positive side of the operational amplifier 86, where the output is fed to the positive side of the operational amplifier 86, where it is inputted via the central microcontroller 30 to I)/Δ converter 52 in the electric oyster 17, as shown in FIG. It is compared with the incoming kilopol I-number request signal 12. An operational amplifier 86 calculates the kilopol-term error signal, which is applied to a comparator 85 where it is compared with the output of a sawtooth generator, not shown in FIGS. 4 and 5. and generates output 01. Output 01 is a pulse width modulated pulse train that controls the inverter. FIG. 6 shows the high-voltage side including the transformer 28 (not just the power supply 1 to the protection circuit that protects the transistor inverter 21) and the high-voltage side that normally originates from the high-voltage side of all X-ray equipment.
There are several types of associated control circuits that are typically affected by arcing, flashing or transient conditions. The output 02 of comparator 88 is (1) very fast in operation and (2) provides overvoltage protection that is responsive to small transients that may occur on the high voltage side. Referring to FIG. 6, the charge signal 12 from the cabinet's central processing unit is applied to the positive input of an operational amplifier 89 and added to a reference signal C, which is considered to be the maximum allowable overvoltage, e.g. 10 kV. The output of the operational amplifier 89 is the required value kV plus 10 kV, 1=signal, and in the comparator 88 this signal is sent from the operational amplifier 84 to the negative input of the comparator 88.
The V return signal is subtracted from or compared to it. Output jO2 of comparator 88 is logic 1 or logic 0, and logic O
A change from 1 to 1 is an indication of overvoltage, and a dedicated inverter/microcontroller will forcibly stop the operation of the device through the software/routine. This overvoltage characteristic, when detected, trips or shuts down the inverter within as little as 10 microseconds. This is 1000 to 2000 times faster than the overvoltage response in conventional devices. This feature thus protects the x-ray tube, high voltage rectifier 29 and high voltage transformer 28 and increases their lifetime. For this reason,
These components must be able to withstand overvoltages of only a few microphones (1 second). Another feature of the protection shown in FIG. 6 is against unbalanced differences in the anode and cathode voltages to ground in the x-ray tube high voltage circuit. This is done through an amplifier 91 having an input 1111L2. These inputs are subtracted and the output is applied to a comparator 92 and compared with a 5 kV reference signal. The unbalance between the anode voltage to earth and the cathode voltage to earth is 5k
If it is greater than V, the comparator 92 is tripped off and the output 03 immediately stops the operation of the device by the inverter microcomputer 41. This protection circuit can detect manufacturing defects that occur in the secondary coil. That is, if the number of turns on the secondary side is incorrect and a difference of more than 5 kV occurs, this error will be detected during testing with the circuit just described. Furthermore, the dynamic performance and operation of the If another unbalance occurs due to a short circuit, the difference in the output will be greater than 51V, and this abnormality can be said to be a fault condition, but it will be detected by the output of the window comparator 92. The logic signal that represents this is data.
Sent to operator console via link. The protective action of comparator 92 thus detects possible faults, damage or deviations in either the secondary coil of the transformer, the high voltage rectifier, the output filter or the x-ray tube itself. Another protection circuit concerns that when the film is exposed to X-rays, most exposures occur when the energy level is greater than 15% of the claimed level. Comparator 93
is provided to receive a kV return signal on its positive input and a signal representing a 75% kV request level on its negative input. This 75% signal is extracted by operational amplifier 90 and voltage divider 95. When the power i~ turns on the transistor inverter 27, the kV output begins to rise and when it reaches 75% of the kV request value, the comparator 93 is tripped on and produces an output 04,
In this way, the dedicated microcomputer 41 is informed that the level of 75% of the kV required value has been reached and that the exposure time is the number of sets J at this time. Related properties of this circuit! The function detects any defect in either the power 1 to transistor inverter 27 or the integrator circuit during the kilovolt rise time and returns the kV after a certain period of time, e.g. 2.5 milliseconds. When the voltage does not reach 75% of the required value, power 1 to transistor inverter 27, its peripheral circuits, high voltage transformer 28, rectifier 29
, assume there is a problem with the filter or feedback circuit. At this time, this signal is used to turn off the device and disconnect it for safety. All three IC protection circuits described above are concerned with sensing deviations in voltage levels. There is also a need to sense when excessive current levels occur and to protect against them. S Two identical circuits are provided, which are for redundancy but have the same setting level.The reason for the redundant circuits is that in the event of an overload, the inverter 27 This is because the inverter tries to generate current and therefore some method must be used to prevent the inverter from doing so.Such overloads can occur due to flashovers in the X-ray tube, or e.g. This may occur due to arcing in the device or a short circuit in the output diode.Furthermore, the saturation prevention circuit 4
When transformer 28 goes into saturation, the current increases and this increase in current tends to lower the rail voltage, leading to problems with the output. For this reason, the current limit circuit is
It is constructed with two redundant circuits with the same setting level to ensure that the device is protected even if one of the current limiting channels fails. To explain the operation, it has a ferrite iron core, is connected in series with the primary side of the transformer 28 at the output of the inverter, and the current limiting action is started by a pair of current transformers 94 and 96. Each output is a differential amplifier 97
．． 98. These amplifiers are inverters,
Caused by radiation etc. 4! ? The exclusion of common styles is very strong in order to avoid noise. Out) Jls. 16 are applied to respective precision rectifiers 99, 101, which produce DC level signals proportional to their respective currents with little delay. This is in contrast to conventional RC'zf' waveform schemes, which have the luxury of long delays, such that the device is no longer sufficiently safe in terms of full response of current limiting action. The outputs of precision rectifiers 99, 101 are applied to the positive inputs of comparators 102, 103, respectively. The two current limit W levels are connected to the respective comparators 102 and 103.
is applied to the negative input of In operation, when the current exceeds the set level of either comparator 102 or 103, output 05 or 06 is turned off and applied to logic control+ device 37, as shown in FIG. The inverter 27 is controlled so that transistors 1 to 1 are selectively shut off, so that the current is automatically reduced and delayed, and then turned on again in the next half cycle. In this case, the current limit is that the energy stored in the transformer windings is circulated through the upper transistor, which was previously carrying 8 current, through the complementary lower diode and its diagonal lower transistor. It is necessary to note that the upper transistors 11 and 12 (see FIG. 4) work by covering the upper transistors 11 and 12 (see FIG. 4) in order to be able to perform the same function. This is the inductive current or inductive energy +17
However, the loop made up of the lower flywheel diode, transformer winding and lower I transistor guarantees attenuation. Such a damping loop is more effective than if all four 1~ transistors were turned off. When the four transistors turn off, inductive energy must be dissipated from the lower flywheel diode through the transformer to the diagonally upper flywheel die suite. In this loop, energy is fed back to the DC rail and decays much faster, allowing the lower flywheel die and lower transistor to conduct. This fast decay of the circulating energy that is fed back to the fixed DC rail via the diodes on the lower and opposite sides of the diagonal leads to an even faster shedding of the current limit, so that the control It is possible to switch transistors on and off at constantly high frequencies. For this reason, the procedure for shutting off the upper transistors (11 to 11) and the operation of the device is -fi, J:
＜ It was found through experiments that this leads to the following. Thus, change 1. The attenuation time of the current when circulating through the lower l transistor and die A-1 passing through the primary side of the F converter is further increased,
This avoids spurious operation at the high frequency of the inverter.Next, we will explain the digital operation of a protection device like the one shown in Figure 7.The center of this device is a dedicated inverter. Microcomputer 41. This microphone [1 computer generates the upward 1-1 slope of the sawtooth generator, synchronizes with the error signal, controls the output 1-transistor, and detects possible faults. case,
In order to confirm this and to record any kind of error detected (if any) in the output circuit, the entire system is There are several signals sent from the microprocessor 30 to the microcomputer 41.1 First, the microprocessor 30
There is a Z>exposure command signal generated within the camera that notifies the microcomputer to start exposure. Before this occurs, the microcomputer 41 checks that (1) the power supply status and the four 1 to 1 transistors are in the correct status;
and (2) to ensure that the power transistor controller 38 is in the correct state before starting ``11''.
The status of the entire device can be checked using different signals. The microcomputer 41 is a microphone [Procera 1)
In addition to the exposure command signal from 30, it also receives an exposure time signal and a phase voltage control signal. The exposure time signal determines the length of the exposure time, and the phase voltage control signal is used to generate output pulses with very small pulse widths to provide output compensation for low energy exposures. . In this way, pulse widths can be scaled small, with durations of a few microseconds, to ensure tight control through closed-loop feedback (ψ) operation. The well-known phase voltage control method uses the 1st transistor on the lower side of the diagonal as the transistor on the 2nd side! 1
11, and for this reason a minimum amount of time, typically 20 microseconds, to ensure that the l<C slowing circuit is completely discharged before switching to the output (A). To,
- If the pulse width of the upper transistor is fixed, then both transistors conduct for a period of time determined by J:τ in closed-loop feedback operation.
, Therefore, if the energy required for exposure is at a low level, the phase voltage controller adjusts and generates a very small pulse width at the diagonal output of the power transistor inverter,
Achieve 1x1 when exposing with high accuracy and low Cornergi. 75% output From the protection circuit shown in Figure 6, the output voltage has reached kVl + 75% of the required value;
The microprocessor 30 also informs the dedicated microcomputer 41 that M number of exposure times should be started. For this purpose, the microcomputer 41 sends a signal back to the central microprocessor 30 called exposure start. At this time, the microprocessor 30 starts counting the exposures. For redundancy, this is also done by the console's microphone [1] monitor via the cabinet screw 1~/console 1-to-1 link. If an overvoltage or unbalance condition occurs during exposure, the microcomputer 41 issues a signal to stop the output to the inverter transistors 1-+-"I-a." No. 564.612 of the pending U.S. Patent Application Serial No. 564.612 filed on the same date as this application.
By driving the power transistor controller 38, output 1 to transistor I"l-l'
A set of theories that control 4! P Ant Goo 1~G4. G
5. G6. There is a G7. Generally speaking, G4 controls the upper transistor 11, G5 controls T2, and G
6 controls 1-3, and G7 controls T4. , the main signal that enters G4 is the drive signal from the microcomputer 41 related to 1-1, and this is
4 and the modulation signal is coupled to the transistor T1 (G4
I want to be able to modulate the output. Similarly, the micro-combination terminals 41 each have gates G5. G6. Drive signals related to 1'2, 'l'3, and -1'4 are supplied to G7. Goo 1~G/I, G5. G6. There are two other inputs to G7, the absence of which can stop modulation. One example where this signal (state PS) is absent is:
This occurs due to a failure of the transistor drive power supply. If it is absent, the transistor I-+--! '
The other signal that acts on the ladder that stops the operation of step 4 is a signal called short circuit protection or shoe 1 to through protection.
This is the optical fiber Ill interface between one transistor, e.g. -11, and its complementary transistor T3; ]1 cannot be switched on either. This is because if such a thing were to occur, vertical shoot-through would occur and damage the second 1-transistor. The feature of this protection is that it is bidirectional; if 1-transistor 3 yields, then 1-
1 cannot be switched on. Conversely, if ]-1 yields .eta., the interlock will control ]-3 before 13 is switched on. The same is true for C for the other vertical branch of power transistors 12 and 14. The signal applied to the upper two transistors]-1 and 1-2 is a line protection signal. The absence of this signal C is caused by a failure of the kV output protector, which is supplied with three different signals for gate G10. This guard line is generated by AND gate G10. AND gate G10 is switched by the second current limit level, overvoltage protection or unbalance protection, so that the output from G10 is 0 and 0 through G4 or G5. , l-transistor]1 and T2. Any of the second current limits, overvoltage or unbalanced circuits will disable the outputs of transistors T1 and T2 to avoid a fault condition occurring in the power output stage. The lower transistors T3 and ]4 do not experience as much weight (?N11) because they span the entire half cycle of the waveform from the high voltage transformer. Regarding the signals applied from terminals 1-3 and 1-4 of the tree to groups 1 to G6 and G7, respectively.
The 7h current limit characteristic represented by the "modulating" signal on G/I and G5 also acts on the upper two transistors 11 and 12. ['''' The encoder 41 generates a synchronization signal, which is applied to the flip knob 404.10 (i. gate G9 is controlled by the first current limit and the output of OR gate G8; The OR gate G8 receives the pulse width modulated pulse train signal and the 2 signal generated from the output of the flip knob 106.
Receives a synchronization signal that is a 0 microsecond monostable pulse. This lowest pulse is generated because the RC series circuit at the output of the transistor inverter is fully discharged before the transistor turns off, thus causing the second breakdown of the transistor. This is to protect FJ from occurring. To explain the operation, if we assume that the second current limit is set to The output is the same 1111 L/S as the second current limit, and the flip 70 knob 106 is associated with the turn on of the 20 microsecond monostable lowest pulse through the first current limit and to ensure RC discharge. After that, the microcomputer 41 generates the main frequency waveform of the transformer, which is immediately applied to the lower one transistor T3.'I'4. The pulse width modulated pulse train signals passing through GB, G9 and flip-flops 104 and 106 are synchronized, and the pulse width modulated pulse train output at the end of B is synchronized with the rise time of the rectangular wave meter IX device. transistors 1 and/or 2. At this time, i.e. at the end of every half cycle, the microphone 1 computer 41 applies voltages to diagonal 1, i.e. from transistors 1 and 14 to diagonal 2,
That is, a signal is generated to switch transistors T2 and T3.
Rir. Finally, the microphone 1]=] computer 41 also generates fault errors and error codes, which are sent back to the control microprocessor '4j30, at the point when one of the possible faults has occurred. Also, let us know what type of failure it is. The binary number is decoded and sent via the data link to the console 49. In this way, the microcomputer 41 controls the power inverter, generates waveforms, communicates with the central microcontroller on the Kisepinetsu I side, receives commands, and, most likely, the power circuit. It has great commercial flexibility in that it provides information to the equipment to make decisions such as whether or not a failure will occur. High-voltage shunt and high-voltage divider feedback circuit The high-voltage shunt 44 and high-voltage divider feedback circuit 46 of FIG. 3 are shown in more detail in FIG. , provides protection against excessive voltages that can affect electronic control circuits.The unique feature is that it replaces the large number of components of a conventional shunt.
Necessary output power 4 from the DC side of the high voltage rectifier 29]
11 is used in a secondary role. The parts that need to be added are the lower resistor 107 and the lower resistor 108 shown in Figure 8, which makes the physical size of the shunt much smaller and , the inherent inaccuracies and costs associated with using a large number of elements, as is done in conventional shunts, are avoided. Ideally, for low-energy generators (in the 0.25 mA S (1) range), the filtering capacitance should be kept to a minimum. For example, in a preferred embodiment of the invention, the capacitance 1 of capacitor 7 is about 5 nanofarads from anode to ground. There is also an identical capacitor and an identical shunt between the cathode and earth. The purpose of using the lower capacitor 107 is to collect the output signal at a low voltage, i.e. in the range of 5 to 15 pols, so that it can be fed back to the control circuit without adverse effects v11. . A related problem that must be addressed is the large electrical spike when the x-ray tube exits the tube. This is a phenomenon that can occur either between the anode and the earth, between the cathode and the earth, or between the anode and the cathode. Of course, if some kind of protection is not provided, there is a possibility that such an overvoltage will damage the control circuit. The scheme used in the present invention to protect the equipment U is such that, in the worst case scenario, the control circuitry connected to the operational amplifier 84 is
To ensure that there are no overvoltages that could damage the equipment, provide a voltage filtering action from the ground down. In order to take advantage of this protective effect, the conditions for the voltage divider in the case of a nominal maximum of 75 kV are determined by multiplying the ratio of capacitor 107 to capacitor 71 by 2,000 as 9. Capacitor 1 () 7 +ff [l; L 10 microphone [about 1 farad, voltage Vx is 37, ! Equal to i volts. The voltage at point Vy in the voltage divider circuit applied to the operational amplifier 84 is connected to the resistors 109 and 1
11C, voltage 13': VZ is approximately half the value of Vy, that is, 4.99 volts or, for example, a voltage close to 5 volts. When considered in relation to the voltage of the same cathode-to-ground voltage divider, at an output level of 150 kV, this 5pol1~
The voltage divider's input voltage becomes 10 pol]-. When the voltage is divided, the casshood filtering action comes into play. First, if a transient occurs and is higher than the predetermined level, the arrester S P 1 provides protection against spikes. In this case, another element that contributes to the wave action is the coaxial couple 112, which has an inherent capacitance and inductance. It has a tendency to reduce the radiation noise coming from the side.As shown in the figure, a second lightning arrester Sl) 2 is installed to ensure complete protection.In addition to this cascade mill wave action, , 113, 114, 116 and a resistor 117 are provided. There is another filtering effect between the resistor 117 and the capacitor 114. Finally, the resistor 109 and the capacitor 11
6 has a filtering effect from y to VZ. resistance 10
9 and capacitor 116 are the resistors 019.9 and 116 mentioned above.
Calculate the effect in relation to +11 divided by 1 effect. The cumulative effect of this is that for an anode voltage of 75,000 pol I~, the nominal value of the voltage VZ is 5 pol 1~. The circuit of FIG. 8 also has diodes 118 and 119. Depending on the polarity of the spike, these diodes either redirect the spike through diode 118 to the 15 volt supply, or VV through diode 119
hl, I and others turn towards the earth. Furthermore, operational amplifier 84 includes diodes 121 . +22 and is protected from overvoltage on its input terminals. (1) Surge arrester SP1, coaxial relay 112, cable 113, air gap A1, lightning arrester SP2 and resistor 109,
111. The filtering effect of the combination of elements including the voltage divider of 117, (2> The filtering effect of the combination of the resistor 117 and the capacitor 114, (3) The filtering effect of the combination of the resistor 109 and the diode υ116, and the diode. The effect, combined with the ability to return energy to the power source via 118 and 119, is that the level of sharp spikes that occur on the high voltage side can be severe enough to cause damage to electronic control circuits. - It is maintained so that energy is not added.Next, regarding the setting of the high voltage shunt 44 (to clarify for the record, its transient response is 1) -11, resistance T1 ndenza 10 with equivalent resistance at 69 and ■×
7, but in steady state,
Its accuracy is related to the equivalent resistance of resistor 69 and Vx. The relatively small value of damping resistor 124 is provided to dampen any extra ripple vibrations on capacitor 71 that may be introduced by the series interface. For this reason, it is preferable that the high vacuum density 71 has a small intufftance value of less than a nanohenry. Resistor 69 and series resistor 117.109.111
The relationship between the equivalent resistance constructed in parallel with the resistor 108C and the capacitors 11 and 107 is such that their time constants must be the same and that the required voltage or voltage of v× is up to 75 It is determined that for an anode voltage of kV, it should be around 40 volts (as required by the Unterritor Laboratories). Very easy adjustment to shunt settings
'c, the only adjustment required is the tolerance of capacitor 51,! lI! It is mentioned here that there is a case where the target is about 5%, and there is a 21-degree 101 mJ adjustment.
Osa), two. In contrast to this single adjustment, traditional flow dividers typically require a large number of adjustments because they use a large number of components; . Another advantage of the current shunt of this invention is that the number of C1 components from the high voltage side to 37.5 pol 1 to Vz is kept to a minimum with only 2 resistors and 2 capacitors. This means that the deviation of errors is minimized. The tolerance required for the resistor is as tight as 4. This is because the number of resistors is only two. For this reason, a flow divider with a fast transient response can be manufactured in a cylinder with high precision at a low cost. Conventional shunts using resistor gates typically do not have good transient response and fast rise times, but the device of the present invention, which operates at high frequencies with very rapid rise times, provides closed-loop feedback. A good transient response can be obtained so that control can be performed. Furthermore, it is possible to operate with greater bandwidth, thus improving the response of the device. Mixing Amplifier and Feedback Controller A high voltage feedback control system is shown in FIG. 9 and includes the well-known single stage for generating a variable voltage command for low level electronic circuits on the order of 0 to 10 volts. I'm here. In the present invention, this signal is generated by a central microprocessor 30 via a digital-to-analog converter 52. The kilovolt feedback signal returning from the output via line 12 is converted by an analog-to-digital converter 53 and fed to the same microphone 11 processor 30 to ensure proper operation of the device as well as feedback voltage. Verify in real-time operation that C follows the command voltage. The preferred power stage shown in FIG. 9 first includes a horn transistor inverter 39. The power stage shown in FIG. This inverter is 24 kV
In order to be able to operate at a voltage of 1 volt, the two fixing taps 1 and It acts on one of (1). In Fig. 9, an equivalent circuit is shown in which the output of the power transistor inverter 39 is connected to the high voltage side (1), which is called the primary side. The components are transformer leakage inductance, inductance and associated series inductance 1.1,'q
They are a P-wave capacitor CF and a variable load RL. For the primary side, this P-wave capacitor is multiplied by the square of the turn ratio of the transformer to its value at high voltage, resulting in a very large value. For the side,
Variable in a very wide range 19 (!!!For type 3, it is 1.25 (1 m8 to 0.1 m△ and 15.0 (+
(There is a large load power change on a one-to-one basis or in some cases historically). Also shown is the previously mentioned high voltage divider. This is a voltage return! mixing, amplifying and controller 34 and central processing unit H3 for
Generates a 100 volt feedback signal to avoid zero operation. One of the most important features of the idle loop feedback control device is the phase advance and return of the voltage feedback performed by the resistor 12G, resistor 127, and capacitor 128. resistance 1
The phase delay of the mixing amplifier 129 is obtained by the time constants of the capacitor I and the capacitor I 132, and their joint action generates a kV return error signal, which changes during the rising period. , in a steady state, it can be regarded as a DC signal, and is compared with the sawtooth wave generator ζ36, and the comparator 13
A pulse train of pulse width modulation is generated at the output of 3. It should be noted that the saturation prevention circuit 47 described above also has a closed loop circuit with a portion of one to several loops. This is because when we dynamically change the slope of the waveform from the sawtooth generator, we are also changing the effective gain of the overall loop controller. Saturation Prevention 1 [Although the fluctuations in path 47 are small and change dynamically according to the saturation level reached by the transformer itself, the variable gain introduced into the loop improves the stability and performance of the device. This is another important feature that must be taken into account for operation. The purpose of the loop delay circuit is to filter out noise in the error feedback signal.
To achieve optimal performance, the summation circuit must be minimized to improve device bandwidth, short pulse response, and feedback tracking. The phase circuit consists of a combination u of a resistor 131 and a capacitor 132.
O) characteristically defined by the time constant and the time constant of the load filter determined by the values of RL and CI.
However, since LC is variable and can vary by typically 15,000 to 1, as explained earlier, the voltage introduced into mixing amplifier 129 by resistor 131 and capacitor 132 is On the other hand, the phase delay caused by the 131 and capacitor 1320 are used to keep the return device responsive to imaging times as short as 1 millisecond.
To avoid this, the time constant must be minimized. In order to obtain a fast response of the device with the feedback of the mixing amplifier 129, a limiting circuit 13 which is available on the market and requires 1q! ) will be established. The action of this circuit is to limit the maximum output of iIt combiner 129 to a level of 10 volts. The level of this 10pol I is the same amplitude as the output of the sawtooth wave meter (G), so the maximum output of the mixing amplifier 129 does not exceed the output level of the sawtooth wave generator. During the latter part of the rise period, it avoids saturation of the mixing amplifier which would cause the response to become even slower.The second action of the phase lag device is to compensate for the noise introduced by the phase lead feature. , above the intended frequency, the phase delay device minimizes the effect of noise on the device.During the rising R, resistors 126, 127 and capacitors
The phase advance effect due to 12II clamps the overshoot to the expected level at the end of the rise time. This may affect dynamic load fluctuations or the kilopol number.
It improves the response of the device to variations in other parameters such as DC rail power level. This C phase advance circuit is used during the rise time of A-Bashu 1~
is controlled and acts as a gradient function. From a control perspective, what this does is reduce the power that the transformer and output DC filter accumulate during the rise time, controlling overshoot 11'J. The main disadvantage of using the phase advance feature is the inherent increase in the noise signal relative to the output of the mixed stem amplifier; This is compensated for at higher frequencies, so that the increase in the AC ripple generated by the phase advance is transferred to the aforementioned lag circuit and related intermediate lag circuits. effectively offset. This circuit consists of a resistor 130 and a content 4 resistor 140 connected in series.
It is composed of C and compensates for dynamic noise. The noise level in the improved device is based on acceptable noise levels and good stability. The bandwidth of the device is preferably in the range 1 to 1.2 to 1z, and the switching frequency of the transformer is typically 6k.
The l-1z culm frequency is higher than that, thus both cut-on frequencies for both the leading phase as well as the intermediate lag are sufficiently lower than the switching frequency of the variable LF converter, and due to the characteristics of the leading phase. Compensate for the increase in AC ripple generated. The stability of the device is not shown in Figure 10, but is shown by the Nicol diagram.
There is. This figure shows an overall gain margin of about 20 dB and a phase margin of 70°. Therefore, the margin of linearity and control stability of the device is very good. The apparatus of the present invention will be better understood when its various machining and performance characteristics are considered. For example, during the 31 rising period, one condition is to increase the current flow to increase the rise time, and the other condition is to avoid A-Vaccord 1 and require large parts. It will be appreciated that there is a conflicting interaction between conditions that can be reversed to limit the flow of current so that it does not occur.During the rise time, the current magnetizes the transformer so that the The flow tends to charge the output filtering capacitor 69, but is only limited by the leakage inductance of the transformer. This leakage inductance is set between the primary and secondary sides. It is preferable to minimize the Fl frequency of the signal waveform of the It is a trade-off between the switching frequency and the tolerable output voltage ripple.A feature of the invention is to increase the frequency during the ramp-up period to reduce the current limiting effect and provide a faster rise time. ,
In this way, it is possible to obtain a waveform very close to the C rectangular wave. In addition to the above mentioned f1, it is necessary to limit the current flow associated with the power supply 1 to the transistor inverter 27 to a level acceptable to the control system. Additionally, the current flow and associated rise time should be limited to avoid A-bashoot at the end of the rise time.
) must be. For this reason, the current must be controlled during the rise time, but at the same time, in order to obtain favorable results in X-ray procedures, especially for short exposures, the device can be A valid x”t J
You must achieve Z time. This kind of thing is
As previously stated, the effect of the increasingly variable mark/space ratio feature and steady state operation is also achieved by operating at higher frequencies. Another aspect of the current level interaction phenomenon described above is that during the rise time, the tank amplifier 129 is in saturation, and during the saturation period, the device quickly recovers from the transformer core saturation. Some control action must be added to ensure that the control action is maintained during this period. As explained in nFi, in order to set this control effect, the output limiting circuit 13 is connected to the feedback of the 4-combiner amplifier 129.
5 to prevent the error signal from going to an extreme position where it takes a long time for the capacitor 132 to recover. The anti-saturation circuit 47 quickly brings the C1 mixing amplifier in the transformer 28 out of saturation and achieves a good control effect at the end of the rise period. At the end of this rise time as well as during the rise time, the anti-saturation circuit During this rise time, the mark/space ratio or the dynamic (2 asymmetrical 1) of the volts 7s must be balanced to ensure that the inverter 21 and the variable The increasing variable mark/space ratio during the rise time controls the slope of the voltage demand from the microcontroller 30 to 1/A converter 52 to increase the rise time. A small pulse at the beginning of the V rise time can be achieved by limiting the current and also controlling the rise period, with a current limit placed on the power inverter. Figure 11A shows the performance of the device of the present invention when operated using various parameters in an X-ray tube Ml-100 manufactured by Electric Company.
A typical exposure time of 32 ms per l using a current load of 640 mA for different kV levels of 9(l and 1(l) kV was observed. It will be observed that the curvature during the rise time is very good at 2M, and the overshoot at the end of the rise time is tightly controlled due to the phase advance compensation. It can be seen that during steady-state operation, the output ripple decreases with the output voltage, and the mark/space ratio increases by 1 and H'J above the voltage.However, in both cases the ripple is negligible. Figure 11F3 shows 12!ikv and h4001IIA, respectively.
The exposure time is shown in 1 millisecond as the voltage and current level change. One thing to notice in this figure is that the effect of the feedback circuit is always fast. The flat top of the kV waveform demonstrates that the feedback is acting within 1 millisecond. By controlling the accumulation of J energy in wave capacitors and transformers,
Some indications of where to prevent Bashu 1 are shown. The operating parameters are that the X-ray generator has very low mA
This is a typical value when working with automatic exposure control. For such very low mA, reproducibility is very difficult to achieve with conventional generators due to large variations in load. The present invention is able to control the number of keys even in such conditions with very fast closed-loop motion that ensures that there is no A-base, a flat top, and a fast response. In order to achieve good performance. FIG. 11C shows exposure at 110 kV and 400 mA, and shows the current waveform of the inverter. This figure clearly shows the difference between the two different 014s. The first period, the non-linear rise period, indicates that the initial flow can flow at a level C much higher than the steady state level, resulting in a faster rise time. Furthermore, for a period of ten minutes,
It shows that the waveforms are highly symmetrical even during steady-state operation. This is due to the action of the saturation prevention circuit 47.
Table 4 shows that there is no saturation phenomenon of the transformer. Furthermore, this figure shows that the transition between the start-up period and the steady state occurs smoothly and rapidly 4) at the end of the phase-advance operation and at the time when the seed part of the kV number is reached. The kV step response is shown in Figure 111), which shows a 200 mA, 7 (l kV waveform with a 15 kV step in between. As can be seen from this figure, the A-barge) -1 to 4, and the drop rj 115 is very fast (about less than 1 millisecond). ). Figure 111 shows the same waveform -C, but with a higher frequency and a superimposed staircase of 7.!1kVp. Figure 11E is similar to Figure 11E, but with 75 kVp superimposed on 7.5 kVl]. However, the time scale has changed and the frequency is 11.'i!+
, 5k+-17°. From the performance data shown in FIGS. 10 and 11, the following conclusions can be drawn regarding voltage feedback controllers. (
1) Across a wide range of systems used in X-ray generators, A-basicote is virtually exclusive. (2) At least 5.5 kl-12 to 1-rough: 1
performance is very good. (3') There is no instability, and rather the linearity and fQ characteristics are very good. (4) Its behavior is that of the primary device 1/(1-+-S-1'). The present invention has been described by way of specific embodiments and examples.
Various modifications will occur to those skilled in the art from the above description. Therefore, it is understood that within the scope of the present invention, this invention may be practiced otherwise than as specifically described herein.

[Brief explanation of the drawing]

FIG. 1 is a schematic diagram of a conventional xm generator, FIG. 2 is a schematic diagram of an X-ray generator of the present invention, and FIGS. 3A and 3B are circuits of the voltage feedback and lil control portion of the preferred embodiment of the present invention. Figure 4 is a circuit diagram of the saturation prevention circuit of the present invention, Figure 5Δ and Figure 5B are the inverter control system of the present invention @
C. Simplified graph showing typical pulses generated, No. 6
The figure is a circuit diagram of the divine protection circuit used in this invention, and Figure 1 is a circuit diagram showing the digital part of the protection circuit of this invention.
FIG. E3 is a circuit diagram of a divider according to a preferred embodiment of the invention; FIG. Figure 10 is a graph showing the 1:1 volt feedback of the present invention, and Figures 11A to 11F are graphs showing various characteristics of the performance of the present invention. This is a graph showing the traces. Description of main symbols 23: 3-phase input 24; rectifier 27: inverter 28: high voltage transformer 29: rectifier 37: logic control device 4γ: saturation prevention circuit 48: current limit circuit 67: current transformer patent applicant General Electric・Company agent (76)
30) Iku Numa Toku 2 FIG, 2 0 5 to Is 2025 303540#l (
8riozu1) FIG, llA 3 IQ Is 20253035404riF...
cit castle<Ki+234 81 o r. FIG.
1. Indication of the case Patent Application No. 268737 of 1982 2. Name of the invention Electric horn generator 3. Person making the amendment Relationship to the case Applicant's office River Road, Schenectaday, New York, 12305, United States of America , No. 1 Name: General Electric Company 4 Address: 4th floor, Kowa Building, 35, 1-14-14 Akasaka, Minato-ku, Tokyo 107 Japan General Electric Co., Ltd. Far East Patent Department Telephone (588) 5200-5207 April 10, 1985 (Shipping date: April 30, 1985) 6. Subject to amendment

Claims (1)

[Claims] 1) In a power generator used to apply a high voltage between the anode and cathode of the XFJI tube, a C
1 a power inverter capable of generating an AC rectangular wave output from the DC power supply; and a power inverter connected to the inverter;
a high voltage transformer that receives the AC output and sends out a high voltage output; a high voltage rectifier that receives the high voltage output and sends out a high voltage DC output to the X-ray tube; and a high voltage rectifier that operates the inverter in response to a control signal. a control means for selectively varying the pulse width of the square wave output of the inverter; and a control means connected between the X-ray tube and the control means and responsive to a voltage level between an anode and a cathode of the X-ray tube. and a voltage feedback circuit for generating a fast control signal to be applied to said control means to maintain a predetermined voltage level between said anode and cathode. 2. In the power generator described in claim 1),
A power generator in which the control means is responsive to selectively vary the frequency of the output of the inverter as well as its pulse width. 3) In the power generator described in claim 1),
A power generator in which the control means includes means for generating a voltage error signal that is a function of the difference between the feedback control signal and the voltage demand level generated in response to control of the A-pelator. 4) In the power generator described in claim 3),
The means for generating the voltage error signal has a phase-advancing feature that changes the voltage error signal during a predetermined rise time, thus protecting against overshoot at the end of the rise time. power generator. 5) discharged into the power generator according to claim 4),
A power generator in which the means for generating the voltage error signal includes phase-lag means for compensating for noise introduced by the phase-lead characteristic. 6. The IC power generator according to claim 1), wherein the control means includes a sawtooth wave generator having an output whose frequency and slope are variable in response to a predetermined control signal. lυ and thus modulating said control means. 7) In the power generator according to feature 8' [Claim 6], the saturation state of the power inverter is sensed in response to the current flowing in the high voltage transformer, and the A power generator having an anti-saturation circuit that supplies a signal to the sawtooth generator without reducing the saturation condition. 8) In the power generator described in claim 1),
9) In the power generator according to claim 1), the power inverter is constituted by a power transistor bridge that controls a high voltage high frequency transformer, wherein the DC power source is a three-phase controlled rectifier. A power generator consisting of a three-phase AC power supply connected to -C. 10) In the power generator described in claim 1)-
C1 A power generator in which the lightning inverter is activated at a relatively high frequency within the range of 1 to 15 Kll. 11) In a high voltage generator that applies a high voltage between the anode and cathode of the X-ray tube and controls this voltage, a DC source and the output of the DC source are converted into an AC voltage supplied to a high voltage transformer. an inverter for converting, a means for rectifying the output of the high voltage transformer to improve the quality of the x-rays, and a means for sensing the voltage level between the anode and cathode of the x-ray tube and generating a signal representative of the actual voltage. and the desired voltage between the anode and cathode t
means for generating a signal representing an actual voltage and a signal representing a desired voltage for selectively switching the inverter on and off to maintain the desired voltage; A high pressure generator connected to a control means. 12. In the high-voltage generator as set forth in claim 11), the sensing means has a shunt circuit having a dual-functioning capacitor, and the capacitor V acts as a high-pressure capacitor of the shunt circuit. The high voltage generator acts as a ripple capacitor for the output of the rectifier means. 13) The high voltage generator according to claim 11), wherein the control means includes a microprocessor, the microprocessor receiving a signal representative of the actual voltage via a D/A converter; A high voltage generator generates a voltage request signal for subsequently comparing the actual voltage with the representation signal via a /D converter. 14) The high voltage generator according to claim 11), wherein the inverter is a rectangular wave inverter operated by pulse width modulation method C. 15) The Iζ high voltage generator according to claim 14, wherein the inverter comprises a power transistor bridge controlling a high voltage high frequency transformer.