JPS5931252B2 - semiconductor switch device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to improvements in semiconductor switch devices, and in particular,
This invention relates to an improvement in a device in which control pole power is supplied from the main power path.

In high-power semiconductor switches such as power transistors, the control pole current itself becomes large, and constantly supplying the control pole current to cope with the large main current from the control circuit complicates the control means and increases control means loss. increase.

For this reason, a method has been proposed in which a current transformer is provided in the main power path to provide a control pole current in response to the main current.

However, when a current transformer is used, since an electromagnetic current transformer operates with alternating current, it is not suitable for operation over a wide range of current-carrying time ratios.

For example, there is nothing that can guarantee current transformer action in direct current or low frequency alternating current (low frequency range in variable frequency).

On the other hand, DC current transformers have the disadvantage of being complicated and expensive.

The present invention was made to solve the above-mentioned conventional drawbacks, and an object of the present invention is to provide a simple control electrode power replenishment method that can handle a wide range of energization time ratios.

FIG. 1a is a circuit connection diagram showing an embodiment of the present invention.
In the figure, a power path (
100-200-XY) inserted in series with a power semiconductor switch 1 (for example a power transistor).

Furthermore, in a power semiconductor switch in which an auxiliary solid state switch (for example, a transistor, a gate turn-off thyristor, etc.) 2 is connected between the first main electrode (collector) and the control pole (base), the primary winding is inserted in series in the power path. N1
and a transformer 3 having a secondary winding N2 inserted in series with the auxiliary solid state switch 2, thereby supplying the secondary winding current in response to the main current flowing through the power path to the control pole, and the secondary winding N2. Line voltage is supplied to the auxiliary solid state switch 2 and the branch to the control pole.

That is, the transformer 3 uses the first main electrode current 11 of the power semiconductor switch 1 as a primary input, and supplies a secondary current 12 to the control pole of the power semiconductor switch via the auxiliary solid state switch 2.

At this time, in order to compensate for the voltage drop of the auxiliary solid state switch 2 and the control pole voltage drop (for example, VBE), the secondary winding voltage VN2 (the polarity of + and -1 in the figure is the forward direction) is supplied. be.

Thereby, even if the voltage between the main electrodes (for example, VCE) is low, a sufficient control electrode potential and thus a sufficient control current can be ensured.

In the transformer of the embodiment shown in FIG. 1, the primary winding N1 is the secondary winding N2.
The sum of the illustrated N2 and the primary winding N1 corresponds to the secondary winding N2.

Now, one embodiment of the present invention further includes a tertiary winding N3, which is connected between the control pole (for example, the base) and the second main electrode (for example, the emitter, the main electrode that also serves as the control pole).

The tertiary winding N3 has a polarity that provides a control pole potential in the forward direction due to the primary winding current, but has a polarity that provides a control pole potential in the reverse direction due to the secondary winding current.

Furthermore, current regulating means 7 can be provided in series with the tertiary winding.

This current adjustment means is used when making the tertiary turns ratio N3/N1 smaller than the secondary turns ratio N2/N1, or when making the tertiary turns close to the secondary turns (however, N3<N2) so that the secondary winding N2
Provided when the control electrode current is shared and supplied by acting in parallel with the control electrode current.

In the former case, for example, a resistor, a diode, or a series body of a diode and a resistor as shown in FIG.

In the latter case, it is resistance.

Now, in the embodiment of FIG. 1, when a forward control voltage (or control current) is applied between terminals S and E, the auxiliary solid state switch 2 first becomes conductive.

On the other hand, the power semiconductor switch 1
Since there is always an EC delay, do not turn on the power immediately.

Furthermore, even if a small number of the main currents 1 are energized, the control current "B2" is insufficient to energize all of the main current iL.

Therefore, again, the voltage VCE of the power semiconductor switch 1 does not drop immediately.

Therefore, the voltage between terminals X and Y is maintained.

And 100-200-X-N knee 2-N3-7-Y
The current flows first in the route that follows.

At this time, the current iL to be applied to the load circuit 2000 is
In the case of a normal load current larger than the excitation current iEX of the current transformer 3, even if the excitation current of the above route flows, the voltage between the terminals X and Y hardly decreases.

In particular, in the case of a load circuit having a flywheel diode antiparallel to a normal inductive load, the X-Y voltage does not decrease when iL>iEX.

Therefore, the X-Y voltage is applied to the series windings N2 and N3.

However, the secondary winding and tertiary winding N in the above route
Since it is a series with opposite polarity to 3, the composite number of turns is (N knee N3
).

Therefore, the excitation current iEx has an excitation ampere turn of A
TEX, 1Ex=ATEX/(N knee N3'
).

Then, for this composite number of turns (N4-N3), the terminal
-Y voltage voltage Y is applied.

Further, due to the above, the following voltages are induced in the secondary windings N and N3, respectively.

Furthermore, both winding electromotive force vN'2. vN3 has negative polarity on the illustrated/marked side.

Therefore, the power semiconductor switch 10 control pole is biased in the opposite direction.

Here, the voltage drop between the x and y terminals of the current adjustment means T is expressed as v7.
If Xy, 7 constituent elements 71, 72, etc. are selected so that I VN 31 >V7 xy.

For example, 70 equivalent resistance display is r7-v7Xy/iEX
Then, select r7<l VNa I,/iEX.

As a result, vBE-1 (1vN31-vXy) reverse biases the control pole of the power semiconductor switch 1, keeping the power semiconductor switch 1 non-conductive.

That is, the tertiary winding N3 is connected to the power semiconductor switch 1 as follows:
Provides reverse bias (cutoff maintenance effect).

On the other hand, regarding the closed loop consisting of N3-7 - terminal e-6 to terminal b-2 - N3, the internal voltage of the control means 6 and the above (1vN3
1-V7Xy) is summarily superimposed, the control current iB2 increases and the auxiliary solid state switch 2 becomes increasingly conductive.

For example, if the inside of the control means 6 is made up of a control power supply 61, an on-control switch 62, and a control resistor 64, then the sum of 61 voltages (I VN3 I-V7x) is 61.
4 and 20 control poles.

Here, if an anti-parallel diode 9□ shown by the dotted line is connected between the control terminals b and e, the control resistor 64 may have a high resistance, or even if 62 is cut off after a predetermined short time, N3 - Auxiliary solid state switch 2 through 7-9-2
0 control current jB2 is maintained.

That is, the tertiary winding N3 has a positive feedback effect on the auxiliary solid state switch 2.

Due to the above action, the period tR during which the power semiconductor switch 1 is cut-off and the auxiliary solid state switch 2 is conductive
sustain.

This period is shown in second operation waveforms t1 to t2.

22(b) and (c) in the figure are the control current iB2 of the auxiliary solid state switch 20, the tertiary winding voltage VN3 (shown with the marked side as positive polarity in FIG. 1), and the control pole current iB1 of the power semiconductor switch 1, respectively.

Furthermore, the control pole voltage VBE of the power semiconductor switch 1 is also
Although the level ratio of each waveform is different, it is roughly jBz in C of the same figure.
It shows a similar pattern.

During the tH period, the magnetic flux of the magnetic core of the current transformer 3 is directed towards the negative saturation level.

That is, the core magnetic flux is reset.

Next, when the core magnetic flux reaches near the negative saturation level, the excitation current iEX increases, the magnetic flux voltage decreases, and (I VNa
I-V7X7) < 0, that is, VBE>O.

That is, the control pole of the power semiconductor switch 10 transitions to forward bias.

This point is time t2 in FIG.

Then, the semiconductor switch 1 becomes conductive tx'
), the X-Y voltage vxy decreases, lVN,1 further decreases, and VBE becomes forward.

Thus, finally, 1 becomes completely conductive and the load current iL
flows through the primary winding N1 and the power semiconductor switch 1.

By the way, the voltage drop V2 (ON) of the auxiliary solid state switch 2
and the required control pole voltage VBE of the semiconductor switch 10 is the desired voltage drop VCE of the power semiconductor switch 1.
(ON) Higher.

That is, VCE(ON)≦V2(ON>+vBE・
(2).

In other words, the voltage drop of the auxiliary solid state switch 2 ■2 (O
It is desirable to reduce the voltage drop across the power semiconductor switch to less than the sum of N) and VBE.

If VcE (ON) ≧ v2 (ON) and 10 vBE are sufficient, the current transformer 3 itself is useless, and in this case, VCE
(ON) is high, leading to a large amount of power loss and making cooling means of power semiconductors and other devices disadvantageous.

In the normal usage range, the relationship shown in equation (2) is established.

In the device of the present invention, when the power semiconductor switch 1 starts energizing, a voltage of 1 (V2(ON)+VBE VCE(ON)) is applied to (N'+N)=N2, and the magnetic core flux changes from the negative saturation level. , begins to change towards the positive saturation level, and the core flux rises through the unsaturated region.

In other words, due to this increase in magnetic flux, (vBE+v2(O
N))), and the primary winding current 1
1 is supplied to the control pole current iB1.

During this time, the current adjusting means 7
Therefore, almost no current flows.

In addition, if the number of tertiary windings is made relatively thick and there is only a simple resistor 71 as shown in Figure 1, then VNa>0 (the mark is positive), so Also control pole current I
It is possible to share the supply of Bt.

This current transformer action continues for a period of time until it is turned off by the control means 6, which is the section t3 to t4 in FIG.

Note that t2 to t3 is a delay time from when the power semiconductor switch 1 is turned on until the voltage between the terminals X and Y drops.

Next, when the control means 6 applies a reverse bias between the terminals b and e, the respective control poles of the power semiconductor switch 1 and the auxiliary solid state switch are reverse biased and turned off.

At this time, if the power semiconductor switch 1 is turned off first, the current transformer voltage vN will transiently become VCE>(V2+VBE).
2゜vN3 has negative polarity, and the section t5 to t at the beginning of Figure 2
6, and some core flux reset is performed during this time.

Moreover, if the auxiliary solid state switch 2 is turned off first, (v2+vBE-v(,E) increases transiently, and the current transformer positive polarity voltage increases, but this voltage increase is caused by the voltage limiting means 4. (For example, a diode, a series body of a diode and a resistor, a breakover switch element, a series body of a nonlinear resistor such as a varistor, and a diode, etc.).

In this case, the dotted line B in the section t5 to t6 at the exit of Figure 2
The current transformer voltage waveform is as follows.

The diode 8 in FIG. 1a is the auxiliary solid state switch 2.
To turn off the power semiconductor switch 1 in advance to reset the magnetic flux, or to turn off the auxiliary solid state switch 2 in advance to prevent the reverse bias current of the power semiconductor switch 1 from running out. belongs to.

As mentioned above, the reverse bias can be applied, for example, to the resistor 64 in FIG. 1a.
A parallel capacitor 65 is provided in the OFF control switch 6, and the charge accumulated during the ON control is used as a pulse power source to turn the OFF control switch 6.
3 can be applied.

The same operation is repeated thereafter.

In the embodiment shown in FIG. 1, according to the present invention, the magnetic flux of the current transformer magnetic core can be reset to near the negative saturation level before conduction.

Therefore, during the conductive current transformer operation, the magnetic core can be fully utilized from the negative saturation level to the positive saturation level.

By the way, the secondary winding N during the aforementioned magnetic flux reset period tR
2 Applied voltage ■RN2 is the tR period voltage Vxy between x and y.
(tR) -E, however, there are N approximate expressions >> N 3, N 2 >> N.

On the other hand, the secondary winding voltage vc'r during current transformer operation is VCT
-(V2 (ON)+VBE-VCE)
-VN2 (ON)
, -...-...-(4).

Since the voltage-time integral value during the rainy period is equal, the on-time tO
N, then VRN2°tR=VcT°toN --(5) Therefore, now, if E-100~300v1(V2(ON
)+VBE VCE' ) -1 to 3V,
toNZtR ratio is 30-300 times.

That is, the CT action can be maintained and the conductivity of the power semiconductor switch can be maintained for a time period that is 30 to 300 times the reset time t R required for magnetic flux reset.

Therefore, at the maximum conduction ratio, the turn-off time t in FIG.
If 4 to t6 are ignored, the maximum energization time ratio αMAX can be as follows.

Moreover, as described above, since the core magnetic flux can be fully utilized from negative saturation to positive saturation, the current transformer can be made smaller.

Further, upon completion of the magnetic flux reset (by reaching near the negative saturation level), the magnetic flux reset mode automatically shifts to the conductive mode.

FIGS. 3 and 4 are connection diagrams showing other embodiments of the present invention, respectively.

In FIG. 3, the secondary winding N2 is connected between the auxiliary solid state switch 2 and the control pole of the power semiconductor switch 1, and the tertiary winding N3 and the secondary winding N2 share a common winding part N3. This is what I was meant to have.

In addition, in FIG. 3, the quaternary winding N4 is connected to the control pole of the auxiliary solid state switch 20 together with the resistor 11 and the diode 10, so as to have a positive feedback effect on the auxiliary solid state switch 2 during the magnetic flux reset period tR. It is something.

Of course, if the input current from the control means 6 is large enough, N
4.10 and 11 are unnecessary.

FIG. 4 is a diagram showing an example in which a four-layer semiconductor switch such as a gate turn-off thyristor is used as the auxiliary solid state switch 2.

In FIGS. 3 and 4, an open two-terminal X-Y connection body is illustrated as the basic unit of the power semiconductor switch device.

In actual application equipment, these current transformers 3 can be inserted into various power supply or load circuit networks (not shown).

FIGS. 5a to 5d are schematic diagrams showing other examples of the auxiliary solid state switch 2 and the power semiconductor switch 1 used in the present invention.

Figure a shows a parallel composite element consisting of a transistor and a gate reverse bias type thyristor, and Figure 5 shows a Darlington connected type composite element with a transistor using a gate reverse bias type thyristor as a Darlington auxiliary switch.
A power semiconductor switch 10 is applied between B and E or between G and E for the purpose of reducing the voltage drop during normal (always) energization.
Become a target.

Of course, it can be used as an auxiliary solid state switch 2.

Figure 0 shows a Darlington-connected transistor composite device consisting of an auxiliary transistor and a main transistor, and Figure d shows a device with characteristics intermediate between a thyristor and a transistor, which has some four-layer parts, and is usually controlled like a transistor to maintain conductivity. A polar current is applied.

In addition, when a switch having two control poles, such as those shown in FIGS.
In addition to the one shown in the figure, a separate one is provided as shown in Figs. 1 to 3).

In the element shown in figure b, it is effective to supply the current transformer secondary current to the control pole KB terminal of the power transistor.

At this time, the built-in gate reverse bias type thyristor has the advantage of increasing overcurrent capability, and the current transformer normally passes through.
It has the advantage of reducing voltage drop and loss in the mold.

In this manner, the present invention is effective for any power semiconductor switch 1 that desires a control pole current for better conductivity.

In the above embodiments, the magnetic flux reset is performed by sharing the secondary winding and the auxiliary solid state switch that controls whether or not to apply the secondary current to the control pole. During periods when the power semiconductor switch 1 is non-conducting, the voltage across its terminals or its responsive voltage can reset the magnetic flux in the transformer core.

Further, in the embodiments described above, the control pole of the power semiconductor switch is directly reverse biased by the tertiary winding to make it non-conductive. For example, in the embodiments of FIGS. 1 to 3, An auxiliary bypass switch is provided to bypass the control pole current of the power semiconductor switch, and by making the auxiliary bypass switch conductive using a tertiary winding, etc., the power semiconductor switch 1 is indirectly made non-conductive by the magnetic flux voltage at reset. can do.

According to the present invention, in a power semiconductor switch having a transformer that supplies control pole current, the magnetic flux of the transformer core can be reset by the voltage between the terminals of the power semiconductor switch during the period when the power semiconductor switch is off. This makes it easy to reset the magnetic flux.

Further, by making the power semiconductor switch non-conductive by the magnetic flux voltage during the reset period, a desired voltage application time integral value for magnetic flux reset can be secured.

Furthermore, by connecting the auxiliary solid state switch and the transformer secondary winding between the control pole and the main electrode, it can be used to supply secondary current to the control pole and to reset the magnetic flux, which further simplifies the process. .

[Brief explanation of drawings]

FIG. 1 is a diagram showing an embodiment of the present invention, FIG. 2 is a waveform diagram for explaining the operation of the embodiment in FIG. 1, and FIGS. 3 and 4 are diagrams showing other embodiments of the invention. The figures shown in Figures 5a-d are
FIG. 7 is a conceptual structural diagram and a circuit/circuit symbol diagram showing other examples of switch elements that can be used in the present invention. In the figure, 1 is a power semiconductor switch, 2 is an auxiliary solid state switch, 3 is a transformer, N1 is the first winding, N2 is the second winding, N3 is the third winding, 4 is the voltage limiting means, and 6 is the control means,
7 is a current adjustment means, and 8 and 9 are diodes.

Claims (1)

[Claims] 1 A semiconductor switch inserted in series in a current carrying path and energized when a control pole current is supplied to the control pole, a primary winding inserted in the current carrying path, and a secondary winding that generates the control pole current. a tertiary winding provided in the transformer; and a semiconductor switch connected in series with the tertiary winding to reset the magnetic flux of the transformer in advance before generating the control pole current. an auxiliary solid-state switch for causing a current to flow through the tertiary winding; further, the secondary winding is connected in series to the series body of the tertiary winding and the auxiliary solid-state switch; A semiconductor switch device, characterized in that one end of the tertiary winding is connected to a control pole of the semiconductor switch, and the other end of the tertiary winding is connected to a second main electrode of the semiconductor switch.