JPS59186440A - Communication system - Google Patents

Abstract

PURPOSE:To eliminate the influence of Doppler effect effectively and improve resistance to disturbance by transmitting spectrum-diffused pulses with a transmission time width that which does not exert the influence of a Doppler shift substantially. CONSTITUTION:A two-phase PSK modulating circuit 15 inputs an M-sequence code in a predetermined random time series from a PN code generator 16 through an output line 161 and adds phase keying by two-phase PSK to the output of a balanced modulator 13 corresponding to variation of the M-sequence code by an incorporated balanced modulator, etc., to perform what is called direct spectrum diffusing modulation. When autocorrelation is performed by the transmit pulses, the reception level of every chip by direct spectrum diffusing modulation in transmit pulses is all integrated, pulse by pulse, into the position of the chip when delay time is zero; and the obtained correlative peak output value is sent out to a phase detector 30 through an output line 271.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a communication system, and more particularly to a communication system with anti-jamming capability that significantly improves interference before jamming and significantly improves confidentiality of communication contents.

When communicating between a transmitting station and a high-speed moving object such as an aircraft using digital radio waves, communication is often It is also well known that communications are often subject to human interference or interception, and that Yang's efforts to improve anti-interference capabilities and improve the confidentiality of communications are well known.

Conventionally, in order to maintain this type of anti-jamming ability and confidentiality, methods have been adopted in which the information to be transmitted is digitized and then transmitted or further encrypted.

Furthermore, recently, a method of superimposing transmission data on a carrier pulse modulated by a modulation means using a spread spectrum technique and transmitting the data has been commonly used.

However, when the transmission pulses of the various transmission signal formats mentioned above are received by a high-speed moving object, they are affected by the Doppler effect, which makes stable reception as in fixed communication difficult, and in severe cases, reception becomes impossible. It is rare that fx<s. Moreover, there is a drawback in that it is almost impossible to predict and deal with the influence of the Doppler effect in advance.

This will be explained in detail as follows. In other words, in the spread spectrum technology normally used for transmission pulses in this type of communication system to provide anti-jamming capability, it is important to synchronize the spread spectrum codes on the transmitter and receiver sides; For methods normally used to acquire synchronization, for example those that assume synchronization to be acquired while sliding, the processing time becomes longer as the length becomes longer due to secrecy. Basically, it is not suitable for receiving high-speed moving objects, and other types of synchronization acquisition means are also not suitable for receiving high-speed moving objects that move at high speeds of Mach units, although there are differences in degree. Therefore, in order to ensure synchronized communication with such high-speed moving objects, it is common to process the received signal using other high-speed receiving means, such as a matched filter or a convolver.

However, even with such means, the Doppler 77 corresponding to the data bit length is
In extreme cases, correlation detection output by a matched filter or the like cannot be obtained.

As is well known, a mated filter synchronously (coherently) integrates the signal represented by each bit that makes up the data pilator at the bit with zero delay time, concentrates all the energy of the signal on this bit, and outputs it. It performs high-speed processing, so-called correlation detection, and as long as each bit of the signal is received within the processing time of this correlation detection operation, there is no problem. If a phase rotation occurs, the output will be reduced in proportion to the amount of Doppler shift obtained by the above-mentioned correlation detection, and in severe cases, the problem will occur that the output will become zero.

For example, consider a transmission pulse that has a carrier frequency of 1500 MHz and is subjected to two-phase P8K (Phase 5-shift keying) direct spread spectrum modulation.The data bits are 4.8 Kb (kilobits) at a speed of 7 seconds, One bit is converted into an M-sequence code by a PN code generator with a 10-stage shift register, that is, 2
P N (Pseu
Assuming that direct spread spectrum modulation is performed using a do No1se (pseudo-noise) code, the transmission data bit time td is expressed by the following equation (1).

td2 1/4.8Kb # 208μS...
・・・・・・・・・(1) This transmission data bit time td
The logical value 111 or 101 of the transmission data is transmitted every time.

Now, the transmission pulse has each section, which is called a chip, divided by 1/1023 by the M-sequence code, and the chip length tc is expressed by the following equation (2).

tc=208/102.3 0.203μs...
(2) That is, the transmission data bit time td is composed of 1023 chips having a chip length tc shown in equation (2). Therefore, the output of the correlation detection by the mass filter is obtained by consolidating the transmission pulses distributed corresponding to each of the 1023 chips at a delay time of zero, and this is performed for each transmission data bit time. As described above, in this matched filtering, each chip receives a phase change Δφ caused by the Doppler shift expressed by the following equation (3) and is received one after another.

Δφ=2π(-・7o)tc ・・・・・・・・・
(3) In equation (3), ■ is the relative speed between the high-speed moving object and the transmitting station, C is the radio wave propagation speed, and fo is the carrier frequency.

By substituting the above-described values into equation (3), the following equation (4) is obtained.

Δφ=6.37xlO= xv (radian)...
...(4) In equation (4), if ■ is 3 times the speed of sound, then 3 times is approximately 102 om/sec, so Δφ is approximately 6.5X10-3 radian,
The phase of the 1023rd chip is rotated by about 3785 degrees.

Although the above numerical example is based on the case where V is 3 matsuha, it is clear that even if V is 2 matsuha or 1 matsuha, an extremely large phase rotation will be brought about.

Therefore, the correlation peak output value of the matched filter obtained by receiving the transmitted pulse subjected to such a Doppler shift and detecting the correlation while correlating the signals of each m chips is 17]
It must be much smaller than X (chip level), and in extreme cases it can become zero.

Therefore, in order to use the transmission pulse for a high-speed moving object as a spread spectrum signal, the Doppler shift must be corrected by some method, or the transmission data bit time must be set within a range that does not need to be affected by the TD75 Doppler shift. It is necessary to configure a communication system that utilizes filters and the like so as not to lose their effectiveness, and also maintains the insufficient anti-jamming ability by supplementing with other signal systems.

The purpose of the present invention is to eliminate the above-mentioned drawbacks, and to provide a Doppler shift method in a communication system in which a transmission station emits a transmission pulse using digital radio waves to a high-speed moving object such as an aircraft to perform communication with anti-jamming properties. The high-speed synchronization characteristics of the matched filter are maintained by transmitting all spread spectrum transmission pulses with a transmission time width that is virtually unaffected by It is an object of the present invention to provide a communication system that can effectively eliminate the influence of the Doppler effect and significantly improve anti-jamming properties by providing a means for time hopping in accordance with a hopping pattern.

The method of the present invention is a communication method in which a transmitting station emits a transmission pulse using digital radio waves to a high-speed moving object such as an aircraft, and performs communication with the high-speed moving object with anti-jamming capability. The transmission data bit time is represented by one bit divided by a preset number of divisions as one bit of transmission data by a carrier pulse modulated by the set spread spectrum modulation method, and the transmission data is divided into two logical values, 111 and 1Ol. The transmission data is generated as one representative bit, and one bit of this transmission data is made to correspond to a preset random time hopping pattern and output as a transmission pulse while sequentially performing random time hopping for each transmission data bit time. a pulse output means, and a carrier wave signal generated in synchronization with the transmission pulse in response to the random time hopping button and uncorrelated with the transmission pulse is synthesized with the transmission pulse as a synchronization pulse with respect to the transmission pulse. a synchronization pulse output means for outputting a synchronization pulse, and a synchronization receiver 4 that receives a transmission signal including the transmission pulse and a synchronization pulse, and allows the high-speed moving object to receive the transmission pulse from the transmission station while maintaining synchronization by the synchronization pulse. It consists of a stage.

FIG. 1(5) is a block diagram showing the configuration of a transmitting section in a transmitting station, and FIG. 1(B) is a block diagram showing the configuration of a receiving section in a high-speed moving body.

The transmitter 1 shown in FIG. 1(5) includes a gate circuit 11.a random time hopping code generator 12. Balanced modulator 1
3. Carrier wave oscillator 14, two-phase PSK modulator 15, P
N Coco Generator 16. Adder 17° Phase shifter 18. Balanced modulators 19, 20, power amplifier 21 and transmitting antenna 2
2.

Further, the receiving section 2 shown in FIG. 1 (13) includes an input amplifier 2
4. Frequency mixer 252 local oscillator 26. Matched filter (~27.Matched filter (B) 28° phase shifter 2
91 phase detector 30. It is configured to include a defruter 31 and a synchronization holding circuit 32. In FIG. 1 (5), the transmission data input via the input line 111 is output from the transmitter 1 as a binary logical value 111-ta (a) corresponding to the content of the transmission data at every transmission data bit time td. The example of FIG. 1 will be explained with reference to the transmission pulse (b).

The transmission data input to the gate circuit 11 is shown in FIG. 2(a).
As shown in FIG. 1, the data is normally unipolar type data having logical values 115 or 101 corresponding to the content of the data to be transmitted at each transmission data bit time td, the number of which corresponds to the number of bits of the transmission data. Figure 2 (a
), this is converted to 111. by the logical value POI. Logical value P
It is expressed as 101 etc. due to O2.

The random time hopping code generator 12 generates an M sequence of 2n-1 bits by combining an n-stage shift register and a logic gate circuit for feeding back pulses output from the n-stage 77 register to the input side of the shift register. A known PN code generation circuit that synthesizes and outputs codes outputs an M-sequence code of 2n-1 pits of a preset random time sequence over the transmission data bit quim td, and has an output gate pattern that presets these codes. Transmit data bit time td through output gate circuit
One pulse is generated randomly in time. At this time, the M-sequence code supplied to the output gate circuit has unipolar data changed to a dipolar waveform by the built-in waveform conversion circuit, and therefore, as a random time hopping code, the transmission data bit time t
Time t1 shown in FIG. 2(b) when d is divided into 2n-1 pieces.
The gate circuit 11 gates the input transmission data using such a random time hopping code, so that when the transmission data takes the logical value 111, the logic value of the M-sequence code is 111. time width t
1 positive pulse and, in the case of logical value 10 I, a negative pulse of time width t1, respectively, at the transmission data bit time td.
It outputs it as a randomly hopping pulse within, and sends it to the balanced modulator 13 via an output line 112.

A carrier wave with a frequency of 10 is input to the balanced modulation circuit 13 from the carrier wave generator 14 via the output line 141 together with the output of the gate circuit 11, and the balanced modulation of these two is performed and pulsed with the output pulse of the gate circuit 11. 2-phase P
The signal is sent to an S K (Phase Shift Keying) modulator 15. The output of the balanced modulator 13 formed in this way is positive when the transmission data has a logical value of 111, and negative when the logical value is 101, and is randomly distributed in the time width t1 and during the transmission data visit time td. It is a time-hopping pulsed carrier wave.

Now, the two-phase ratio PSK modulation circuit 15 inputs a predetermined random time series M-sequence code from the PN coco generator 16 via an output line 161, and changes the sign of the M-sequence code using a built-in balanced modulator or the like. Corresponding to the balanced modulator 1
Phase keying using two-phase PSK as described below is added to the output of No. 3 to perform so-called direct spread spectrum modulation.

That is, the positive pulse of time width 11 outputted by the balanced modulator 13, that is, the pulse corresponding to the logical value Jl of the transmission data, has a code change corresponding to the sign change of the M sequence code generated by the PN coco generator 16. When the logical value is 11'', the carrier wave phase 77 is zero, and when the logical value is l □ l level, cis vector direct spread modulation is applied by phase keying with the carrier wave phase 77 to 180 degrees, and The negative pulse of time width t1 outputted by the balanced modulator, that is, the negative pulse corresponding to the logical value 'O1 of the transmission data, has its phase set to 1 in advance.
After shifting by 80 degrees and performing spectral direct spread modulation similar to the positive pulse, the pulse shown in Fig. 2 (b
) and send them to the adder 17 via an output line 151.

The transmission pulse P1 shown in FIG. 2(b) is generated corresponding to the transmission data PO1 shown in FIG. The pulse width is the time t1 divided by the code length 2n-1=p, and over this pulse width, the code bit width t2, so-called chip The transmission pulse P1 is a carrier wave that has been subjected to the above-mentioned phase keying and direct spread spectrum modulation for each signal, and this transmission pulse P1 randomly time-hops at time t1 during the transmission data bit time td. Where this transmission pulse P1 occurs in the transmission data bit time td is limited only to the sender and receiver who know the content of random time hopping in advance.

In this way, the transmission data POI is represented by the transmission pulse P1, and the transmission data PO2 is represented by the transmission pulse P2 formed in the same way, so that the transmission pulses are made to correspond to the transmission data and are generated one after another. . Dharma friend =
The only difference between the transmitted pulse P2 and the transmitted pulse 'P'1 is that the phase of the carrier wave is 180 degrees. Thus, even when the transmitted data has a logical value of 101, only the phase of the carrier wave is 180 degrees. By generating a transmission pulse that is shifted by a degree, the transmission power can be set to 1, which is the logical value of the transmission data.
It does not change in either case of 11 or 10w, and as in the case of conventional transmission of this type, the 8/ N (Signal
to No. 1se) by 3 dB.

Now, the synchronization pulse that is output in synchronization with the transmission pulse is formed as follows.

The carrier wave output from the carrier wave oscillator 14 is also sent to a phase shifter 18, whereby the carrier wave is subjected to a phase shift of 90 degrees, and then sent to the balanced modulator 19 via an output line 181. On the other hand, the outputs of the random time hopping code generator 12 and the PN code generator 16 are sent to the balanced modulator 20, and the balanced modulator 20 balance-modulates these two signals to generate the transmission pulse shown in FIG. P.I.

A composite code is generated by removing 'j# transmission wave from P2 etc.

This is sent to the balanced modulator 19 via the output line 201.

The balance modulator 19 balances and modulates the two-man power supplied in this way, and generates synchronizing pulses P81 and PS shown in FIG. 2(C).
2 etc. and sends these to adder 17 via output line 191.

Synchronous pulses PS1h and PS2 generated in this way
are obtained as carrier waves whose phase is only 900 degrees knots from the transmission pulses Ply and P2, respectively, and are generated one after another in synchronization with the transmission pulses.

This synchronization pulse is formed in synchronization with the generation of the transmission pulse, and it is necessary to form the synchronization pulse so that the mutual correlation coefficient with the transmission pulse is as small as possible for mutual separation in the reception processing of the reception section 2. In order to make the cross-correlation coefficients smaller than each other, it is not necessarily necessary to make the spreading codes of the transmission pulse and the synchronization pulse different; for example, it is not necessary to make the spreading codes of the transmission pulse and the synchronization pulse different. There is no problem even if the code phase is shifted.

This is because the autocorrelation characteristic of the spreading code becomes extremely small even if the spread code is shifted by one bit. Focusing on this point, in this embodiment, the code sequence of the synchronization pulse and the transmission pulse are The same code sequence but with a phase shift is used, and this is combined with the transmission pulse and emitted as a transmission signal.

Now, the adder 17 adds and synthesizes the transmission pulse input via the output line 151 and the synchronization pulse input via the output line 191 in analog form, and uses this as a transmission signal via the output line 171 to the power amplifier 21. Send to.

The power amplifier 21 amplifies this transmission signal to a predetermined level, and then emits it as a transmission radio wave to the receiving section 2 via the output line 211 and the transmission antenna 22.

The transmission pulse obtained in this way can be formed as a pulse of much shorter time than the conventional method of transmitting the transmission data bit time tdi corresponding to each pit of the transmission data. The transmission time of the signal is also greatly shortened, making it easy to reduce the Doppler shift in the high-speed moving object receiving the signal to a negligible level in terms of system operation. The effect of improving interference ability can be obtained as a synergistic effect.

Regarding the improvement of the anti-jamming capability, for example, direct spread spectrum modulation within the transmitted pulse can be performed using PN with code length 31.
When it is done with a code, the transmission pulse obtained by this spreading is
5 dB, and the transmission data bit time t shown in FIG.
When the number of divisions of d is pe31, the irregularity of the transmitted pulse obtained by time hopping corresponding to this number of divisions improves the anti-jamming ability by about 15 dB, and the synergistic effect of both improves the anti-jamming ability by about 30 dB. In the embodiment shown in FIG. 1, the inside of the transmission pulse is modulated by direct spread spectrum modulation.
This may be modulated by other techniques in the spread spectrum technology, such as frequency hopping spread modulation, or a composite modulation method of direct spread spectrum modulation and frequency hopping spread modulation, in which case the synergistic effect of frequency hopping is further enhanced. can get. Regarding these spread spectrum techniques, including spectrum despreading,
For example, it is described in detail in Rin, C., Dixon, "Spread Spectrum Communication System J," published by JATEX, and many others.

On the other hand, receiving section 2 receives a transmission signal via receiving antenna 23 and supplies it to input amplifier 24, amplifies it to a predetermined level, and then sends it to frequency mixer 25 via output line 241.

The frequency mixer 25 is supplied with the local oscillation frequency output from the local oscillator 26 via an output line 261, and performs frequency mixing of both.

The local oscillation frequency output from the local oscillator 26 is the frequency f of the carrier wave of the transmitted pulse and the received pulse. The transmission signal is also shifted by the intermediate frequency, and is converted to an intermediate frequency by frequency mixing and outputted via the output line 251, and is output to the matched filter (5) 2.
7. They are respectively supplied to matched filters (B) 28.

These mated filters perform spectral despreading at high speed isometrically while undergoing direct spread spectrum modulation and receiving the transmit signal (synchronous/ It is configured as a matched filter with correlation detection characteristics to perform correlation detection and obtain the diffused energy as a correlation peak output concentrated at one point, and also has a matched filter (B).
Reference numeral 28 is constructed as a matched filter having correlation detection characteristics for synchronizing pulses.

Both the transmission pulse and the synchronization pulse are input to the matched filter (A) 27, but only the power whose carrier wave phase is shifted by 90 degrees from the transmission pulse is obtained.

The autocorrelation due to the transmitted pulse is calculated as follows for each transmitted pulse input:
An autocorrelation process is performed in which all received levels for each chip subjected to direct spread spectrum modulation in the transmitted pulse are integrated to the chip position at zero delay time, and the obtained correlation peak output value is phase detected via the output line 271. vessel 30
Send to.

As shown in FIG. 2(b), the transmission pulse is sent out at a time t1 that is sufficiently small compared to the transmission data bit time td, and therefore it is clear that the high-speed processing time of the matched filter (A) 27 is sufficient to process it. It is.

In exactly the same way, the matched filter (B) 28 obtains the correlated peak output of the synchronization pulse and sends it to the phase shifter 29 via the output line 281, thereby changing the phase of the carrier wave in the transmitter 1. The phase is shifted by 7 feet to cancel out the added 90 degree phase shift to make it the same phase as the carrier frequency of the transmitted pulse P1, etc., and this is sent to the phase detector 3 via the output line 291.
In this way, the transmission pulse input to the phase detector 30 and the matched filter output of the synchronization pulse are in phase with each other, and the effects of damping are exactly the same, so if the synchronization pulse is transmitted As long as the pulse is stably detected along with the pulse, a stable synchronous detection output can be obtained by the phase detector 30, and when the transmitted data has a logical value of 11, the pulse is of positive polarity, and when the transmitted data is the logical value of 101, the pulse is of negative polarity. It is not outputted as a demodulated transmission data pulse, but is sent out to the output line 301 as a demodulated transmission data pulse.

The defrouter 31 is a circuit for initial synchronization acquisition. For example, when only the synchronization pulse is transmitted as a preamplifier, the output of the matched filter (B) 28 is input to the envelope detector 311, thereby detecting the envelope. The tapped delay line 312 receives N synchronized signals that are transmitted in the same random time hopping pattern as the transmission signal, and its output is sent through the output line 31 to the tapped delay line 312. N output tabs 3121-1, 3121-2, given a delay amount corresponding to the time interval of the pulses.
3121-N, and when the output from the envelope detector 311 is due to the synchronization pulse input in the random time hopping pattern described above, the envelope detection output due to the synchronization pulse input through each of these output taps is They are output at the same time after being time corrected so that there is no time difference between them.

Therefore, the output tap 3121 of the tapped delay line 312
-2 gives the output tab 3121-1 for the envelope detection output of the first synchronization pulse (start synchronization using a random time hopping pattern) (delay time for the transmission time interval between the first synchronization pulse and the second synchronization pulse, and the same goes for the rest) The setting is such that a delay time corresponding to the time interval of the next synchronous pulse is given from output tab 3121-N with zero delay amount to the envelope detection output of the Nth synchronous pulse.

Therefore, when synchronization pulses are input according to a preset random time pattern, the envelope detection output is the maximum due to the tapped delay line 312, and the envelope detection output due to the number of synchronization pulses equivalent to one week of random time hopping is reached. It will be output without any time difference.

How many of the N envelope detection outputs of the N synchronization pulses output from the tapped delay line 312 are output for one week of synchronization pulses is determined by the synchronization judgment as described below. The SZN ratio in the detection of the synchronization pulse,
It can be set in advance, taking into consideration the determination accuracy and the like.

When synchronization pulses are input, this addition amount becomes a level corresponding to the number of input synchronization pulses, and the adder 313
Next, this level is applied to the built-in Schmitt trigger circuit, which uses a preset judgment level as the trigger threshold (sreso hold level), and when it exceeds this threshold, it is determined to be a synchronization pulse. The output of the trigger circuit is sent as a synchronization determination signal to the envelope detector 311 via the output line 3131, but in order to compensate for the time required for output processing, this synchronization determination signal is shifted by the transmission data bit td and output. That's what I do. When the envelope detector 311 receives this synchronization determination signal, it performs envelope detection on the input synchronization pulse and sends it to the multiplier of the synchronization holding circuit 32 via an output line 3112. Synchronization holding circuit 32 via
The random time hopping code generator 322 is also sent to the random time hopping code generator 322 .

The synchronization holding circuit 32 includes a multiplier 321. Random time hopping code generator 322. VCO (Voltage
Control 0scillato? )323.

Pulse converter 324 and L P F (Low Pa
5sFilter) 325 f-, receiving the envelope detection output of the synchronization pulse determined to be synchronized by the defruter 31 to the multiplier 321, and receiving the synchronization determination signal to the random time hopping code generator 322,
This is to maintain the synchronization of the synchronization pulses determined to be synchronized by the defruter 31.

VCO 323 generates a clock signal that sets the transmission time t1 of the synchronization pulse and transmission pulse shown in FIG. 2, and sends this clock signal to random time hopping code generator 322 via output line 3231.

The random time hopping code generator 322 generates a random time hopping code that is the same as the random time hopping code of the synchronous clock according to the input clock pulse, and sends it to the output line 3221. is connected to the adder 314 of the defruter 31 via the output line 3141.
The conversion starts when a synchronization determination signal is received from , that is, when the synchronization pulse confirms the synchronization determination, and is converted into this random time hopping code and synchronization maintenance ping code (a) and the conversion obtained by converting this random time hopping code. Random time hopping code (b)
FIG. 2 is a diagram showing the characteristics of a random time-converted random time hopping code.

From the random time hopping code generator 322, a random time voice tubbing code with a time width t1 as shown in FIG. 3(a) is outputted via an output line 3221, one for each transmission data bit time shown in FIG. A preset random time hopping pattern is output, and this is a code sequence that determines the timing of generation of the transmission pulse and synchronization pulse shown in FIGS. 2(b) and 2(C).

This random time hopping code is used by pulse converter 3
24 and is converted into the converted random time hopping code Ph shown in FIG. 3(b), which converts the random time hopping code Ph into a duty
) is converted into a pair of positive and negative pulses of 50%,
Pulse converter 324 sends this converted random time hopping code to multiplier 321 via output line 3241.

Multiplier 321 multiplies this converted random time hopping code by the envelope detection output and sends the result to I, PF 325 via output line 3211. In this case, if the phases of the two inputs supplied to the multiplier 321 match, the output of the LPF 325 will be zero, and the output of the random time hopping code generator 322 will be output as is, but the converted random time hopping code will be When the phase is behind the envelope detection output, a positive output corresponding to the delay is generated, and when the phase is ahead, a negative output corresponding to the advance is generated, and this output is output from the output line. 3251 to VC0323, and uses this as a control voltage to control and change the oscillation frequency of VC0323, and frequency matching of VC0323 is performed by this feedback control loop until the output of LPF 325 becomes zero.

In this way, it is sent out on output line 3221 while always maintaining the correct random time hopping code and thus the synchronization determined by the defruter.

The transmission data output from the phase detector 30 is
The synchronization was maintained/Damn Time Ho! The shipping code is used as a synchronization clock to be read one after another, and in this way, accurate synchronization with the transmitting section 1 is maintained, and reception processing for the transmitted data is executed.

The present invention is based on a communication method using a digital communication format that requires anti-jamming capability between a high-speed mobile object such as a starvation aircraft and a transmitting station, by dividing the transmission data bit time of the digital format to be transmitted by a preset number of divisions. The unit obtained by dividing is one bit of the transmission pulse, and this makes it representative of one bit in the transmission data bit time, and the transmission pulse is subjected to spread spectrum modulation, and this transmission pulse is divided into the time frame of the transmission bit time. In addition, the influence of the Doppler 7ft can be greatly suppressed by means of time hopping inside the station, transmitting it from the ground station along with a synchronization pulse that is synchronized with this transmission pulse, and receiving this transmission pulse synchronously by a high-speed moving object. The basic feature is that the anti-jamming ability is significantly improved, and various modifications of the embodiment of the present invention shown in FIG. 1 are conceivable.

For example, in the embodiment shown in FIG. 1, the transmission pulse generated in the transmitting section 1 uses a combination of direct spread spectrum modulation; It is obvious that other spread spectrum modulation, such as composite modulation with

Furthermore, in this embodiment, the synchronization pulse uses the carrier wave of the transmission pulse shifted by 90 degrees in phase, but the carrier wave of this synchronization pulse has a negligible correlation with the carrier wave of the transmission pulse. It is clear that the same implementation is possible even if the signal is formed using the code modulation method.

Furthermore, in the receiving section 2, matched filters are used to despread and demodulate the received transmission pulses and synchronization pulses, and these matched filters have a Doppler shift of approximately zero, and each transmit pulse and synchronization pulse are known in advance.
Although it is equipped with a fixed correlation detection characteristic that corresponds to the characteristics of the synchronization pulse, it is also possible to further eliminate the influence of the Doppler shift, which exists slightly, to increase the synchronization reception ability. Filter ('B)
A Doppler frequency correction circuit or the like is provided on the output side of the 28, and other autocorrelation detection means, such as a convolver, etc., which operates by following the Doppler frequency correction circuit using information about the amount of Doppler correction obtained thereby, can be used. It is obvious that this can be easily implemented by a method such as taking an autocorrelation between the received signal and a temporally conjugate reference signal.

Furthermore, it is clear that both the defrouter 31 and the synchronization holding circuit 32 can be replaced with other known circuits having the same function as these and can be implemented in substantially the same manner.

In this embodiment, the operating state in which the transmitting unit and the receiving unit are each set is explained as an example, but a desired plurality of transmissions and receptions can be performed using mutually different hopping patterns at the transmission data bit time td shown in FIG. It is obvious that the transmission and reception density can be easily implemented by implementing the above during the same transmission bit time to increase the transmission and reception density, and all of the above can be easily implemented without jeopardizing the gist of the present invention.

As explained above, according to the present invention, in a communication system that requires anti-jamming properties, the transmitted data bits are represented by a transmitted pulse with a time width in which 1 bit is a unit obtained by dividing the transmitted data bits by a preset number. In addition, this transmission pulse is assumed to be spread spectrum modulated, time frame A of the transmission bit time is time-hopped in a preset hopping turn, and is transmitted together with a synchronization pulse that is synchronized with this transmission pulse. By providing a means of synchronously receiving these signals, a communication system has been created that can significantly reduce the influence of Doppler shift to the point where it can be ignored operationally, significantly improve anti-jamming properties, and significantly improve transmission and reception density. There is an effect that it can be done.

The transmission data (a) in the embodiment shown in FIG. 1 (5) is transmitted using random data showing the characteristics of the random time hopping code (a) and the converted random time hopping code (b) in the embodiment shown in FIG. 1 (B). FIG. 3 is a characteristic diagram of a time-converted random time hopping code.

1... Transmitting section, 2... Receiving section, 11.
...Gate circuit, 12 ... Random time hopping code generator, 13 ... Balanced modulator, 14 ... Carrier wave oscillator, 15 ...・
2-phase P8 modulator, 16...PN code generator, 17...adder, 18...phase shifter,
19.20...Balanced modulator, 21...
Power amplifier, 22... Transmission antenna, 23...
...Receiving antenna, 24...Input amplifier,
25... Frequency mixer, 26... Local oscillator, 27... Matched filter (5), 2
8...Matched filter (B), 29...
... Phase shifter, 30 ... Phase detector, 31 ...
... Defruter, 32 ... Synchronization holding circuit, 31
1... Envelope detector, 312... Delay line with tap, 313... Adder, 321...
... Multiplier, 322 ... Random time hopping code generator, 323 ... VCo, 32
4...Pulse converter, 325...LP
F.

7th block

Claims (1)

[Claims] In a communication system in which a transmitting station emits a transmission pulse using digital radio waves to a high-speed moving object such as an aircraft and performs communication with the high-speed moving object with anti-jamming capability,
The transmission data bit time is divided by a preset number of divisions as one bit of transmission data using a carrier pulse modulated by a preset spread spectrum modulation method.
In addition, the transmission data is generated as one bit with binary logical values 111 and 101 as the representative, and one bit of this transmission data is transmitted in correspondence with a preset random time hopping pattern. a transmission pulse output means for outputting a transmission pulse while performing random time hopping one after another for each data bit time; synchronization pulse output means for combining a carrier wave pulse with the transmission pulse as a synchronization pulse for the transmission pulse and outputting the result; a synchronization pulse output means for receiving a transmission signal including the transmission pulse and the synchronization pulse; 1. A communication system characterized in that a high-speed moving object is provided with synchronous reception means for receiving transmission pulses from the transmission station.