JPS58163286A - Controller for drive of induction motor - Google Patents

Abstract

PURPOSE:To prevent the variation of reactive power at a receiving terminal by simple constitution by controlling the amplitude value and supplied frequency of currents supplied to the induction motor by control signals from a means controlling reactive power at the receiving terminal and the generating torque control means of the induction motor. CONSTITUTION:Three-phase AC voltage and three-phase AC currents are detected by a transformer PTS and a current transformer CTS set up at the receiving terminal, and inputted to a reactive-power computing element VAR. The reactive-power computing element VAR obtains the instantaneous value of reactive power at the receiving terminal. A comparator C1 compares the instantaneous value of the reactive power and command value, transmits the outputs to a comparator C2 through a control compensation circuit HQ(S), and compares the value with outputs from a current transformer CTO. Outputs from the comparator C2 are tansmitted to a phase control circuit PH1 through an operational amplifier K1. Outputs from a rotary pulse generator PG and a revolution-speed setter VRW are processed by a speed control compensation circuit HN(S) ring counter RC, a counter PC and a logic circuit LC, and transmitted to a gate control circuit GC.

Description

Detailed Description of the Invention tat Description of the Technical Field The present invention relates to a drive control device for an induction motor that is capable of controlling reactive power at a receiving end to an arbitrary value.

(bl Description of Prior Art) Figure 1 shows a configuration diagram of a conventional drive control device for an induction motor.

In the figure, %BU8 3-phase AC power supply wire, TR is power transformer, SS1 is separately excited converter, 88*h self-excited converter, L @ ke DC reactor, IMd induction motor body, P
G is the rotation pulse, % CT is the DC current detector, VR
w Fi rotation speed setter, C1, C.

is a comparator, HN (8) is a speed control compensation circuit, F
-V is a frequency → voltage converter, V-F is a voltage → frequency converter, FN, , FN is a function generator, PH1 is a phase control circuit of SS1, KIFi operational amplifier s Al is a pulse adjuster, pcFi counter device, PCFi ring counter, L
Cii logic circuit and GC are gate control circuits of S81.

First, the slip frequency fsl of the induction motor IM is controlled as follows.

Rotary pulse generator PG per 360° electrical angle of IM,
Generate n pulses. In other words, if the rotational frequency of IM is fm, then from PG it is l a (1・f
A pulse with a frequency of m' (Hz) is generated. In addition, the voltage-frequency converter V-F outputs the set slip frequency fsl.
When J-n-fsl (HzJ pulse is the main sin PG
It adds or subtracts the output signal of the V-F converter and the output signal of the V-F converter, and sends a pulse of WA wave number l-n (fm connection ±fal) to the next counter PC. Furthermore, I
When operating M in electric motor mode, 81 becomes an adder,
When the IM is operated in generator mode (regenerative operation mode), A1 becomes a subtracter. Counter P Ctd Generates an output signal every n counts and outputs an output signal to the next ring counter, RC (1=
clock). Therefore [RC output signal has frequency f
e=fm±fsl, resulting in a 60° energizing rectangular wave (6 pieces). The next logic circuit LC converts the 60′ energizing square wave into 120”
Gate control circuit G that converts into energizing square waves (6 pieces)
The IfC gate signal is supplied to the self-excited converter 8B via C. Therefore, the output frequency of sSz is fe=fm±fsl
becomes.

That is, since the primary frequency of the induction motor IM is fe and the rotational frequency fm, the secondary frequency fsl of the IM is fsl==fe-1m=fsl. This relationship holds true regardless of the value Vc of the rotational frequency fm.

Next, the speed control operation of the electric motor IM will be explained.

The speed setter VRω provides a speed command value ω for the electric motor IM. Further, the output signal of the one-rotation pulse generator PG is converted into an analog quantity by the frequency-voltage converter FV, and is used as the speed detection value ω. The speed command value ω and the detected value ω are compared by the comparator CI, and the deviation t
Input N=ω-ω to the control compensation circuit HN (S)・
HN (S) usually has an integral of 1! The constant deviation gN is made zero. The output of 1(N(S) is the motor I
With the torque command T1 of M, the function generator FN1. DC current command I and slip frequency command [item]
is giving. The smooth frequency command [fFl] is an analog quantity, and is converted to 1-n-t^ (Hz) via a V-F'i converter.
) is converted into a digital quantity.

On the other hand, the current detector CT detects the direct current (2), and the comparator C compares it with the current command value (5).

The deviation go=I''-I・ is amplified by an operational amplifier and inputted to the phase control circuit PH1.The phase control circuit PH controls the firing phase of the separately excited converter S81, and uses a known method. is used.

That is, when the input of PH is set to 61, the firing phase angle of SS1 is α1 times α1 = COr1gl, and the output voltage vdl of S81 becomes a value expressed by the following equation.

vdl ==k @ Va @ CO8 (1,
Proportionality constant, Vs: ml When the source voltage actual tfile is smaller than its command value 0, the deviation 1 is positive, and the above S
By increasing the DC output voltage vdl of S1, the DC current 1o is increased. Also I,)1. +2), 'e
<O9, reduce vdt, reduce io. Finally I, Ki I? It calms down.

The generated torque Te of the induction motor IM is related to the primary current amplitude value, that is, the current value IO and the slip frequency fsl. In particular, when operating with a constant slip frequency fsl, the generated torque Te is 2gI of DC current 15fF and Io.
I! It becomes proportional to , and DC series characteristics are obtained.

Also, the function generator FN1. Torque command using FN?
By setting the DC current command value IFEL and the slip-slip frequency command value (fsl) to a certain -u relationship, it is possible to give the blind motor IM a DC shunting characteristic. Here, for convenience of explanation, the general operation will be explained with fsl=-9.

When the rotational speed ω of the IM is smaller than the command value ω, the deviation εN=ω−ω becomes a positive value and the torque command is increased by T%. Correspondingly, the DC current command value 1 increases, causing the generated torque Te of the electric motor IM to increase. Therefore, the electric motor IM accelerates and the rotational speed ω approaches its command value ω. Conversely, if ω〉J.

Since g N < 0, - is decreased, and the generated torque Te of the IM is decreased. Therefore, IM decelerates and settles down to ω in ω.

Such a conventional drive control device for an induction motor has a drawback that the reactive force at the power receiving end varies greatly in conjunction with the speed control described above.

That is, the reactive power Q at the receiving end of the separately excited converter SS1
ssij, power supply voltage Vs, direct fl voltage Wt, flo,
When the ignition phase angle of 881 is α11 proportional unit, it can be expressed as follows.

Qs@=kl *Va@Ie*sin α.

Here, the proportionality constant kl and the power supply snow pressure Vs are constant values,
The DC current IO and the ignition phase angle α1 change in accordance with the speed control of the induction motor IM.

That is, when the IM rotates at low speed, the DC output chamber pressure vdl of SSl is small and the one-firing phase angle α1 is close to 90°, which is sin α1+1, but as the speed of the IM increases, vdI increases, α1 moves in the direction of 06, so M of sin αl becomes very small, reducing Qss. Moreover, when the deviation 11N between the speed command value ω and the actual speed ω becomes smaller, the above-mentioned DC ′v1 flow ■o
becomes smaller, and the reactive power Qss at the receiving end of SS1
becomes even smaller. Therefore, in the low starting speed range of IM, Qs
s becomes a large value, and as the rotational speed ω increases, s Q
When ss decreases and approaches the command value J, the DC current I
Qss becomes a negligible value due to the decrease in . In particular, since a drive motor of a rolling mill or the like performs rapid reversible operation, the reactive power Qss at the receiving end of SS1 fluctuates significantly. Fluctuations in reactive power not only increase the amount of power supply system equipment required, but also cause voltage fluctuations, which have various negative effects on electrical equipment connected to the same system.

Therefore, recently, a method has been adopted in which a reactive power compensator is connected to the power receiving end of the separately excited converter SS1 to cancel out the fluctuations in the reactive power of the SS1 and take only active power from the W source. .

This reactive power compensator is composed of, for example, a phase leading capacitor that takes a constant leading reactive power and a phase control reactor device that takes a variable lagging reactive power, and the separately excited converter S
The phase control reactor device is controlled so that the delay reactive power consumed by the phase control reactor device 81 and the delay reactive power taken by the phase control reactor device are exactly equal to the lead reactive power taken by the phase advance capacitor. The basic scorer rate at the edge can be controlled to 1.

However, this phase control reactor device requires a thyristor circuit to control the current flowing to the accelerator, and its capacity is comparable to the capacity of the separately excited converter SS1, so the equipment cost is high. It had the disadvantage of being expensive.

(C) Purpose of the Invention The present invention was created in view of the above, and provides an induction motor that reduces reactive power fluctuations at the receiving end without using a reactive power compensator such as a queen phase control reactor device. An object of the present invention is to provide a drive control device.

ldl Configuration of the Invention 1 and 2 Figures 1 and 2 are configuration diagrams showing an embodiment of a drive control device for an induction motor according to the present invention.

In the figure, Bus is a three-phase AC electric wire, CAL' is a phase advance capacitor, %Ta is a power transformer, 881 is a separately excited converter, and 8S! is the self-excited converter%, IjO is the DC reactor, IM is the induction machine body, PG is the rotating pulse generator, P
T, is a transformer, CTs, CTo Fi current transformer.

■R#-i reactive power calculation circuit, Ct-Cs are comparators.

VRq is the reactive power setting device, VRw is the rotation speed setting device, HQ (8) is the reactive power control compensation circuit, HN (8
) is the speed control compensation circuit, F-V is the frequency-voltage converter,
V-F is a voltage-frequency converter, KI is an operational amplifier, P
H, is a phase control circuit of separately excited converter SSl, 81 is a pulse adder/subtractor, PC is a counter, RC ring counter,
LC is a logic circuit, and GC is a gate control circuit for the self-exciting converter S82.

The method of controlling the slip frequency fsl of the electric chair IM is the same as described for the conventional device.

Further, the amplitude value of the DC current I, that is, the current supplied to the motor IM, is the output voltage vdl of the separately excited converter 8S1.
This is similar to the conventional device shown in FIG. 1.

The major difference between the device of the present invention and the conventional device is that in order to control the reactive power Q at the receiving end, the above ik N, 11 RI
The above-mentioned slip frequency is controlled in order to control the torque Te of the 1 & machine IM.

(e) Effect of the invention First, the transformer PTs and current transformer CT installed at the power receiving end
Three-phase AC voltage and three-phase AC current are detected by svC and input to the thermodynamic power calculator VAR. Using VAR, shift the phase of ii 3-phase AC voltage by 90 degrees, calculate the product with S* for each phase, add the three phases, and then the power receiving comparator C1 adjusts the reactive power detector Q and the reactive power setting. Command value q from device VRQ
), and the difference εQ=QQ is input to the control compensation circuit HQ (S). Hq(8)K is normally used by an integral element and is controlled to make the foot deviation IQ zero. The output IP of the control compensation circuit HQ (8) is the direct current 1. This is the command value.

As mentioned above, the reactive power Qss at the receiving end of the separately excited converter 881 is proportional to the value of the DC current .

The reactive power Q at the receiving end is the sum of the delayed reactive power Qss consumed by the SS1 and the advanced reactive power Q cap consumed by the phase advance capacitor CAP, and can be expressed as follows. However, the delay is assumed to be a positive value.

Q = Qs% - Q cap If the detected value Q is smaller than the command value Q', the deviation #Q =
Q'-Q becomes a positive value and increases the DC current command value I/f-. Since the DC current generator is controlled to be equal to its command value IFVc, the delayed reactive power Qsi of 881 increases, causing the reactive power Q at the receiving end to increase. JIP-wise Q=
Q'' and settles down. Conversely, if Q >Q'', g
Q(O) and decreases the DC current IO. Therefore, KQs
s decreases and is controlled so that Q=Q.

On the other hand, the speed control 1]Fi of the induction motor IM is performed as follows.

The digital signal from the pulse generator PG is converted into an analog quantity ω by a frequency-voltage converter FV.

This is the rotational speed detection value of the IM. The comparator Cs compares the speed command value ω from the speed setter VRw with the speed detected value ω, and the deviation tN=ω−ω is sent to the control compensation circuit HN(S
). A proportional ten-integral element is used in HN(S), and its output T% is the generated torque Te of the electric mower IM.
becomes the command value.

FIG. 3 shows the relationship between the generated torque Te and the slip frequency fslK of the induction motor IM, using the DC current IO as a parameter. That is, when fsl is positive, Te also becomes a positive value, and IM is a motor operating j1. Conversely, when fsl is negative, Te also becomes a negative value, and IMVi
Generator operation (regenerative brake operation) is performed.

Also, gfs+ is -fs(max) <fsl <' + fs(max
), the generated torque Te is equal to the slip frequency fsl
It can also be seen that it is roughly proportional to .

Therefore, the slip frequency fsl of the induction machine IM can be changed to a(m
ix) to +fs (max), the generated torque Te of the IM can be controlled.

That is, the torque command T is applied to the voltage-frequency converter V-F via a limiter circuit (not shown) (a circuit that satisfies -fs(mix) <fsl <+IA1maX)).
The rotational speed of the IM is controlled by inputting a slip frequency setting value fs''l proportional to the torque command value T''.

In the case of ω5ω, the deviation gN becomes a positive value, and the torque command T
%increase. As a result, the slip frequency set value til increases in proportion to -, and the generated torque Te of the induction machine IM increases. Therefore, IM is accelerated and the rotational speed ω approaches its command value -. vice versa.

When ω or ω, kIN<O, and the value of the T angle increases accordingly. Therefore, the slip frequency setting value fml also becomes a negative value, and the pulse adder/subtractor performs subtraction. Therefore, IM becomes a generator operation (regenerative braking operation) and decelerates, and approaches the command value ω. It is controlled so that it finally becomes ωsoω.

In this way, the device f1t of the present invention controls the reactive power Q at the power receiving end and the speed or torque of the induction motor IM, but there is mutual interference between the two control systems as described below.

First, we will discuss the influence of the speed control system on the reactive power control system.

When 1=ω, the torque command T1 is zero, so the slip frequency fsl is also controlled to be zero. IM slip frequency f
The fact that sl=o means that all of the current supplied to the IM becomes the excitation current, and the DC side voltage vdl of the self-exciting converter SS3 at this time has a value close to zero. Therefore, the DC voltage vdl of separately excited converter SSl also becomes a small value, and the firing phase angle α of 8B becomes a value close to 901.
When the sine value Final becomes 1 and the torque command - is increased to a positive value as 0ω5ω, fs
l) O and the secondary electric current of the induction machine IM increases. Therefore, since the DC side voltage vd2 of SSs increases, the direct sweat voltage vdl of 8Sl also increases accordingly. Therefore, the ignition phase angle α1 changes from 90° in the direction of OO, and its sine value 1inaI#i becomes smaller. When the slip frequency fsl of the induction machine IM is increased, the rate at which the direct at pressure vd2 of the self-excited converter SS increases is related to the rotational speed ω of the induction machine IM. That is, when the value of ω is large, K
When fsl is increased, vdl increases rapidly, but when fsl is increased when the value of O is small, 4. vdl does not become very large.

Therefore, when the value of ω is small, the influence from the speed control system on the reactive power control system is small, so here we will explain the case where the value of ω is large.

As a result of the decrease in the sine fE ginα, the reactive power Qss4 at the receiver 1' end of SS1 decreases, and Q<Q.

Therefore, in order to increase the DC current I0 so that KQ=QK, the generated torque Te of the induction machine IM and the shear frequency fsl
In addition to the increase in I, the IM is increased by the increase in I. As a result, the rotational speed ω of the IM approaches its command value ω, the torque command − and the slip frequency 131
decreases and the DC current I. returns to its original value. final [Q=Q
, ω = ω and settles down.

Even when ω is set to ω, it ultimately settles to ω=ω and Q=Q while giving a shadow to the reactive power control system.

Next, the influence I#I from the reactive power control system on the speed control system will be described.

When Q>Q, the DC current Io increases to 8B.

It was mentioned earlier that the delayed reactive power Qss increases and is controlled so that Q = Q''K.At this time, the induction-like IM
If the slip frequency fml of is a certain positive value and the generated torque Te of the fi1M and the load torque TL are balanced, ■
As o increases, the generated torque T of the induction machine IM
e increases, IM is accelerated, and ω>ω. Therefore, the torque command T~ decreases, and the IM slip frequency fss also decreases. As a result, the DC voltage vdl decreases and approaches the sine value sinαl. Therefore, Qss increases and becomes Q>Q. Therefore, this time I・ decreases and Te(Tt
, and fsl Vi increases slightly. Finally Q
It settles down to a DC current value I· and a slip frequency fsl such that =Q and ω=ω.

Even when Q"<Q, the final state KQ=ζ and ω=ω" will settle while affecting the speed control system in the same way.

In order to reduce the vibration effect caused by mutual interference between the two control systems and bring it to a steady state more quickly, the constants of the control compensation circuits HQ (8) and "j): HN (S) are set to optimal values. To be elected.

(f) Other Embodiments FIG. 4 is a block diagram showing another embodiment of the control circuit of the apparatus shown in FIG. In the figure, SQ is a square calculation circuit DIV is a divider. Other symbols refer to those in FIG.

The DC current command value ■, which is the output of the reactive power control compensation circuit HQ (8), is input to the square calculation circuit SQK.

(■・) Find 1. This is input into the denominator side of the divider DIV. The torque command value T%, which is the output of the speed control compensation circuit HN (8), is also input to the numerator input input a of the upper Akanemi divider DIV. The divider DIV calculates a/b, and the result (AH) is input to the voltage-frequency converter F-V. Quantity) It's huge.
1. As stated in 1, , -, the generated torque Te of a conduction IM
Fi current is proportional to the square of IO. Therefore, when the DC current command IO is increased by the reactive current 5 and force control, the slip frequency command S of the induction machine IM becomes as shown in Fig. 4.
i) as (t”l) = ksL−'T”/(I(1”)
2ksl: By giving it as a proportional constant, the generated torque Te of 84 aircraft IM
It is possible to control the reactive power at the power receiving end without changing the power. In other words, the 4th M control circuit is a book that reduces the influence of the reactive power control system on the speed control system. Note that when the overall frequency fsl'f changes L with speed control, the reactive power control system is affected, but the effect does not bounce back from the reactive power control system to the speed control system.
Vibration phenomena due to mutual interference become extremely small. Therefore 2
The determination of the constants of the two control system compensation circuits HQ (8) and HN (8) is extremely simple compared to the embodiment shown in FIG.

FIG. 5 is a block diagram showing another embodiment of the induction motor drive control device of the present invention.

In the figure, BUS is the electrical line of the 3-phase AC power supply, CAP is the phase advance capacitor, TRFi power transformer, CC is the 3-vessel pressure sinusoidal output cycloconverter, sIM is the induction machine body, PG is the rotating pulse generator, and PTs is the transformer. , CTs, CTtFi flow accumulator, VAft is a reactive force calculation circuit VR1t reactive wealth setter, and VRw is a rotation speed setter.

Self, CI, Cu, CV, and Cw are comparators, HQ(S) is a reactive power control compensation circuit, HN(8)ii speed control compensation circuit, LIM is an IJ i vine circuit, and F-V is a frequency-voltage converter. % Adder, PTGFi 3-phase sine wave generator, ML
u, MLv, MLw are multipliers, KU, KV, KW
is an operational amplifier, and PHv, PHv, PHw are phase control circuits of the cycloconverter CC.

The cycloconverter CC is a three-vessel pressure sinusoidal snow flow Iu.

Since Iv and Iw are supplied to the induction motor IM, current control is performed for each phase. It is described in detail in VI [Thyristor Application Handbook, edited by Shota Miyairi], W, Chapter 9, 1 Cycloconverter 1.

A simple $WC explanation of the operation of the electric 6LM is as follows. Comparator C
The deviation 6υ-IL4u is input to the phase control circuit PHu via the operational amplifier Ku. PHu
It controls the firing phase angle αυ of the i1 U-phase cycloconverter, and applies a voltage proportional to the phase control manual force gU*Ku to the U-phase armature winding 1 of the induction machine IM.

In the case of Itf)Iu, 'gu) becomes 0 and the applied voltage is increased to increase the current IU. On the contrary, i'ff, <, to
In the case of j IJ (0, the applied voltage is reduced and the current IU is decreased. In other words, the U-phase armature current IU
is controlled to be equal to the command value IW. If the command value I is given in the form of a sine wave, the actual current fiIυ flows in response to it. Armature current Iv, I of phase ■ and W phase
Similarly, w is controlled according to the command values r and xi.

CTL is a current transformer that detects the above currents IU+V and IW.

The cycloconverter CC employs a so-called natural transposition method in which a thyristor as a constituent element is commutated using a power supply voltage. Therefore, when rounding from the power supply side, the delayed reactive power Qcc is consumed, and when the conventional operation method is used, the induction i1. The reactive power varies greatly depending on the magnitude of the current supplied to M and the load conditions, and a problem similar to that explained in FIG. 1 occurs.

Therefore, returning to the explanation of the diagram tJIJ5, in this embodiment, the amplitude value Im of the sine wave current that is not supplied to the induction motor IM
By controlling the frequency fe, the reactive power Q at the receiving end and the torque Te generated by the induction motor IMo are controlled.

First, the reactive power Q at the receiving end is transformed by the transformer PTs.

Next, the comparator CIK compares the reactive power command value from the reactive power setter VRq, ie, the reactive power detected value Q, and inputs the deviation gQ-4-Q to the control compensation circuit Hq(S)&. The output of HQ (S) is the amplitude command value Im of the sinusoidal currents Iu, Iv, and Iw supplied to the induction machine IM. The amplitude value Im of the sine wave current corresponds to the DC current I shown in the configuration diagram of FIG. 2, and the reactive power Qc c on the power supply side of the cycloconverter CC is the amplitude value Im.
is proportional to. Therefore,! >Q, then gQ) becomes 0,
Increase Im and increase Qcc. The reactive power Q at the receiving end is the delayed reactive power Qc c of the cycloconverter CC.
and the sum of the leading reactive power Qcap of the phase advancing capacitor CAP as follows.

However, i! ! Take a positive value.

Q = Qcc - Q cap Since Q cap is a constant value, * If Qcc increases, Q also increases, and control is performed so that Q=Q. On the contrary, ζ
(In the case of Q) jIm decreases, and Q = Q'', which is 1
controlled as follows.

On the other hand, the rotational speed ω of the induction motor IM is controlled as follows.

First, a one-rotation pulse generator PG generates a pulse with a frequency proportional to the rotational speed of the IM, and a frequency-voltage transformer F-V obtains an analog quantity ω.

Next, the comparator CI compares the speed command value ω from the speed setter VRw with the speed detected value ω, and the deviation IN=J
-ω is input to the control compensation circuit HN (S).

The output of Hs (S) becomes the torque command value - of the conductive motor IM, and provides the slip frequency command value fSl via the limiter 1 circuit LIM.

Input phase sine wave generator PTGK. The three-phase sine wave generator PTG has a unit three-phase sine wave ψU with a frequency fe.

It generates ψV and ψW, which can be expressed by the following formula.

ψU=sin(ω@*1) ψV=sil(ωe・t2π/3) ψw=sinrωe−t+2r/3) However, ωe=2πfe fe = fm+fs1 By combining, the one-pass command value can be expressed by the linear equation is obtained.

Ax, fence.

IU = bn * stn (ωee t )★
Kaoru IV= Im * stn (ωem t-2r/3)
■ ≠ 1w= Im e stn (ωe111-1-2π/
'3) Therefore, the frequency fe of the IM primary current (armature current)
Since the rotational frequency is fm, the slip frequency fsl at this time is fsl = fe - fnl = (fm + fsl ) - fm = fsl The generated torque Te of the induction machine IM uses the amplitude value Im of the supply current as a parameter. In the case of slip control number fsl WC, characteristics similar to those shown in FIG. 3 are obtained.

Therefore, when ω〉ω, εN=ω−ω increases the positive torque command −, and the slip frequency command fsl becomes O(f
Increase within the range of sl<+fs(max).

As a result, the generated torque Te of the IM increases, accelerating the IM and increasing the speed ω to the command value -. Conversely, if ω〈ω, then 1
Since N<0, T is decreased and the generated torque Te of the IM is decreased. It is controlled so that finally ω=ω.

At this time, if the deviation εN is a large negative value, T"<, and the frequency command f:l is -f@(max)<rF
Bring it to l<:0. In this case, the IM generates a negative torque that rapidly decelerates the IM.

This speed control has the following effects on the reactive power control system.

When T''=O, fsl=fs'i=0, and all of the currents Iυ, 'IT, and IN supplied to 1M become excitation currents.

Assuming that the resistance of the armature winding is sufficiently small.

The phase of the U-phase current 11+) is delayed by 90@ with respect to the output voltage Vu of the U-phase microconverter. That is, when the value of voltage Vu is small (the firing phase angle α of the converter
Since the current IU is at its maximum (a value close to 90" for U), the reactive power on the power supply side of the converter becomes a large value.
The same applies to the W phase, and the reactive power Qcc of the cycloconverter CC becomes a large value.

Therefore, the current amplitude value Im for maintaining Q=Q becomes small.

When 'V>0 and single layer sl -fsf', l>0, the secondary current component of IM increases and the voltage Vu
On the other hand, the phase of the current ■υ shifts in the direction of O@. Then, when the voltage Vυ is large (the firing phase angle α of the converter
(U is a value close to 0°) The current Iυ becomes maximum, and the reactive power on the power supply side of the converter decreases. The same applies to the ■ phase and the W phase. Therefore, since Qcc decreases, the current amplitude value Im is increased so as to maintain Q-Q.

As a result of the increase in h, the generated torque Te of IM also increases, accelerating the induction motor IM.

Then, since Qcc decreases, Im is increased. In this case, the IM is subjected to a rounding regenerative brake that generates negative torque, causing the IM to rapidly decelerate.

Further, the influence on the speed control system due to reactive power control is similar to that explained in the second embodiment. here,
The DC current value IO shown in FIG. 2 may be replaced with the current amplitude value Im shown in FIG. 5.

(g) Effects of the Invention As described above, the present invention controls the amplitude value of the current supplied to the induction motor and the supply frequency using the control signal from the means for controlling the reactive power at the receiving end and the means for controlling the generated torque of the induction motor. By controlling the power, the reactive power at the receiving end of the power converter that supplies variable voltage and variable frequency power to the induction motor can be controlled to a constant value. In this case, the fundamental wave power factor at the receiving end can be controlled by 1.

That is, expensive equipment such as a conventional phase control reactor device can be omitted, and an economical device can be provided.

[Brief explanation of the drawing]

Fig. 1 is a block diagram of a conventional induction motor drive control device, Fig. 2 is a block diagram showing an embodiment of the induction motor drive control device of the present invention, and Fig. 3 explains the device of Fig. 11E2. FIG. 4 is a block diagram showing another embodiment of the control circuit portion of the device shown in FIG. 2, and FIG. 5 is a block diagram showing another embodiment of the device of the present invention. BtJS...Electric line of 3-phase AC power supply CAP...Phase advance capacitor TR...Power transformer SS1...Passive converter 881...Self-excited converter L0...DC reactor IM...Induction machine Main body PG...Rotating pulse generator PTs...Transformer CTs, CTo...Current transformer VAR...Reactive power calculation circuit C1-C3...Comparator VRq...Reactive power setting device Vru W... Rotation speed setter HQ (8)... Reactive power control compensation circuit HN (8)...
・Speed control compensation circuit F-■...Frequency-voltage converter V-F...Voltage-frequency converter Kl...Operation amplifier PH,...Phase control circuit A1...Pulse adder/subtractor RC...Ring counter PC...Counter GC
'...Gate control circuit LC...Logic circuit CC...
...3-vessel pressure sinusoidal output cycloconverter CTL...
Current transformer Cu, CV, Cw...Comparator LIM
...Limiter circuit A...Adder PTG...
Three-vessel pressure sinusoidal generator ML u , MLv , MLw = multiplier Ku
, Kv, Kw... operational amplifier

Claims (3)

[Claims] (1) An induction motor, a power converter interposed between the induction motor and an alternating current power source and supplying variable voltage and variable frequency power to the induction motor, and controlling reactive power at the receiving end of the cough power converter. means for controlling the generated torque of the induction motor; and means for receiving a control signal from the reactive power control means to control the amplitude value of the current to be supplied to the induction motor; and means for controlling the amplitude value of the current to be supplied to the induction motor; receive a control signal,
- A drive control device for an induction motor, comprising: means for controlling an applied frequency or a slip frequency of the conductive motor. (2) A power converter that is interposed between the induction motor and the AC power source and supplies variable voltage and variable frequency power to the induction motor, and controls reactive power at the receiving end of the power converter. means for controlling the generated torque of the induction motor; means for receiving a control signal from the pre-distribution effective power control means for controlling the amplitude value of the current to be supplied to the induction motor; and the reactive power control means. and means for receiving a control signal from the generated torque control means and controlling the total slip frequency of the induction motor. (3) The reactive power control means includes a phase advance capacitor that is connected to the receiving end of the power converter and shares a certain amount of leading reactive power, and the leading reactive power and the delay that is shared by the pre-distribution power converter. 3. The drive control device for an induction motor according to claim 1, wherein the reactive power at the power receiving end is controlled so that the sum of the reactive power and the reactive power reaches zero.