JPH11261445A - Adaptive type spread spectrum receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To allow a receiver to operate satisfactorily even when the timing cycle of a desired wave is incomplete or when there is a multi path element of a small delay time difference. SOLUTION: ADS-CDMA-receiver signal sampled at a constant interval of time is supplied to plural matching filter 22-1 and 22-2 and plural quadrature filters 23-1 and 23-2. The matching filters 22-1 and 22-2 determine a filter coefficient regarding the matching filter 22-1 so that the matching filter 22-2 matches the spreading code of a desired wave distorted only for a quarter tip cycle(TC). Also, the spreading code of the quadrature filters 23-1 and 23-2 is made orthogonal with the spreading code of the desired wave delayed by 0 and Tc/4. Outputs of these filters are multiplied to by a weight coefficient, added by an adder and the weight coefficient is set so that their average power becomes minimum under a restricting condition.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a receiver used in a direct spread code division multiple access system in spread spectrum communication.

[0002]

2. Description of the Related Art In recent years, various spread spectrum systems have been studied for effective use of frequencies in digital mobile communication (MK Simon, JK Om).
ura, R .; A. Scholtz and B.S. K. L
Evitt, “SpreadSpectrum Co.
mmmunication ”, Computer Sci
ence Press, 1985). In particular, CDMA (Code Division Multi) using a direct sequence (Direct Sequence: DS) method.
Since the ple Access) method has a relatively simple configuration, a practical method has been studied.

In the DS-CDMA system, a plurality of users (users) use the same carrier frequency at the same time.
Each user uses a different spreading code, but since these spreading codes have a correlation with each other, components of other users are mixed into the despread signal even when the despreading is performed with the spreading code of the desired wave. Therefore, when the number of other users is large, the level of the interference wave component mixed into the despread signal increases, and the transmission characteristics deteriorate significantly. This deterioration becomes more and more problematic when the reception level of another user becomes higher than the reception level of the desired wave. Therefore, it is conceivable to control the transmission power of each user to make the level at the reception point of each user constant, but it is very difficult to completely perform this transmission power control. The deterioration of the transmission characteristics due to the cross-correlation between spreading codes can be solved by adding an interference canceller to the receiver.

A conventional interference canceller (K. Fukaw)
a, and H .; Suzuki, "Orthogona
licensing Matched Filter (OM
F) detection for DS-CDMA m
obile radio systems ”, IEEE
Globecom 1994, pp. 139-143. 385-38
9, Nov. 1994. 2) shows the basic configuration of FIG. This receiver operates as follows. A received signal composed of the in-phase component amplitude and the quadrature component amplitude of the received wave is input from the input terminal 11. The sampler 12 samples the received signal at regular intervals and outputs the sampled signal to the terminal 13.
The signal extracting means 14 receives the sampled signal, performs despreading and linear combination operations, and outputs the combined signal to the terminal 15.
The demodulation means 16 demodulates the synthesized signal and outputs a determination signal to the terminal 17. The timing control means 18 controls the operation timing of each means.

Next, FIG. 3 shows a configuration example in which the signal extracting means 14 is a cascade connection of despreading means and linear combining means. In the figure, the spreading factor is set to 4 and the number of users using the same frequency is set to 4 to simplify the explanation. First, a sampling signal is input from the input terminal 13. The despreading means 21 includes a matched filter 22 and three orthogonal filters 23
-1 to 23-3. The matched filter 22 uses a spread code of a desired wave, and the orthogonal filters 23-1 to 23-
Reference numeral 3 uses a spreading code orthogonal to the spreading code of the desired wave and orthogonal to each other. Matched filter 22 and orthogonal filter 23
Sampled signal in -1～23-3 and performs a correlation operation between the spread code and outputs the despread signal _{x 1 (i) ~x 4 (} i).

The linear synthesizing means 25 includes multipliers 26-1 and 26-2.
6-4 and an adder 27. The despread signals x ₁ (i) to x ₄ (i) are multiplied by weighting coefficients W _{1 to} W ₄ , respectively, and added to generate a composite signal. , Output terminal 1
Output from 5 The coefficient control unit 28 receives a plurality of despread signals x ₁ (i) to x ₄ (i) and the combined signal as inputs, and obtains by an algorithm that minimizes the average power of the combined signal under the constraint of the weighting coefficient. The weighted coefficients W _{1 to} W ₄ are output. Matching filter 22 and orthogonal filters 23-1 and 23-2
3-3 can be replaced with a correlator, and the same applies to a matched filter and a quadrature filter described below.

If the optimum value of the four-dimensional weighting coefficient vector at this time is Wo = [Wo ₁ Wo ₂ Wo ₃ Wo ₄ ] ^T , then Wo = αR ⁻¹ T (1). Here, α is a certain scalar value, R is a correlation matrix of a 4 × 4 despread signal, and T is a four-dimensional steering vector. R is a despread signal X (i) = [x ₁ (i) x ₂ (i) x
₃ (i) with ^{_{x 4 (i)] T R}} = <X (i) X H (i)> to become like (2). Here, i is the time in units of the symbol period T, and Wo _j is the optimum value of W _j (j = 1, 2, 3,
4), x _j (i) is the despread signal at time i in the j-th filter (matched filter or orthogonal filter) of the despreading means, ^T is the transposed matrix, ^H is the complex conjugate transposed matrix, and <> is the set average. This R can be approximated as follows:

R = [X (1) X ^H (1) + X (2) X ^H (2) +... + X (N _t ) X ^H (N _t )] / N _t (3) where N _t is extremely large Is a large natural number. The steering vector T is in this case T = [1 0 0 0] ^T (4).

The control of the weighting coefficient is performed so that the signal level of the desired wave included in the composite signal is kept constant. Since the spreading code of the quadrature filter 23-1～23-3 is orthogonal to spreading codes of the desired wave, x ₂ (i) ~x to ₄ (i) is not included signal component of a desired wave. Taking this into consideration, the above constraint condition of the weighted coefficient is expressed by W ^H T = 1 (5). The above α is determined so that Wo satisfies this constraint.

Various algorithms for obtaining the optimum value Wo of the weighting coefficient vector can be considered. As a simple method, the method of Frost, which is an LMS with constraint conditions (Frost, OL, "Analysis")
for linearly constrained
adaptive array processin
g ", Proc. IEEE, vol. 60, No. 8,
PP. 926-935, August 1972).

[0011]

[Problems to be solved by the invention]
Incomplete synchronization or small delay time difference
If there is a multipath component of the wave, the above orthogonal filter 23
Output signals x of -1 to 23-3_Two(i) -x_Four(i)
No. component leaks. This leaked desired signal component is
Output signal x of matched filter 22₁Hope signal included in (i)
There is a correlation with the number component. Therefore, the constraint expressed by equation (5)
Weighted to minimize the average power of the synthesized signal under conditions
By controlling the coefficient, x₁The desired signal component included in (i)
x_Two(i) -x _FourCanceled by the desired signal component leaked into (i)
The average power of the desired signal component contained in the composite signal is reduced.
The error rate characteristic is greatly degraded.

In order to suppress this deterioration, it is necessary to suppress the signal component of the desired wave from leaking into the output signal of the orthogonal filter group. SUMMARY OF THE INVENTION It is an object of the present invention to provide an adaptive spread spectrum receiver that operates well even when the timing synchronization of a desired wave is incomplete or there is a multipath component of the desired wave having a small delay time difference.

[0013]

The adaptive spread spectrum receiver comprises: (1) sampling means for sampling a received signal at regular intervals to output a sampled signal; and (2) despreading with the sampled signal as input. And (3) demodulation means for demodulating the synthesized signal and outputting a judgment signal, and (4) timing control means for controlling the operation timing of each means. In particular, in the present invention, the signal extracting means, which is a central part of these means, uses (5) a spread signal of a plurality of delay-shifted desired waves and a plurality of codes orthogonal to the sampled signal. Despreading means for despreading and generating a despread signal;
(6) a linear combining means for generating a combined signal by multiplying and combining the despread signal by a weighting coefficient, and (7) taking the despread signal and the combined signal as inputs, and obtaining an average power of the combined signal under a constraint condition. And a coefficient control means for outputting a weighting coefficient obtained by an algorithm for minimizing. Operation The basic operation of the present invention is as follows. (1) The sampling means samples the received signal at regular time intervals and outputs a sampled signal. (2) The signal extraction means takes the sampled signal as input, performs despreading and linear synthesis operations, and converts the synthesized signal. (3) The demodulation means demodulates the synthesized signal to output a decision signal. (4) The timing control means controls the operation timing of each means. The synthesizing means and the coefficient control means particularly operate as follows. (5) The despreading means despreads the sampled signal by using a plurality of delay-shifted desired-wave spreading codes and a plurality of codes orthogonal thereto, and outputs a despread signal, and The synthesis means is
The despread signal is multiplied by a weighting coefficient and synthesized to generate and output a synthesized signal. (7) The coefficient control means receives the despread signal and the synthesized signal as inputs, and obtains an average power of the synthesized signal under constraint. Estimate and output the weighting coefficient obtained by the algorithm for minimizing.

The difference from the prior art is that the despreading means performs despreading using a plurality of spread codes of the delay-shifted desired wave.

[0015]

DESCRIPTION OF THE PREFERRED EMBODIMENTS In the embodiment of the present invention, a particularly characteristic part is a signal extracting means. In FIG. 1, the components corresponding to those in FIG. The difference from the conventional configuration shown in FIG. 3 is that there are a plurality (two in the figure) of matched filters that use the spreading code of the desired wave in the despreading means 21. In this figure, the matched filters 22-1 and 22-2 use spread codes of the delay-shifted desired waves. For example, the timing filters and delay times from 0 to Tc / 4 (Tc is the chip period of the spread code) are used. In order to deal with this, 0 is set for each of the matched filters 22-1 and 22-2.
And the spreading code of the desired wave delayed and shifted by Tc / 4. In order to generate a spread code of a desired wave delayed and shifted to a value smaller than Tc such as Tc / 4, a spread code for each Tc is filtered by a filter such as a cosine roll-off to obtain an interpolated value between samples. . The interpolation value corresponding to the delay shift value may be used as the tap coefficient of the matched filter. The desired wave delayed from 0 to Tc / 4 is 0 and Tc
It can be approximated as a linear combination of the desired wave spreading code shifted by / 4 delay. Therefore, if the spreading codes of the orthogonal filters 23-1 to 23-2 are set to be orthogonal to 0 and the desired wave spreading code shifted by Tc / 4 delay, the signal component of the desired wave leaks into the output signal of the orthogonal filter group. Disappears. As a result,
The desired wave signal components included in the matched filters 22-1 and 22-2 are not canceled out, and the deterioration of the error rate characteristic can be suppressed.

In this configuration, a constraint different from that of the prior art must be imposed on the control of the weighting coefficient. The matched filters 22-1 and 22-2 include the signal component of the desired wave, and are not included in the orthogonal filters 23-1 and 23-2. In view of this, constraint weighting factor mentioned above _{^{W H T 1 = 1 (6}} ) W H T 2 = 1 (7) , and the respective steering vector T ₁ and T ₂ are T ₁ = [1 0 0 0] ^T (8) T ₂ = [0 1 0 0] ^T (9) Constraint equation (6) and (7) is equivalent to fixing the weighting coefficients W ₁ and W ₂ with respect to the output signal of the matched filter 22-1 and 22-2 to 1.

In this configuration, the number of matched filters of the despreading means is two, but can be extended to three or more.
Further, it is easily conceivable to extend to diversity reception and RAKE reception. In the above description, despreading is performed using a filter. However, a sliding correlator may be used that correlates an input signal with a despread code and synchronizes the despread code with the input signal.

[0018]

As described above, the present invention operates satisfactorily even when the timing synchronization of the desired wave is incomplete or when there is a multipath component of the desired wave having a small delay time difference. It is effective to use the same carrier frequency in a wireless system shared by many users. In a mobile communication, since a user's call changes in a time series, such information is automatically extracted from a received wave, and it is particularly effective for a receiver adaptive to the change.

[Brief description of the drawings]

FIG. 1 is a diagram showing an example of a functional configuration of a signal extracting unit according to an embodiment of the present invention.

FIG. 2 is a block diagram showing a general functional configuration of an adaptive spread spectrum receiver.

FIG. 3 is a block diagram showing a functional configuration of a conventional signal extracting unit in FIG. 2;

Claims (1)

[Claims] 1. Sampling means for sampling a received signal at regular time intervals and outputting a sampled signal, and signal extracting means for receiving the sampled signal as input, performing despreading and linear synthesizing operations, and outputting a synthesized signal. An adaptive spread spectrum receiver comprising: a demodulation means for demodulating the synthesized signal to output a determination signal; and a timing control means for controlling operation timing of the signal extraction means and the demodulation means. Means for despreading the sampled signal by using a plurality of spread codes of a desired wave delayed and shifted by one chip or less and a plurality of codes orthogonal thereto, respectively, to generate a plurality of despread signals; A spreading means, a linear synthesis means for generating and outputting the synthesized signal by multiplying and synthesizing a plurality of the despread signals by a weighting coefficient, and And a coefficient control means for outputting the weighting coefficient obtained by an algorithm for minimizing the average power of the synthesized signal under a constraint condition with the synthesized signal as an input. Receiving machine.