JPH1098333A - Antenna for mobile radio - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an antenna for mobile radio by saving spaces, such as a vertical size while maintaining a prescribed characteristic. SOLUTION: A radiator of a first antenna 32 at an upper stage consists of an internal conductor 7a and a metallic pipe 10 within a hollow and non- conductive radome, and the radiator of a second antenna 34 at a lower stage consists of metallic pipes 14 and 15 within the same radome 2. Two PEs 31 are offset-arranged, lower than a power-feeding point 9 nearly in parallel with the first antenna 32, and two PFs 33 are offset-arranged higher than a power feeding point 13 nearly in parallel with the second antenna 34. Thereby a tilting angle is freely set in the range of about ±10 deg. by adjusting a PE length and the external size of the antenna is reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an antenna mainly used in mobile radio, and more particularly to a mobile radio antenna suitable for use in a base station.

[0002]

2. Description of the Related Art In recent years, mobile radios such as portable telephones and personal handy phone systems (hereinafter abbreviated as PHS) have been actively used. Since many stations must be installed, space saving of base stations is strongly required.

As a condition required for an antenna used for a mobile radio base station, it is preferable that the position of the mobile station cannot be specified, so that the directivity in the horizontal plane is as low as possible. It is desirable that the tilt angle can be set in a range of 0 degree or a small range and the gain is high. FIG. 24 shows a conventional example of such a base station antenna for mobile radio. FIG. 24 shows a conventional two-stage collinear array antenna. In FIG. 24, a non-conductive radome 115 for housing the antenna is a radome canopy 115a, a radome bottom lid 1
15b, a radome side wall 115c, and a coaxial power supply line 112 between the radome canopy 115a and the radome bottom lid 115b.
Is stretched. The first dipole antenna 109 is
An inner conductor 112a above the coaxial feed line 112 and a metal pipe 113 held by spacers 114 made of an insulating material such as Teflon and supplied with power from the outer conductor of the coaxial feed line 112.
And the second dipole antenna 110
Is formed by vertically symmetrically arranging a metal pipe 113 held below by a spacer 114 across an annular slit 112X provided in the outer conductor of the coaxial feed line 112, and the point of the annular slit 112X Powered by

In the above configuration, the first and second dipole antennas 109 and 110 are vertical dipole antennas, respectively, and have a substantially circular directivity in a horizontal plane, and are stacked in the vertical direction. The directivity is sharp and the required gain is realized. In this configuration, the tilt angle is determined by the feed point interval between the first dipole antenna 109 and the second dipole antenna 110. When tilting downward (-Z direction), the interval is narrowed and upward (+ Z direction). When tilting to, set by widening the interval.

[0005]

The conventional antenna as described above has the characteristics of a mobile radio base station antenna, but has a large vertical dimension. For example, for a frequency of 1.9 GHz, the total length of the antenna in the vertical direction is required to be approximately 177 mm. In particular, when the antenna is configured as a vertical diversity antenna, for example, if the distance between the upper and lower antennas is 2.5λ (395 mm), the total length of the antenna is approximately In addition, when the beam is tilted upward, the total length of the antenna is further increased (at +10 degrees, 177 mm becomes 191 mm), and there is a problem that the installation location is restricted.

SUMMARY OF THE INVENTION An object of the present invention is to provide a mobile radio antenna capable of saving space while maintaining desired characteristics in consideration of the above-mentioned problems of the conventional antenna.

[0007]

A vertical dipole antenna having a feed point, a plurality of parasitic elements, and an insulator, each of the plurality of parasitic elements being substantially parallel to the vertical dipole antenna, and Support means for supporting the vertical dipole antenna and the plurality of parasitic elements such that the height of the center of each of the plurality of parasitic elements differs from the height of the feeding point by a predetermined distance, and Each of the
The mobile wireless antenna is a linear or band-shaped conductive element having an electrical length within a range of ± 10% with respect to a used wavelength.

According to the present invention, a plurality of parasitic elements having a length within a range of ± 10% with respect to a used frequency are arranged so that the center thereof is different from the feed point of the vertical dipole antenna by a predetermined distance. By doing so, it is omnidirectional in the horizontal plane, increases the gain in the vertical direction with respect to the antenna axis in the vertical plane without forming a collinear array configuration, and has a small vertical height and a tilt angle of 0 degree or A mobile radio antenna that can be set in a small range can be configured, and the antenna height can be reduced even in the case of vertical diversity.

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A vertical dipole antenna having a feed point, a plurality of parasitic elements, and an insulator, each of the plurality of parasitic elements being substantially parallel to the vertical dipole antenna and having a plurality of parasitic elements. Support means for supporting a vertical dipole antenna and a plurality of parasitic elements, so that the height of the center of each of the parasitic elements differs from the height of the feeding point by a predetermined distance, each of the plurality of parasitic elements Is a mobile wireless antenna which is a linear or band-shaped conductive element having an electrical length within a range of ± 10% with respect to a used wavelength.

According to the present invention, a mobile radio antenna which is omnidirectional in a horizontal plane, has a small tilt angle in a vertical plane with respect to a direction perpendicular to the antenna axis, and has a small antenna outer shape is provided. It is possible to increase the directivity gain in a vertical plane without using a collinear array configuration.

In the above invention, at least two sets of the vertical dipole antenna and the plurality of parasitic elements can be configured to overlap in the vertical direction. Even in such a case, the features of the above invention can be maintained.

An embodiment of a mobile radio antenna according to the present invention will be described below with reference to the drawings. (Embodiment 1) FIG. 1 is a transparent perspective view of a mobile radio antenna according to Embodiment 1 of the present invention, and FIG. 2 is a sectional view thereof. In FIG. 1, an antenna holding bracket 1 holds a hollow, non-conductive radome 2. The flexible coaxial feeders 5 and 6 are connected to the rigid coaxial feeders 7 and 8 in the antenna by rigid-flexible converters 5a and 6a, respectively, in the antenna holder 1. First
The antenna 32 includes an inner conductor 7a having a length of approximately λ / 4 at the upper end of the coaxial feed line 7 in the antenna, and a metal pipe 10 having a length of approximately λ / 4 connected to the outer conductor at the feeding point 9. As λ
/ 2 dipole, and is a sleeve antenna having the metal pipe 10 as a sleeve. In the vicinity of the first antenna 32, a parasitic element (parameter) which is a parasitic element provided inside the radome 2 as a support means.
Two sitic elements (hereinafter abbreviated as PE) 31 are arranged at target positions substantially parallel to the first antenna 32.
This PE31 has an electrical length of ± 10% with respect to the operating frequency.
The center of the conductive element is located below the feed point 9 and is offset by a predetermined distance. On the other hand, similarly, the second antenna 34
, Two equal-length metal pipes 1 fed by the inner conductor and the outer conductor of the rigid coaxial feed line 8 in the antenna, respectively.
A λ / 2 dipole is formed by using 4 and 15 as radiators,
The length is approximately λ / 2. In the vicinity of the second antenna 34, two PEs 33 provided inside the radome 2 are provided.
The book and the second antenna 34 are arranged at substantially symmetrical positions substantially parallel to the second antenna 34. The PE 33 is a conductive element whose electric length is within a range of ± 10% with respect to the working frequency, and the center thereof is arranged above the feeding point 13 with a predetermined offset. That is, the upper end of the upper parasitic element is located near the feeding point, and the lower end of the lower parasitic element is located near the feeding point.

Here, both antennas 32 and 34 are used as a height (vertical) diversity configuration.
If both antennas are excited at the same frequency and the same phase, the directivity in the vertical plane can be further enhanced by a two-stage stack, and the gain can be increased. Further, the first and second antennas 32 and 34 can be used in two frequency bands. If there is no need, only one set of antennas may be used.

In this configuration, the upper antenna 32 is arranged as shown in Japanese Patent Application No. 5-307846 (FIG. 25).
), A tilt angle occurs in the vertical in-plane directivity,
In the state shown in FIG.
If the length of E is increased, the value of the tilt angle becomes negative (downward angle) gradually decreases as shown in FIG. 3, becomes zero at a certain point, and becomes more positive (upward angle), and gradually increases. . For general use, there is no practical problem if the tilt angle is within ± 10 °. In addition, the position of the PE may be slightly moved up and down from the state shown in FIG. 1 to find the best point. As a result, the length of the PE can be used within a range of about ± 10% with respect to the wavelength, and the tilt angle can be changed by adjusting the length of the PE.

In this case, in the case where the tilt angle is negative, the position of the lower parasitic element is opposite to that of the upper stage, so that the tilt angle is negative in the upper stage and PE in the lower stage is the same as in the upper stage. If the PE length is set so that the tilt angle becomes positive when the positional relationship is established, the tilt angle can be turned upside down so that the lower and upper stages have substantially the same negative value.

As described above, according to the present embodiment, a PE having an electrical length of about ± 10% with respect to the wavelength is provided at least at an equal angle and at equal intervals at one end at substantially the same height as the feed point of the radiator. With this arrangement, it is possible to configure a mobile radio antenna that is omnidirectional in the horizontal plane, has a small tilt angle in the vertical plane with respect to the antenna axis in the vertical plane, and has a small antenna outer shape. The directivity gain in the vertical plane can be increased without using an array configuration.

Next, the operation of the mobile one-body radio antenna of the above embodiment will be described with reference to the drawings.

Paying attention to the second antenna 34 in FIG. 1, the current distribution on the dipole antenna and the PE surface is obtained by the simulation of the moment method, and based on the state, the horizontal omnidirectional mechanism and the tilt can be adjusted only by adjusting the PE length. The mechanism by which the angle can be controlled will be described.

FIG. 4 shows the calculation results of the current distribution. FIG. 4A shows a case where the PE wire diameter is changed, and FIG. 4B shows a case where the PE length is changed. The solid line and the broken line show the prototype antenna. FIG. 4C shows a simulation model, which assumes that the present antenna is applied to the 1.9 GHz band. As shown in FIG. 4, the current on the PE has a phase inverted at the center and has two peaks of amplitude, so that it operates as a one-wavelength resonator. It can be seen that the currents are also opposite to each other. From this, the current on the element is schematically drawn as shown in the center of FIG.

FIG. 5 shows a process of simply modeling the second antenna 34 of FIG. 1 of interest. From this figure, it is assumed that the present antenna is an array of five elements, and the current distribution is A uniform distribution is assumed for both amplitude and phase. Then, a point wave source is placed at the position of the antinode of each current, and an array factor is obtained. The operation mechanism will be considered based on this array factor.

First, the horizontal plane directivity will be described.

FIG. 6 shows a point wave source model when the antenna is viewed from above. At this time, the lower and upper array factors are as follows.

[0023]

(Equation 1) ar ₁ and ar ₂ are the lower and upper current amplitudes, and α ₁ and α ₂ are the phases with respect to the radiator (# 1) of the lower and upper PEs.

7 and 8 show the calculation results of the array factor.
You. FIG. 7 shows a PE radius of 0.15 mm and d = 2 in FIG.
Parameter of current distribution for 0 mm, α₁= 180 °,
ar ₁= 0.4, α_Two= 0, ar_Two= 0.3 was applied
Things. As you can see, the lower array
Kuta (e_l ) Is strong in the antenna axial direction (X direction)
It has a cocoon-shaped directivity that weakens in the angular direction (Y direction).
You. This is because the PE current phase is opposite to the radiator and
This is because radiation is suppressed. On the other hand, the upper array
Kuta (e_h ) Is weak in the X direction and strong in the Y direction, contrary to the lower part.
It's getting worse. In other words, the upper and lower maximum radiation directions are
Are orthogonal to each other, and as a result, the directivity
(E_t ) Is closer to omnidirectional. FIG. 4 (a)
Shows that the gain increases as a result of directivity synthesis
I understand.

FIG. 8 shows the parameters of the current distribution with a PE radius of 1.5 mm, ar ₁ = 0.5, a
This is the case where r ₂ = 0.2. As can be seen from FIG. 4 (a), when the PE is made thicker, the amplitude on the PE is larger at the lower part and smaller at the upper part, and the sum of the amplitude of the radiator and the amplitudes of the two PEs in FIG. .15 mm) and cancel, and the lower array factor emits less radiation in the Y direction. As a result, in the combined directivity ( _et ), the radiation in the Y direction is weakened, and the nondirectionality is lost.

From the above, it is possible to understand the mechanism of omnidirectionality in the horizontal plane and the mechanism in which the directivity changes depending on the PE diameter. In order to realize the omnidirectionality in the horizontal plane, a certain limited thickness is required. It turns out that it needs to be PE.

Next, the vertical plane directivity will be described.

An array factor is determined in the same manner as in the horizontal plane, and the mechanism of beam tilt is considered. FIG. 9 shows a vertical point wave source model. Wave source # 1 is radiator, # 2 and # 3 are PE
The lower part, # 4, and # 5 represent the upper part of the PE, and the lower and upper wave sources of the PE are vertically separated by an interval S. The array factor for this model is as follows:

[0029]

(Equation 2) ar ₁ and ar ₂ are the lower and upper current amplitudes, and α ₁ and α ₂ are the phases with respect to the radiator (# 1) of the lower and upper PEs.

From FIG. 4B, the parameters of the current distribution when the PE length L was changed were obtained as follows. L = 150 mm: α ₁ = 200 °, ar ₁ = 0.3, α ₂ =
20 °, ar ₂ = 0.3 L = 160 mm: α ₁ = 180 °, ar ₁ = 0.4, α ₂ =
0 °, ar ₂ = 0.3 L = 170 mm: α ₁ = 160 °, ar ₁ = 0.3, α ₂ =
-20 °, ar ₂ = 0.2 where S = 95 mm d = 20 mm f = 190
0 MHz FIG. 10 shows the result of calculating the array factor from the above parameters. When L = 160 mm, the maximum radiation direction is the horizontal direction (X direction), and the beam tilt angle is 0 °. On the other hand, the tilt is downward at L = 150 mm and upward at L = 170 mm, which is consistent with the results obtained by the moment method (FIGS. 13, 14, and 17). This indicates that the mechanism of beam tilt can be explained by the simple point wave source model of FIG. Therefore, the phase distribution shown in FIG.

FIG. 11 shows the result of calculating the array factor for various combinations of the wave sources shown in FIG. 9 when L = 150 mm. In FIG. 11, (# 1, # 2, #
3), (# 4, # 5) and (# 2, # 3, # 4, # 5)
The directivity calculated from is oriented in the horizontal or vertical direction. On the other hand, the directivity of (# 1, # 4, # 5) is close to the total directivity, and shows the tilt state well.
Therefore, beam radiator (# 1) and PE
It can be seen that the upper (# 4, # 5) current phases are significantly involved.

From the above, the mechanism of beam tilt can be generally explained as follows.

In FIG. 4B, the change in the phase distribution depending on the PE length L is as shown in FIG. That is, when the phase of the radiator is expressed as a reference, the phase on the PE is a leading phase when L is short and a lagging phase when L is long. At this time, the shape of the phase does not change, and the phase difference between the upper portion and the lower portion is approximately 180 °.
When L = 160 mm, the phases of the radiator and the current above the PE coincide, and the combined directivity is in the horizontal direction. On the other hand, when L = 150 mm, the phase of the upper part of the PE is tilted downward because the phase of the upper part of the PE advances from the radiator, and when L = 170 mm, the phase of the upper part of the PE is conversely delayed and tilted upward.

[0034]

[Table 1] This means that the tilt angle can be controlled only by the PE length. In a diversity configuration, the tilt angle can be made uniform by changing the PE length even in a vertically inverted structure.

In the above embodiment, the input impedance is not always close to the target of 50 ohms, but can be easily set to 50 ohms by matching.

FIG. 15 is a cross-sectional view of the above embodiment.
FIG. 15 (a) shows a method in which metal pipes 42, 43 and PE 44 as radiators are inserted or integrally molded into a spacer 41 made of an insulating material such as Teflon, for example, in a radome 40. It has been inserted. FIG. 15B shows a case where a PE 46 is attached to a radome 45, and a metal wire or a metal plate 46 may be attached as shown in FIG.
As shown in (d), the metal wire 46 may be formed integrally with the radome 45a. Further, as shown in FIG. 15E, a pattern 45b of a conductive substance may be formed by a method such as printing. For maintenance, the inner surface of the radome is desirable, but may be provided on the outer surface. 15 (f) and 15 (g), a method of printing a conductive pattern 46b on the resin film 45b by printing or plating and inserting or attaching the conductive pattern 46b into (or outside) the radome 45 is adopted. You can also.

In the vertical diversity configuration, as the distance between the feeding points between the upper antenna and the lower antenna (diversity interval) increases, the correlation between the two antennas decreases and the diversity effect increases, but the antenna length increases. Becomes Therefore, with the configuration of the present embodiment, the diversity interval can be increased when the height of the entire antenna is the same. FIG. 1 of the present embodiment
When the feeding point interval is 2.5λ (395 mm), the total length of the antenna is about 466 mm, which is 1 unit smaller than that of the conventional antenna.
By making the positions of the PEs of the upper and lower antennas different from each other by 90 ° with respect to the horizontal plane, the correlation between the upper and lower antennas can be further reduced.

In FIG. 2, the coaxial feed line 7 in the antenna is bent in the middle so as to be located at the center of the metal pipe 10 as the radiator at the feed point 9, but as shown in FIG. It may be extended straight upward without being used. There is almost no deterioration in characteristics due to non-coaxiality with the sleeve due to eccentricity, processing is easy in manufacturing, there is no variation, and the cost can be reduced.

Hereinafter, a specific example of the embodiment of the present invention will be described with reference to 1,895 to 1,920M for PHS.
A case in which the present invention is applied to a frequency range of Hz will be described. (Embodiment 2) FIG. 16 shows Embodiment 2 of the present invention, which is constituted by one set of antennas. The radiator had an outer diameter of 3 mm, and the PE had an outer diameter of 0.3 mm. The dimensions of the antenna body excluding the antenna holding bracket and the like were 170.4 mm in length and 40.3 mm in outer diameter. 17 (a) to 17 (d) show various characteristics calculated by the moment method using an expansion function of a trigonometric function and a test function. Set the frequency f under conditions that do not cause tilt.
At r = 1900 MHz, the maximum gain is 2.67 dB.
d, the average gain was 2.40 dBd, and the directivity ripple was 0.47 dBp-p in the horizontal plane. Also Zi
n = 29.6 + j9.1Ω, R _L = 11.1 dB, SW
R = 1.78.

Next, based on the state of two PEs in FIG. 16, the PE length is set to 1.01λ so that the beam tilt angle becomes 0 °.
Constant and the PE offset amount is 0.25λ, 0.35λ,
In each case of 0.5λ, a simulation of changing the PE diameter using the distance between the PE and the radiator as a parameter was performed by the moment method. Hereinafter, data are all based on the wavelength λ for generalization. FIG. 21 shows the case of the PE offset amount of 0.25λ, FIG. 22 shows the case of the PE offset amount of 0.35λ, and FIG.
The case of λ is shown.

FIGS. 21 (a), 22 (a), 23
In (a), the average gain in the XY plane (horizontal plane) with respect to the diameter of PE was determined. If the target average gain is set to 2 dBd or more, the PE offset amount of 0.35λ achieves the target over a wide range of the PE diameter. FIG. 21 (b), FIG.
In (b) and FIG. 23 (b), the directivity ripple with respect to the PE diameter was obtained. With the target value set to 0.5 dB or less, a range within the target under each condition is obtained. FIG.
In (c), FIG. 22 (c), and FIG. 23 (c), the VSWR with respect to the PE diameter was obtained. If the target value is set to 3 or less, the case where the PE offset amount is 0.5λ falls within the target value over the widest range. From the above data, if it is assumed that the target is slightly loosened and the range is widened, if the number of PEs is two, the PE diameter becomes 0.0
Below 1λ, the distance between the PE and the radiator is 0.05λ to 0.
It is practical to find a suitable combination in the range of 2λ and the offset amount of 0.25λ to 0.5λ. Also P
The E length is practically 0.9λ to 1.1λ as a range in which a beam tilt angle of about ± 10 degrees can be obtained from FIG. (Embodiment 3) The above-described Embodiment 2 is a result of a simulation for one set of antennas.
As a third embodiment of the present invention, measurements were made by actually producing an upper and lower two-stage antenna. FIG. 18 is based on FIG.
A choke 37 is inserted between the upper first antenna 35 and the lower second antenna 36 in order to improve the isolation between the upper and lower stages, and two antennas are used to match the input impedance of the lower antenna 36. Are short-circuited by a conductor 38 between the outer conductors between the coaxial power supply lines.

PHS band frequencies 1,895 to 1,
As a result of measurement at 920 MHz, the maximum gain of the upper antenna was 2.97 dBd and the average gain was 2.53 dBd.
The SWR became 1.38 or less. 19A shows the directivity in the horizontal plane of the upper antenna, and FIGS. 19B and 19C show the directivity in the vertical plane.

The lower antenna also has a frequency of 1,89.
As a result of measurement at 5 to 1,920 MHz, the maximum gain was 2.77 dBd and the average gain was 2.16 dBd.
The SWR also became 1.50 or less. FIG. 20A shows the directivity in the horizontal plane of the lower antenna, and FIGS.
(C) shows the directivity in the vertical plane.

In the third embodiment composed of the two sets of antennas, a measurement result similar to the result of the second embodiment simulated for the above-mentioned one set of antennas could be obtained.

In the above embodiment, the number of parasitic elements is two, but three or more parasitic elements may be used. In that case, the directivity in the horizontal plane becomes closer to a perfect circle, and the size can be further reduced.

Also, in the above embodiment, two
Although an example in which a set of antennas is provided is shown, the number of antennas provided in the vertical direction is not limited to this, and may be three or more.

[0047]

As described above, the mobile radio antenna of the present invention is omnidirectional in the horizontal plane, and the tilt angle in the vertical plane with respect to the direction perpendicular to the antenna axis is adjusted by adjusting the PE length. The mobile radio antenna can be freely set within a range of about ± 10 degrees and the antenna outer shape can be reduced, and the directivity gain in the vertical plane can be increased without using a collinear array configuration.

Further, the present invention can be configured so that at least two sets of a vertical dipole antenna and a plurality of parasitic elements are overlapped in the vertical direction, and in that case, the above characteristics can be maintained.

[Brief description of the drawings]

FIG. 1 is a perspective perspective view of a mobile wireless antenna according to a first embodiment of the present invention.

FIG. 2 is a sectional view of the mobile radio antenna according to the first embodiment;

FIG. 3 is a characteristic diagram showing a tilt angle and a maximum gain with respect to a length of a parasitic element in the first embodiment.

4A is a diagram showing a current distribution when the PE wire diameter is changed, FIG. 4B is a diagram showing a current distribution when the PE length is changed, and FIG. Shows the simulation model used to calculate the current distribution

FIG. 5 is a diagram showing a process of simply modeling the second antenna of FIG. 1;

FIG. 6 is a diagram showing a point wave source model for horizontal plane directivity;

FIG. 7 is a diagram showing a calculation result of an array factor in a horizontal plane when a PE radius is 0.15 mm;

FIG. 8 is a diagram showing a calculation result of an array factor in a horizontal plane when the PE radius is 1.5 mm.

FIG. 9 is a diagram showing a point wave source model for vertical plane directivity;

FIG. 10 is a diagram showing a calculation result of an array factor on a vertical plane.

FIG. 11 is a diagram showing a change in directivity due to a combination of wave sources.

FIG. 12 is a diagram showing a change in a phase distribution according to a PE length L;

FIG. 13 is a diagram showing a change in directivity on a horizontal plane depending on a PE diameter.

FIG. 14 is a diagram showing a change in a beam tilt angle depending on a PE length.

FIGS. 15A, 15B, and 15F are perspective views showing the configuration of the parasitic element according to the first embodiment, and FIGS. 15C, 15D, 15E, and 15E; (G) is a sectional view showing the configuration of the parasitic element in the same manner.

FIG. 16 is a perspective view of an element configuration of a mobile radio antenna according to a second embodiment of the present invention.

FIG. 17 is a characteristic diagram by simulation of the mobile radio antenna according to the second embodiment.

FIG. 18 is a perspective view of a mobile wireless antenna according to a third embodiment of the present invention.

FIG. 19 is a measured characteristic diagram of the upper antenna of the mobile radio antenna according to the third embodiment.

FIG. 20 is a measured characteristic diagram of the lower antenna of the mobile wireless antenna according to the third embodiment.

FIG. 21 is a characteristic diagram obtained by simulating the mobile radio antenna according to the second embodiment by changing the parasitic element diameter with the distance from the radiator as a parameter at the parasitic element offset amount of 0.25 wavelength.

FIG. 22 is a characteristic diagram obtained by simulating the mobile radio antenna according to the second embodiment by changing the parasitic element diameter by using the parasitic element offset amount of 0.35 wavelength and the distance from the radiator as a parameter.

FIG. 23 is a characteristic diagram obtained by simulating the mobile radio antenna according to the second embodiment by changing the parasitic element diameter by using the distance from the radiator as a parameter with the parasitic element offset amount of 0.5 wavelength.

FIG. 24 is a side sectional view of a conventional mobile radio antenna.

25 (a) is a perspective view of a conventional mobile radio antenna, FIG. 25 (b) is a characteristic diagram of its horizontal plane directivity, and FIGS. 25 (c) and (d) are vertical plane directivity. Characteristic diagram of sex

[Explanation of symbols]

 2 Radome 3,4 Arm 7a Inner conductor 9,13 Feed point 10,14,15 Metal pipe 11,16,20,22,31,33 Parasitic element 12,21,32,35 First antenna 17,23, 34, 36 second antenna 38 conductor

Claims (11)

[Claims] 1. A vertical dipole antenna having a feeding point, a plurality of parasitic elements, and an insulator, wherein each of the plurality of parasitic elements is substantially parallel to the vertical dipole antenna; Supporting means for supporting the vertical dipole antenna and the plurality of parasitic elements such that the height of the center of each of the parasitic elements is different from the height of the feeding point by a predetermined distance. A mobile wireless antenna, wherein each of the parasitic elements is a linear or belt-shaped conductive element having an electrical length within a range of ± 10% with respect to a used wavelength. 2. The vertical dipole antenna is located between the plurality of parasitic elements, and a horizontal distance between each of the plurality of parasitic elements and the vertical dipole antenna is 0.05 wavelength or less. The mobile radio antenna according to claim 1, wherein the predetermined distance is 0.2 wavelength, and the predetermined distance is a distance corresponding to 0.25 wavelength to 0.5 wavelength. 3. The apparatus according to claim 1, wherein said vertical dipole antenna, said plurality of parasitic elements, and said support means are provided in a plurality of sets, and each set of said plurality of sets is arranged in a vertical direction. Or the mobile radio antenna according to 2. 4. The height of the center of each of the plurality of parasitic elements included in the uppermost set of the plurality of sets is a predetermined distance greater than the height of the feeding point included in the uppermost set. And the height of the center of each of the plurality of parasitic elements included in the lowermost set of the plurality of sets is higher than the height of the feeding point included in the lowermost set by the predetermined distance. The mobile radio antenna according to claim 3, wherein: 5. The mobile radio antenna according to claim 3, wherein a beam tilt angle of each of the plurality of sets is set downward. 6. The mobile radio antenna according to claim 3, wherein an arrangement of the plurality of parasitic elements in a horizontal plane in each of the plurality of sets is different from each other in each set. 7. The mobile radio antenna according to claim 3, wherein the outer conductors of the plurality of sets of coaxial feeders for the vertical dipole antenna are short-circuited to each other. 8. The apparatus according to claim 1, wherein the supporting means has a protection portion parallel to the vertical dipole antenna, and each of the plurality of parasitic elements is attached to a surface of the protection portion. The mobile wireless antenna according to 1. 9. The apparatus according to claim 1, wherein the supporting means has a protection portion parallel to the vertical dipole antenna, and each of the plurality of parasitic elements is formed in the protection portion. The mobile wireless antenna according to 1. 10. The apparatus according to claim 1, wherein the supporting means has a protection portion parallel to the vertical dipole antenna, and each of the plurality of parasitic elements is formed on the protection portion by printing. The mobile wireless antenna according to 1. 11. The supporting means has a plurality of protection portions arranged at equal intervals in parallel with the vertical dipole antenna, and each of the plurality of parasitic elements is formed on a thin film of an insulator by printing or plating. The mobile radio antenna according to claim 1, wherein the antenna is a conductor that is inserted or attached to each of the plurality of protection portions.