JPH1022859A - Direct conversion receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent deterioration of the noise factor characteristic and to prevent occurrence of a receiving fault by controlling the gains of both high frequency and base bands by means of the individual control signals. SOLUTION: This direct conversion receiver offsets the local oscillation frequency of orthogonal detection to the receiving-signal frequency and eliminates the adjacent waves which are dropped into a desired signal band, via an image elimination mixer consisting of a digital multiplier, a digital subtracter and a digital adder. In such a constitution, the base-band gain control amplifiers 8 and 9 are controlled by a control signal 59 that is produced by the output signal of an A/D converter. Then a high-frequency band gain amplifier 2 is controlled by a control signal 50 that is produced by the output signal of a root Nyquist filter, and the amplifiers 2, 8 and 9 are controlled by the individual control signals, respectively.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a receiver for use in a radio equipment used for digital mobile communication and the like, and more particularly, to individually control the gain in a high frequency band and the gain in a base band using individual control signals. In addition, the present invention relates to a receiving apparatus that prevents deterioration of noise figure characteristics and does not cause a reception failure.

[0002]

2. Description of the Related Art FIG. 8 shows a configuration of a conventional direct conversion receiver. In FIG. 8, the direct conversion receiver includes an aerial battle 1 for receiving a signal,
A high-frequency band gain control amplifier 2 for controlling the output level of the received signal, a phase shifter 3 for obtaining a cos wave and a sine wave from the input local oscillation signal, and a mixer for mixing the received signal and the local oscillation signal. Mixers 4 and 5 for obtaining Q signals, low-pass filters 6 and 7 for removing unnecessary frequency components from the I and Q signals, and baseband band gain control amplifiers 8 and 9 for controlling output levels of input signals. And A / D converters 11 and 12 for converting analog signals into digital signals.
Digital multipliers 12, 13, 14, 15, digital subtractors 16, 22, digital adder 17, root Nyquist filters 18, 19 for performing waveform shaping on the I and Q baseband signals, An amplitude detection circuit 20 for detecting amplitude information of a signal, an averaging circuit 21 for performing an averaging process, and a D / A converter 23 for converting a digital signal into an analog signal.
It is composed of

In the direct conversion receiver configured as described above, the output level of the received signal 24 received by the antenna 1 is controlled by the high-frequency band gain control amplifier 2, and a signal 25 is obtained. The signal 25 is input to the mixers 4 and 5, respectively.

[0004] Here, assuming that the frequency of the desired wave is f and the frequency of the local oscillation signal 26 is f-fo, the adjacent wave having the frequency of f + 2fo falls into the desired wave band after the quadrature demodulation. Therefore, consider a case where there is one f-2fo adjacent wave having a frequency other than the desired wave. In this case, each signal 25 is represented by the following equation.

S (t) = {I (t) cos2πft + Q (t) sin2πft} + {II (t) cos2π (f-2fo) t + QQ (t) sin2π (f-2fo) t} (1) Where I (t); baseband I signal (desired wave component) Q (t); baseband Q signal (desired wave component) II (t); baseband I signal (adjacent wave component) QQ (t); base Band Q signal (adjacent wave component)

The local oscillation signal 26 is input to the phase shifter 3,
A cos wave 27 and a sine wave 28 are output and input to the mixers 4 and 5, respectively.

The signal 25 is mixed with the cos wave 27 by the mixer 4 and down-converted to obtain a signal SI (t) 29. The signal SI (t) is expressed by the following equation.

[0008] SI (t) = {I (t) cos2πft + Q (t) sin2πft} cos2π (f-fo) t + {II (t) cos2π (f-2fo) t + QQ (t) sin2π (f- 2fo) t} cos2π (f-fo) t = {I (t) cos2πfot + Q (t) sin2πfot} / 2 + {II (t) cos2πfot-QQ (t) sin2πfot} / 2 + {I (t) cos2π (2f-fo) t + Q (t) sin2π (2f-fo) t} / 2 + {II (t) cos2π (2f-3fo) t + QQ (t) sin2π (2f-3fo) t} / 2 ( 2)

Similarly, the signal 25 is converted by the mixer 5 into sin
Mixed with wave 28 and downconverted to signal SQ
(t) 30 is obtained. The signal SQ (t) is expressed by the following equation.

SQ (t) = {I (t) cos2πft + Q (t) sin2πft} sin2π (f-fo) t + {II (t) cos2π (f-2fo) t + QQ (t) sin2π (f- 2fo) t} sin2π (f-fo) t = {-I (t) sin2πfot + Q (t) cos2πfot} / 2 + {II (t) sin2πfot + QQ (t) cos2πfot} / 2 + {I (t) sin2π (2f-fo) t + Q (t) cos2π (2f-fo) t} / 2 + {II (t) sin2π (2f-3fo) t + QQ (t) sin2π (2f-3fo) t} / 2 (3)

Next, the signal 29 is input to the analog low-pass filter 6, from which unnecessary frequency components are removed, and the signal SSI (t)
31 is obtained. The signal SSI (t) is represented by the following equation.

SSI (t) = {I (t) cos2πfot + Q (t) sin2πfot} / 2 + {II (t) cos2πfot-QQ (t) sin2πfot} / 2 (4)

Similarly, the signal 30 is input to the analog low-pass filter 7, where unnecessary frequency components are removed, and the signal SSQ
(t) 32 is obtained. The signal SSQ (t) is represented by the following equation.

SSQ (t) = {− I (t) sin2πfot + Q (t) cos2πfot} / 2 + {II (t) sin2πfot + QQ (t) cos2πfot} / 2 (5)

The output levels of the signals 31 and 32 are controlled by the baseband gain control amplifiers 8 and 9, respectively, to obtain the signals 33 and 34, respectively.

The signal 33 is converted into a digital signal by the A / D converter 10, and a signal DI (nT) 35 is obtained. Signal DI (n
T) is expressed by the following equation.

DI (nT) = {I (nT) cos2πfonT + Q (nT) sin2πfonT} / 2 + {II (nT) cos2πfonT-QQ (nT) sin2πfonT} / 2 (6) where n = 0,1, 2, ... T; sampling period

Similarly, the signal 34 is converted into a digital signal by the A / D converter 11, and a signal DQ (nT) 36 is obtained.
The signal DQ (nT) is represented by the following equation.

DQ (nT) = {− I (nT) sin2πfonT + Q (nT) cos2πfonT} / 2 + {II (nT) sin2πfonT + QQ (nT) cos2πfonT} / 2 (7)

Next, the signal 35 is supplied to the digital multipliers 12 and 13.
Is multiplied by the cos signal 37 and sin signal 38, respectively.
A signal 39 and a signal 40 are obtained, respectively. Similarly, signal 36
Are cos signals by digital multipliers 14 and 15, respectively.
37 and the sine signal 38 are multiplied to obtain a signal 41 and a signal 42, respectively.

Next, the signal I1 (nT) 39 and the signal Q2 (nT) 42 are inputted to the digital subtractor 16, and the adjacent wave dropped into the desired signal band is removed to obtain the signal I (nT) 43. . Here, in the signal 39 and the signal 42, the desired signal exists as a differential component, but the adjacent wave dropped into the desired signal band exists as an in-phase component, so that the adjacent wave dropped into the desired signal band can be removed. It becomes possible.

Similarly, the signal Q1 (nT) 40 and the signal I2 (nT) 41 are input to the digital adder 17, and the adjacent wave dropped into the desired signal band is removed to obtain the signal Q (nt) 44. . Here, in the signal 40 and the signal 41, the desired signal exists as an in-phase component, but the adjacent wave dropped into the desired signal band exists as a differential component, so that the adjacent wave dropped into the desired signal band can be removed. It becomes possible.

Finally, the signal I (nT) 43 and the signal Q (nT) 44 are waveform-shaped by the root Nyquist filters 18 and 19, and adjacent waves outside the desired signal band are removed. And the baseband Q signal 46 is obtained.

Here, the signals 45 and 46 are connected to the amplitude detection circuit 20.
, Squared, added, and further subjected to a square root operation to obtain a root Nyquist filter output signal.
47 amplitude information signals are obtained. Next, the signal 47 is subjected to an averaging process by the averaging circuit 21, and a signal 48 is obtained.
Next, the signal 48 is converted to the reference signal by the digital subtractor 22.
Subtraction from 49 gives a signal 50. Finally, signal 50
Is converted into an analog signal by the D / A converter 23, and a control signal 51 is obtained. The control signal 51 controls the gains of the high frequency band gain control amplifier 2 and the base band gain control amplifiers 8 and 9.

[0025]

In the direct conversion receiver having the above-described structure, the input signal level of the baseband gain control amplifier is higher than the output signal level of the root Nyquist filter because the adjacent wave cannot be sufficiently removed only by the analog low-pass filter. It becomes larger by the remaining adjacent wave level. Therefore, when the adjacent wave component is much larger than the desired signal component, there is a problem that the baseband gain control amplifier is saturated.

In order to solve such a problem, instead of using a root Nyquist filter output signal from which adjacent waves have been removed, an output signal from an A / D converter in which adjacent waves remain is used. There is a method of generating a control signal for a gain control amplifier. However, the gain figure of the gain control amplifier generally degrades when the gain is reduced, and the bit error rate characteristic greatly deteriorates when the noise figure characteristic of the high frequency band gain control amplifier is deteriorated. Therefore, when this method is used, there is a problem that a reception failure occurs when an adjacent wave is very large with respect to a desired signal.

The present invention solves such a conventional problem, in which a baseband gain control amplifier is controlled by a control signal generated from an A / D converter output signal, and a high frequency band gain control amplifier is controlled by a root Nyquist. The object is achieved by controlling with a control signal generated by a filter output signal and controlling a high-frequency band gain control amplifier and a baseband band gain control amplifier using individual control signals.

[0028]

According to the present invention, a baseband gain control amplifier is controlled by a control signal generated by an A / D converter output signal, and a high frequency band gain amplifier is controlled by a root Nyquist filter output signal. This object is achieved by controlling the high frequency band gain control amplifier and the baseband band gain control amplifier using individual control signals.

[0029]

BEST MODE FOR CARRYING OUT THE INVENTION
A direct control receiver that offsets the local oscillation frequency in quadrature detection with respect to the received signal frequency, and removes the adjacent wave that has fallen into the desired signal band using an image removal mixer composed of a digital multiplier, digital subtractor, and digital adder. A direct conversion receiver, comprising: a high-frequency band amplifier, a mixer, and a baseband band amplifier, wherein the gain of the high-frequency band and the gain of the baseband band are individually controlled using individual control signals. This has the effect of preventing the gain control amplifier from saturating, and preventing the occurrence of reception failure due to the deterioration of the noise figure characteristic of the high frequency band gain control amplifier.

According to a second aspect of the present invention, the high-frequency band gain control amplifier is constituted by using an amplifier and a variable attenuator, and the gain control is performed discretely. This is a conversion receiver, and has an effect that a dynamic range can be further expanded and a highly accurate gain control can be performed as compared with the invention of the first aspect.

According to a third aspect of the present invention, the high-frequency band gain control amplifier is constituted by using an amplifier, a fixed attenuator and a switch, and the amplifier and the fixed attenuator are switched by a switch to perform discrete gain control. The direct conversion receiver according to the second aspect of the present invention has an effect that distortion can be reduced as compared with the invention according to the second aspect.

According to a fourth aspect of the present invention, in addition to the gains of the high frequency band amplifier and the baseband band amplifier, the gain of the mixer is also switched. A direct conversion receiver according to any one of the third aspect, which has an effect that the dynamic range can be further expanded as compared with the invention according to the second aspect.

According to a fifth aspect of the present invention, the image removing mixer comprises a polarity inverter, a multiplexer switch, a digital subtractor, and a digital adder. Claim 3,
A direct conversion receiver according to any one of claims 4 to 6, which has an effect that the amount of calculation can be further reduced as compared with the invention according to claim 1.

The invention according to claim 6 of the present invention is characterized in that gain control of the base band in the base band amplifier is performed discretely.
According to a third aspect of the present invention, there is provided a direct conversion receiver according to any one of the third, fourth, and fifth aspects. It has the effect that it can be performed.

According to a seventh aspect of the present invention, the control of the gain of the mixer is performed using a control signal for controlling the gain of the baseband in the baseband amplifier. The direct conversion receiver according to any one of claims 5 and 6, which has an effect that distortion can be further reduced as compared with the invention according to claim 4.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

(First Embodiment) FIG. 1 shows a first embodiment of the present invention.
3 shows the configuration of the receiving apparatus of the embodiment. In FIG. 1, a receiving apparatus includes an antenna 1 for receiving a signal, a high frequency band gain control amplifier 2 for performing output level control on the received signal, and a phase shift for obtaining a cos wave and a sin wave from an input local oscillation signal. Mixers 3 and mixers 4 and 5 for mixing received signal characteristic local oscillation signals to obtain I and Q signals;
Low-pass filters 6 and 7 for removing unnecessary frequency components from signals, baseband gain control amplifiers 8 and 9 for controlling output levels of input signals, and A / D conversion for converting analog signals to digital signals. Units 10, 11, digital multipliers 12, 13, 14, 15 and digital subtractor 1
6, 22 and 53, a digital adder 17, root Nyquist filters 18 and 19 for shaping the waveforms of the I and Q baseband signals, and amplitude detecting circuits 20 and 52 for detecting amplitude information of the input signal. Averaging circuit 21, which performs averaging processing,
53, D / A to convert digital signal to analog signal
It is composed of converters 23 and 55.

In the direct conversion receiver configured as described above, the output level of the received signal 24 received by the antenna 1 is controlled by the high-frequency band gain control amplifier 2, and a signal 25 is obtained. The signal 25 is input to the mixers 4 and 5, respectively.

Here, assuming that the frequency of the desired wave is f and the frequency of the local oscillation signal 26 is f-fo, the input wave of the desired wave band after quadrature demodulation is an adjacent wave having a frequency of f + 2fo.
Therefore, consider a case where there is one f-2fo adjacent wave having a frequency other than the desired wave. In this case, each signal 25 is represented by the following equation.

S (t) = {I (t) cos2πft + Q (t) sin2πft} + {II (t) cos2π (f-2fo) t + QQ (t) sin2π (f-2fo) t} (1) Where I (t); baseband I signal (desired wave component) Q (t); baseband Q signal (desired wave component) II (t); baseband I signal (adjacent wave component) QQ (t); base Band Q signal (adjacent wave component)

The local oscillation signal 26 is input to the phase shifter 3,
A cos wave 27 and a sine wave 28 are output and input to the mixers 4 and 5, respectively.

The signal 25 is mixed with the cos wave 27 by the mixer 4 and down-converted to obtain a signal SI (t) 29. The signal SI (t) is expressed by the following equation.

SI (t) = {I (t) cos2πft + Q (t) sin2πft} cos2π (f-fo) t + {II (t) cos2π (f-2fo) t + QQ (t) sin2π (f- 2fo) t} cos2π (f-fo) t = {I (t) cos2πfot + Q (t) sin2πfot} / 2 + {II (t) cos2πfot-QQ (t) sin2πfot} / 2 + {I (t) cos2π (2f-fo) t + Q (t) sin2π (2f-fo) t} / 2 + {II (t) cos2π (2f-3fo) t + QQ (t) sin2π (2f-3fo) t} / 2 ( 2)

Similarly, the signal 25 is converted by the mixer 5 into sin
Mixed with wave 28 and downconverted to signal SQ
(t) 30 is obtained. The signal SQ (t) is expressed by the following equation.

SQ (t) = {I (t) cos2πft + Q (t) sin2πft} sin2π (f-fo) t + {II (t) cos2π (f-2fo) t + QQ (t) sin2π (f- 2fo) t} sin2π (f-fo) t = {-I (t) sin2πfot + Q (t) cos2πfot} / 2 + {II (t) sin2πfot + QQ (t) cos2πfot} / 2 + {I (t) sin2π (2f-fo) t + Q (t) cos2π (2f-fo) t} / 2 + {II (t) sin2π (2f-3fo) t + QQ (t) sin2π (2f-3fo) t} / 2 (3)

Next, the signal 29 is input to the analog low-pass filter 6, where unnecessary frequency components are removed, and the signal SSI (t)
31 is obtained. The signal SSI (t) is represented by the following equation.

SSI (t) = {I (t) cos2πfot + Q (t) sin2πfot} / 2 + {II (t) cos2πfot-QQ (t) sin2πfot} / 2 (4)

Similarly, the signal 30 is input to the analog low-pass filter 7, where unnecessary frequency components are removed, and the signal SSQ
(t) 32 is obtained. The signal SSQ (t) is represented by the following equation.

SSQ (t) = {− I (t) sin2πfot + Q (t) cos2πfot} / 2 + {II (t) sin2πfot + QQ (t) cos2πfot} / 2 (5)

The output levels of the signals 31 and 32 are controlled by the baseband gain control amplifiers 8 and 9, respectively, to obtain the signals 33 and 34, respectively.

The signal 33 is converted into a digital signal by the A / D converter 10, and a signal DI (nT) 35 is obtained. Signal DI (n
T) is expressed by the following equation.

DI (nT) = {I (nT) cos2πfonT + Q (nT) sin2πfonT} / 2 + {II (nT) cos2πfonT-QQ (nT) sin2πfonT} / 2 (6) where n = 0,1, 2, ... T; sampling period

Similarly, the signal 34 is converted into a digital signal by the A / D converter 11, and a signal DQ (nT) 36 is obtained.
The signal DQ (nT) is represented by the following equation.

DQ (nT) = {− I (nT) sin2πfonT + Q (nT) cos2πfonT} / 2 + {II (nT) sin2πfonT + QQ (nT) cos2πfonT} / 2 (7)

Next, the signal 35 is supplied to the digital multipliers 12 and 13.
Is multiplied by the cos signal 37 and sin signal 38, respectively.
A signal 39 and a signal 40 are obtained, respectively. Similarly, signal 36
Are cos signals by digital multipliers 14 and 15, respectively.
37 and the sine signal 38 are multiplied to obtain a signal 41 and a signal 42, respectively.

Next, the signal I1 (nT) 39 and the signal Q2 (nT) 42 are input to the digital subtractor 16, and the adjacent wave dropped into the desired signal band is removed to obtain the signal I (nT) 43. . Here, in the signal 39 and the signal 42, the desired signal exists as a differential component, but the adjacent wave dropped into the desired signal band exists as an in-phase component, so that the adjacent wave dropped into the desired signal band can be removed. It becomes possible.

Similarly, the signal Q1 (nT) 40 and the signal I2 (nT) 41 are input to the digital adder 17, and the adjacent wave dropped into the desired signal band is removed to obtain the signal Q (nt) 44. . Here, in the signal 40 and the signal 41, the desired signal exists as an in-phase component, but the adjacent wave dropped into the desired signal band exists as a differential component, so that the adjacent wave dropped into the desired signal band can be removed. It becomes possible.

Finally, the signal I (nT) 43 and the signal Q (nT) 44 are waveform-shaped by the root Nyquist filters 18 and 19, and adjacent waves outside the desired signal band are removed. And the baseband Q signal 46 is obtained.

Here, the signal 35 and the signal 36 are connected to the amplitude detection circuit 52.
, Are squared, added, and further subjected to a square root operation, whereby an amplitude information signal of the A / D converter output signal 56 is obtained. Next, the signal 56 is subjected to an averaging process by the averaging circuit 53, and a signal 57 is obtained. Next, signal 57
Is subtracted from a reference signal 58 by a digital subtractor 54 to obtain a signal 59. Finally, the signal 59 is converted into an analog signal by the D / A converter 55, and a control signal 60 is obtained. Control signal 60 controls the gain of baseband gain control amplifiers 8 and 9.

Similarly, the signals 45 and 46 are output from the amplitude detection circuit 20.
, Squared, added, and further subjected to a square root operation to obtain a root Nyquist filter output signal.
47 amplitude information signals are obtained. Next, the signal 47 is subjected to an averaging process by the averaging circuit 21, and a signal 48 is obtained.
Next, the signal 48 is converted to the reference signal by the digital subtractor 22.
Subtraction from 49 gives a signal 50. Finally, signal 50
Is converted into an analog signal by the D / A converter 23, and a control signal 51 is obtained. The control signal 51 controls the gains of the high frequency band gain control amplifier 2 and the base band gain control amplifiers 8 and 9.

As described above, according to the present invention, the baseband gain control amplifier is controlled by the control signal generated by the A / D converter output signal, and the high frequency band gain control amplifier is controlled by the root Nyquist filter output signal. And control the high-frequency band gain control amplifier and the baseband band gain control amplifier using separate control signals, thereby preventing the gain control amplifier from being saturated and deteriorating the noise figure characteristics of the high-frequency band gain control amplifier. This can prevent a reception failure from occurring.

(Second Embodiment) FIG. 2 shows a second embodiment of the present invention.
3 is a configuration of a receiving device according to the embodiment. The difference between the receiving device of the second embodiment and the receiving device of the first embodiment is that the receiving device of the second embodiment has an amplifier 61 and a variable attenuator 62 instead of the high frequency band gain control amplifier 2.

Here, the members, signals, and the like of the receiving apparatus according to the second embodiment corresponding to the members, signals, and the like described in FIG. 1 are denoted by the same reference numerals, and detailed description is omitted.

The operation of the receiving apparatus according to the second embodiment will be described with reference to FIG.

The received signal 24 received from the antenna 1 is amplified by the amplifier 61 to obtain a signal 64. Signal 64 is
The level is controlled by the variable attenuator 62, and the signal 25 is obtained.

The subsequent steps up to obtaining a baseband I signal 21 and a baseband Q signal 22 are the same as those of the receiving apparatus of the first embodiment.

Further, the baseband gain control amplifier 8,
9 is the same as that of the receiving apparatus of the first embodiment. The generation of the control signal for controlling the variable attenuator 62 is the same as that of the receiving apparatus of the first embodiment until the signal 50 is obtained. Signal 50
Is determined by the determination circuit 63, and a control signal 65 is obtained. The control signal 65 controls the variable attenuator 62. In the configuration of the receiving apparatus according to the second embodiment, a high-frequency band gain control amplifier is configured using an amplifier and a variable attenuator, and gain control is performed discretely. The dynamic range can be further expanded as compared with the configuration described above, and highly accurate gain control can be performed.

(Third Embodiment) FIG. 3 shows a third embodiment of the present invention.
3 is a configuration of a receiving device according to the embodiment. The receiver of the third embodiment is different from the receiver of the second embodiment in that switches 66 and 68, an amplifier 61, and a fixed attenuator 67 are provided instead of the amplifier 61 and the variable attenuator 62. In the configuration.

Here, members, signals, and the like of the receiving apparatus of the third embodiment corresponding to the members, signals, and the like described in FIG. 1 are denoted by the same reference numerals, and detailed description thereof will be omitted.

The received signal 24 received by the antenna 1 is
When the signal is input to the amplifier 61 by the switch 66, the signal is amplified by the amplifier 61 and a signal 64 is obtained. When the signal is input to the fixed attenuator 67, the signal is attenuated by the fixed attenuator 67, and a signal 69 is obtained. Switch 68 connects signal 64 and signal 69
, And a signal 25 is obtained.

The subsequent steps up to obtaining the baseband I signal 21 and the baseband Q signal 22 are the same as those of the receiving apparatus of the second embodiment.

The generation of the control signal is the same as that of the receiving apparatus according to the second embodiment.

Generally, the distortion due to the switch is smaller than the distortion due to the variable attenuator. In addition, since the fixed attenuator can be constituted only by a resistor without using a semiconductor element, distortion does not basically occur. Therefore, in the configuration of the receiving apparatus according to the third embodiment, distortion can be reduced as compared with the configuration of the receiving apparatus according to the second embodiment.

As described above, in the configuration of the receiving apparatus according to the third embodiment, by using the switch, the amplifier, and the fixed attenuator, the distortion is smaller than that of the receiving apparatus according to the second embodiment. Can be reduced.

(Fourth Embodiment) FIG. 4 shows the configuration of a receiving apparatus according to a fourth embodiment of the present invention. The difference between the receiver of the fourth embodiment and the receiver of the second embodiment is that the gain of the mixer is switched in addition to the gain of the high-frequency amplifier and the baseband amplifier.

Here, members, signals, and the like of the receiving apparatus of the fourth embodiment corresponding to the members, signals, and the like described in FIG. 2 are denoted by the same reference numerals, and detailed description is omitted.

FIG. 4 shows the operation of the receiving apparatus according to the fourth embodiment.
This will be described with reference to FIG.

Except that the gains of the mixers 4 and 5 are switched by the control signal 65, this is the same as the receiving apparatus of the second embodiment.

In the configuration of the receiving apparatus according to the fourth embodiment, the gain of the mixer is switched in addition to the gain of the high-frequency band amplifier and the baseband amplifier, so that the configuration of the receiving apparatus of the second embodiment is changed. The dynamic range can be further expanded.

(Fifth Embodiment) FIG. 5 shows a fifth embodiment of the present invention.
3 is a configuration of a receiving device according to the embodiment. The difference between the receiving device of the fifth embodiment and the receiving device of the first embodiment is that instead of the digital multipliers 12, 13, 14, and 15, polarity inverters 70, 71, 72, and 73 are used. And multiplexer switches 74, 75, 76 and 77.

Here, the members, signals, and the like of the receiving apparatus of the fifth embodiment corresponding to the members, signals, and the like described in FIG. 1 are denoted by the same reference numerals, and detailed description is omitted.

FIG. 5 shows the operation of the receiving apparatus according to the fifth embodiment.
This will be described with reference to FIG.

Up to obtaining the signal DI (nT) 35 and the signal DQ (nT) 36, the operation is the same as that of the receiving apparatus of the first embodiment.

The signal 35 is inverted in polarity by the polarity inverters 70 and 71 to obtain a signal 78 and a signal 79, respectively. Similarly,
The polarity of the signal 36 is inverted by the polarity inverters 72 and 73, and a signal 80 and a signal 81 are obtained.

Next, the signal 35 and the signal 78 are selected and output in chronological order at the timing of the sampling cycle by the multiplexer switch 74, and the signal I1 (nT) 39 is obtained. Signal 39
Is represented by the following equation.

I1 (nT) = DI (nT); n = 4k 0; n = 4k + 1-DI (T); n = 4k + 20; n = 4k + 3 (8) where k = 0, 1,2 ...

Here, when the signal I1 (nT) is multiplied by cos2πfonT, I1 (nT) cos2πfonT is obtained. However, the signal when fo = 1 / 4T (when the oversampling of one cycle of the local signal is 4) is obtained. I1 (nT) cos2πfonT can be modified as in the above equation (8). Therefore, the above equation (8) becomes I1 (nT).
This is equivalent to multiplying by cos2πfonT.

Similarly, the signal 35 and the signal 79, the signal 36 and the signal 80, and the signal 36 and the signal 81 are selected and output in chronological order by the multiplexer switches 75, 76, and 77 at the timing of the sampling cycle, respectively. nT) 40, signal
I2 (nT) 41 and signal Q2 (nT) 42 are obtained. The signals 40, 41, and 42 are represented by the following equations, respectively.

Q1 (nT) = 0; n = 4k DI (nT); n = 4k + 10; n = 4k + 2-DI (nT); n = 4k + 3 (9) I2 (nT) = DQ (nT); n = 4k 0; n = 4k + 1 -DQ (nT); n = 4k + 30; n = 4k + 3 (10) Q2 (nT) = 0; n = 4k DQ (nT); n = 4k + 1 0; n = 4k + 2-DQ (nT); n = 4k + 3 (11)

[0093] The subsequent steps up to obtaining the baseband I signal 21 and the baseband Q signal 22 are the same as those of the receiving apparatus of the second embodiment.

The generation of the control signal is the same as that of the receiving apparatus of the first embodiment.

In the configuration of the receiving apparatus according to the fifth embodiment, the image removing mixer is constituted by a polarity inverter, a multiplexer switch, a digital subtractor and a digital adder without using a digital multiplier. The amount of calculation can be further reduced as compared with the configuration of the receiving device of the first embodiment.

(Sixth Embodiment) FIG. 6 shows a sixth embodiment of the present invention.
3 is a configuration of a receiving device according to the embodiment. The difference between the receiver of the sixth embodiment and the receiver of the second embodiment is that, instead of the gain control amplifiers 8 and 9, switches 82, 83, 84 and 85 and amplifiers 87 and 89 are fixed. Attenuator 86, 88
In the configuration provided with.

Here, members, signals, and the like of the receiving apparatus of the sixth embodiment corresponding to the members, signals, and the like described in FIG. 2 are denoted by the same reference numerals, and detailed description thereof will be omitted.

Up to obtaining signals 31 and 32, the operation is the same as that of the receiving apparatus of the second embodiment. The signal 31 is switched by a switch 82 between an input to an amplifier 87 and an input to a fixed attenuator 86. When the signal 31 is input to the amplifier 87, the signal 31 is amplified by the amplifier 87, and a signal 92 is obtained. When the signal is input to the fixed attenuator 86, the signal is attenuated by the fixed attenuator 86, and a signal 91 is obtained. switch
84 switches and outputs the signal 91 and the signal 92, and the signal 33 is obtained.

Similarly, the signal 83 is switched by a switch 83 between input to an amplifier 89 and input to a fixed attenuator 88. The signal 32 is switched between input to the amplifier 89 and input to the fixed attenuator 88. When the signal 32 is input to the amplifier 89, the signal 32 is amplified by the amplifier 89 to obtain a signal 94. When the signal is input to the fixed attenuator 88, the signal is attenuated by the fixed attenuator 88 and a signal 93 is obtained.
The switch 85 switches and outputs the signal 93 and the signal 94, and the signal 34 is obtained.

[0097] The subsequent steps until the baseband I signal 21 and the baseband Q signal 22 are obtained are the same as those of the receiving apparatus of the second embodiment.

The generation of a control signal for performing gain control in a high frequency band is the same as that of the receiving apparatus according to the second embodiment.

The generation of the control signal for performing the gain control in the base band is the same as that of the receiving apparatus of the second embodiment until the signal 59 is obtained. Signal 59 is
The signal is determined by the determination circuit 90, and a signal 95 is obtained. signal
95 controls the switches 82, 83, 84, 85.

As described above, in the configuration of the receiving apparatus of the sixth embodiment, the configuration of the switch, the amplifier, and the fixed attenuator makes it possible to further improve the configuration of the receiving apparatus of the third embodiment. , The dynamic range can be expanded, and highly accurate gain control can be performed.

(Seventh Embodiment) FIG. 7 shows the configuration of a receiving apparatus according to a seventh embodiment of the present invention. The receiving apparatus according to the seventh embodiment differs from the receiving apparatus according to the fourth embodiment in that the gain control of the mixer is controlled using a control signal for performing gain control in a baseband.

Here, the members, signals, and the like of the receiving apparatus of the fourth embodiment corresponding to the members, signals, and the like described in FIG. 2 are denoted by the same reference numerals, and detailed description will be omitted.

The operation of the receiving apparatus according to the seventh embodiment is shown in FIG.
This will be described with reference to FIG. Except that the gains of the mixers 4 and 5 are controlled using a control signal 60 for performing gain control in the baseband, the configuration is the same as that of the receiving apparatus of the fourth embodiment.

In the configuration of the receiving apparatus according to the seventh embodiment, the gain control of the mixer is controlled by using a control signal for controlling the gain in the baseband, whereby the receiving apparatus according to the fourth embodiment is controlled. The distortion can be further reduced as compared with the configuration described above.

[0105]

According to the present invention, as is apparent from the description of the above-described embodiment, the baseband gain control amplifier is provided with an A / D converter.
The high-frequency band gain control amplifier is controlled by the control signal generated by the root Nyquist filter output signal, and the high-frequency band gain control amplifier and the baseband band gain control amplifier are individually controlled by the control signal generated by the converter output signal. , The gain control amplifier is prevented from saturating, and there is an effect that it is possible to prevent the occurrence of reception failure due to the deterioration of the noise figure characteristic of the high frequency band gain control amplifier.

[Brief description of the drawings]

FIG. 1 is a configuration diagram of a direct conversion receiver according to a first embodiment of the present invention;

FIG. 2 is a configuration diagram of a direct conversion receiver according to a second embodiment of the present invention;

FIG. 3 is a configuration diagram of a direct conversion receiver according to a third embodiment of the present invention;

FIG. 4 is a configuration diagram of a direct conversion receiver according to a fourth embodiment of the present invention;

FIG. 5 is a configuration diagram of a direct conversion receiver according to a fifth embodiment of the present invention;

FIG. 6 is a configuration diagram of a direct conversion receiver according to a sixth embodiment of the present invention;

FIG. 7 is a configuration diagram of a direct conversion receiver according to a seventh embodiment of the present invention;

FIG. 8 is a configuration diagram of a conventional direct conversion receiver.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Air combat 2 High frequency band gain control amplifier 3 Phase shifter 4, 5 Mixer 6, 7 Analog low-pass filter 8, 9 Baseband band gain control amplifier 10, 11 A / D converter 12, 13, 14, 15 Digital multiplier 16, 22, 54 Digital subtractor 17 Digital adder 18, 19 Root Nyquist filter 20, 52 Amplitude information detection circuit 21, 53 Averaging circuit 23, 55 D / A converter 24 Received signal 26 Local oscillation signal 29 Analog I signal 30 Analog Q signal 34 Digital I signal 35 Digital Q signal 51,60 Gain control signal 61,87,89 Amplifier 62 Variable attenuator 63,90 Judgment circuit 66,68,82,83,84,85 Switch 67,86,88 Fixed attenuator 70, 71, 72, 73 Polarity inverter 74, 75, 76, 77 Multiplexer switch

Claims (7)

[Claims] 1. A local oscillation frequency in quadrature detection is offset with respect to a reception signal frequency, and an adjacent wave dropped into a desired signal band is removed by an image removal mixer including a digital multiplier, a digital subtractor, and a digital adder. In a direct control receiver, a high-frequency band amplifier, a mixer, and a baseband band amplifier are provided.
A direct conversion receiver characterized in that individual control is performed using individual control signals. 2. The direct conversion receiver according to claim 1, wherein said high-frequency band gain control amplifier is constituted by using an amplifier and a variable attenuator, and performs gain control discretely. 3. The gain control amplifier according to claim 2, wherein the high-frequency band gain control amplifier comprises an amplifier, a fixed attenuator and a switch, and the amplifier and the fixed attenuator are switched by a switch to perform discrete gain control. Direct conversion receiver as described. 4. The direct conversion according to claim 1, wherein the gain of the mixer is switched in addition to the gains of the high-frequency band amplifier and the baseband band amplifier. Receiving machine. 5. The image removing mixer according to claim 1, wherein the image removing mixer comprises a polarity inverter, a multiplexer switch, a digital subtractor, and a digital adder. Direct conversion receiver according to 1. 6. The baseband amplifier according to claim 1, wherein gain control in a baseband band is performed discretely.
Direct conversion receiver according to any of the above. 7. The control device according to claim 4, wherein the control of the gain of said mixer is performed using a control signal for controlling the gain of a base band in said base band amplifier. Direct conversion receiver described in any of them.